LU506246B1 - Method for fully-digital phase noise measurement with millihertz-level frequency resolution - Google Patents
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- G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
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Abstract
The present disclosure relates to the field of phase noise measurement, and provides a method for fully-digital phase noise measurement with millihertz-level frequency resolution. The method includes: converting a measured signal into a digital signal, generating two digital signals with orthogonal phases and no phase truncation spur based on a frequency control word of the digital signal, mixing the two digital signals with the digital signal separately, and performing multi-segment decimation and filtering, phase unwrapping, frequency offset elimination, power spectrum estimation, and segmental splicing. The method uses only one analog-to-digital converter (ADC) to measure phase noise of a signal, which has a simple implementation principle and is easy to achieve a miniaturized system. The multi-segment decimation and filtering can reduce a data rate after mixing while filtering out a high-frequency component.
Description
LU506246
METHOD FOR FULLY-DIGITAL PHASE NOISE MEASUREMENT WITH
MILLIHERTZ-LEVEL FREQUENCY RESOLUTION
[0001] The present disclosure relates to the field of phase noise measurement, and more specifically, to a method for fully-digital phase noise measurement with millihertz-level frequency resolution.
[0002] In some special occasions, a signal in a 10k Hz to 500k Hz frequency band has a measurement requirement for millihertz-level frequency resolution. For example, in the "Tiangin Project", a plurality of key technologies need to be implemented in order to detect cosmic gravitational waves in a 0.001 Hz to 0.1 Hz frequency band. À capacitive displacement sensor is one of key components, and its sensing performance plays an important role in sensitivity of gravitational wave detection. In the capacitive displacement sensor, one carrier needs to be injected into a plurality of capacitive plates around a test mass (TM). If a carrier used for modulation has poor frequency stability a millinertz-level low-frequency differential capacitive signal may be covered by phase noise of the carrier.
Therefore, it is of great research significance to measure phase noise extremely close-in to the carrier (millihertz frequency offset).
[0003] However, no phase noise instruments currently on the market can achieve millihertz-level measurement for phase noise extremely close-in to a signal in the 10k Hz to 500k Hz frequency band.
LU506246
[0004] In view of the above defects or improvement requirements in the prior art, the present disclosure provides a fully-digital phase noise measurement method with millihertz-level frequency resolution, to resolve a technical problem that an existing phase noise instrument cannot achieve millihertz-level measurement for phase noise extremely close-in to a signal in a 10k Hz to 500k Hz.
[0005] In order to achieve the above objective, according to a first aspect of the present disclosure, a method for fully-digital phase noise measurement with millihertz-level measurement frequency resolution is provided, including:
[0006] S1: converting a measured signal into a digital signal, and obtaining a frequency control word of the digital signal;
[0007] S2: generating two digital signals V,7 and Vp2 with orthogonal phases and no phase truncation spur based on the frequency control word, and mixing the Vhs and the Vh2 with the digital signal to obtain V,7 and V,2 respectively:
[0008] S3: separately performing multi-segment decimation and filtering on the Var and the V,2' to obtain a filtered signal of the V;7 and a filtered signal of the Vp2' under each decimation factor; and under each decimation factor, performing division and four-quadrant arctangent on the filtered signal of the V;7 and the filtered signal of the V,2 to obtain a corresponding initial phase difference sequence, and then performing phase unwrapping and frequency offset elimination on the initial phase difference sequence in turn to obtain a corresponding target phase difference sequence; and
[0009] S4: performing power spectrum estimation on the target phase difference sequence under each decimation factor to obtain a single-sideband phase noise spectral density of the target phase difference sequence, and splicing the single-sideband phase noise spectral density of the target phase difference sequence under each decimation factor to obtain a complete phase noise measurement curve.
[0010] According to a second aspect of the present disclosure, a device for fully-digital phase noise measurement with millihertz-level measurement 7906246 frequency resolution is provided, including:
[0011] a first processing module configured to convert a measured signal into a digital signal, and obtain a frequency control word of the digital signal;
[0012] a second processing module configured to generate two digital signals Vas and Vn2 with orthogonal phases and no phase truncation spur based on the frequency control word, and mix the Vhs and the Viz with the digital signal to obtain
Vhs and Vp2 respectively;
[0013] a third processing module configured to: separately perform multi-segment decimation and filtering on the Vm and the Vp2' to obtain a filtered signal of the
Var and a filtered signal of the V,2" under each decimation factor; and under each decimation factor, perform division and four-quadrant arctangent on the filtered signal of the Vhs and the filtered signal of the Vx." to obtain a corresponding initial phase difference sequence, and then perform phase unwrapping and frequency offset elimination on the initial phase difference sequence in turn to obtain a corresponding target phase difference sequence; and
[0014] a fourth processing module configured to perform power spectrum estimation on the target phase difference sequence under each decimation factor to obtain a single-sideband phase noise spectral density of the target phase difference sequence, and splice the single-sideband phase noise spectral density of the target phase difference sequence under each decimation factor to obtain a complete phase noise measurement curve.
[0015] According to a third aspect of the present disclosure, a system for fully-digital phase noise measurement with millihertz-level measurement frequency resolution is provided, including a computer-readable storage medium and a processor, where
[0016] the computer-readable storage medium is configured to store an executable instruction; and
[0017] the processor is configured to read the executable instruction stored in the computer-readable storage medium to execute the method described in the first aspect.
[0018] According to a fourth aspect, a computer-readable storage medium is provided, where the computer-readable storage medium stores a computer 7906246 instruction, and the computer instruction is configured to enable a processor to execute the method in the first aspect.
[0019] In general, compared with the prior art, the above technical solutions conceived by the present disclosure can achieve following beneficial effects:
[0020] 1. The method provided in the present disclosure uses only one analog-to-digital converter (ADC) to measure phase noise of a signal, which has a simple implementation principle and is easy to achieve a miniaturized measurement system, thereby facilitating a site test. Multi-segment decimation and filtering can reduce a data rate after mixing while filtering out a high-frequency component. This reduces a quantity of operation points required for power spectrum estimation, thus achieving measurement with millimeter-level frequency resolution for phase noise extremely close-in to the signal. For example, phase noise extremely close-in to a signal in a 10k Hz to 500k Hz frequency band can be measured, making up for a deficiency that a commercial instrument cannot achieve the millimeter-level frequency resolution for the signal in the frequency band.
[0021] 2. The method provided in the present disclosure can generate two mutually orthogonal signals with no phase truncation spur by using a low-spurious orthogonal digital signal synthesis technology, which can reduce measurement interference caused by spurs.
[0022] 3. The method provided in the present disclosure can reduce an impact caused by a slight frequency difference between a measured signal and a synthetic signal by using phase unwrapping and frequency offset elimination algorithms.
LU506246
[0023] FIG. 1 is a flowchart of a method for fully-digital phase noise measurement with millihertz-level frequency resolution according to an embodiment of the present disclosure;
[0024] FIG. 2 is a schematic diagram of fully-digital phase noise measurement according to according to an embodiment of the present disclosure;
[0025] FIG. 3 is a schematic diagram of equal-precision frequency measurement;
[0026] FIG. 4 is a schematic diagram of a low-spurious orthogonal digital signal synthesis method;
[0027] FIG. 5 is a schematic diagram of comparing a single-sideband phase noise spectral density of a 500 kHz signal according to an embodiment of the present disclosure; and
[0028] FIG. 6 is a schematic diagram of splicing a single-sideband phase noise spectral density of a 500 kHz signal according to an embodiment of the present disclosure.
[0029] In order to make the objectives, technical solutions, and advantages of the present disclosure clearer, the present disclosure is further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are only intended to explain the present disclosure and are not intended to limit the present disclosure. Further, the technical features involved in the various implementations of the present disclosure described below may be combined with each other as long as they do not constitute a conflict with each other.
[0030] In a plurality of phase noise measurement techniques, a method of directly using a spectrum analyzer is affected by background noise of the spectrum analyzer itself and is not suitable for measuring a signal with high frequency stability. A beat method is difficult to achieve measurement with millihertz-level frequency resolution extremely close-in to the signal due to a limitation of a counter. A frequency discrimination method has poor sensitivity when measuring near-end phase noise.
[0031] A phase discrimination method has many physical devices and a large-volume phase discrimination system, and needs to perform compensation 7906246 for a loss of a low frequency. A fully-digital method is free from the limitations above, and has advantages of a simple implementation principle, easy implementation of a miniaturized system, and convenient field testing.
[0032] In view of this, an embodiment of the present disclosure provides a method for fully-digital phase noise measurement with millihertz-level signal frequency resolution. As shown in FIG. 1, the method includes following steps.
[0033] S1: A measured signal is converted into a digital signal, and a frequency control word of the digital signal is obtained.
[0034] Specifically, a to-be-measured signal (for example, a sine wave) is sampled by an ADC and converted into a digital signal, and a frequency control word of the digital signal is obtained by using an equal-precision frequency measurement method, a direct measurement method, or a periodic measurement method.
[0035] When equal-precision frequency measurement is adopted, the frequency control word of the to-be-measured signal can be expressed by count values of a standard signal and the to-be-measured signal.
[0036] S2: Two digital signals Vhs and Vr2 with orthogonal phases and no phase truncation spur are generated based on the frequency control word, and are mixed with the digital signal to obtain V,7 and Vn2 respectively.
[0037] Preferably, in the step S2, the Vis and the Vi. are generated by using a phase truncation-free frequency synthesis method, a ROM table compression frequency synthesis method, or a coordinate rotation digital computer (CORDIC) frequency synthesis method.
[0038] Specifically, based on the frequency control word generated through the equal-precision frequency measurement, a low-spurious orthogonal digital signal synthesis technology can be used to generate two low-spurious digital signals that have orthogonal phases and whose frequencies are similar to a frequency of the to-be-measured signal, which can reduce interference of a spur on phase noise measurement of the to-be-measured signal.
[0039] S3: Multi-segment decimation and filtering is separately performed on the Var and the V,2' to obtain a filtered signal of the V;7 and a filtered signal of the Vp2' 0000068 under each decimation factor; and under each decimation factor, division and four-quadrant arctangent are performed on the filtered signal of the V;7 and the filtered signal of the Vi, to obtain a corresponding initial phase difference sequence, and then phase unwrapping and frequency offset elimination are performed on the initial phase difference sequence in turn to obtain a corresponding target phase difference sequence.
[0040] Specifically, the two signals obtained through low-spurious orthogonal digital signal synthesis are separately mixed with the to-be-measured signal in a digital domain. Through the multi-segment decimation and filtering, a high-frequency component after the mixing can be filtered out, and a low-frequency signal after the mixing can be obtained. In addition, multi-segment decimation can achieve multi-resolution measurement, which can reduce a quantity of data points required to achieve millinertz-level frequency resolution.
[0041] S4: Power spectrum estimation is performed on the target phase difference sequence under each decimation factor to obtain a single-sideband phase noise spectral density of the target phase difference sequence, and the single-sideband phase noise spectral density of the target phase difference sequence under each decimation factor is spliced to obtain a complete phase noise measurement curve.
[0042] Specifically, the division and the four-quadrant arctangent are performed on the two signals obtained after the multi-segment decimation and filtering to obtain the initial phase difference sequence. The phase unwrapping is performed on the phase difference sequence to recover a jump of the phase difference sequence at
TT.
[0043] The digital signal generated by the low-spurious orthogonal digital signal synthesis technology has a slight frequency difference from the measured signal.
In order to eliminate a measurement error caused by the frequency difference, the frequency offset elimination needs to be performed on a phase obtained after the phase unwrapping sequence.
[0044] Preferably, the step S4 includes: segmenting a frequency range offset from the carrier frequency, and selecting different sampling frequencies for different biased carrier frequency bands to achieve the power spectrum estimation for the frequency bands.
[0045] The method provided in the present disclosure will be explained in detail below with reference to FIG. 2. 17000668
[0046] As shown in FIG. 2, the method for fully-digital phase noise measurement with millihertz-level frequency resolution provided in the present disclosure mainly includes following steps: collection by an ADC, equal-precision frequency measurement, low-spurious orthogonal digital signal synthesis, multi-segment decimation and filtering, four-quadrant arctangent, phase unwrapping, frequency offset elimination, power spectrum estimation, and segmental splicing.
[0047] Step 1: A measured signal is converted into a digital signal by using the ADC.
[0048] Step 2: À frequency control word of the digital signal is obtained by using an equal-precision frequency measurement method, to provide a variable frequency reference for next-stage low-spurious orthogonal digital signal synthesis.
[0049] Step 3: Two digital signals with a phase difference of 1/2 and no phase truncation spur are generated by using a low-spurious orthogonal digital signal synthesis method based on the frequency control word provided in the step 2. The two signals will be mixed with the measured signal in a digital domain.
[0050] The low-spurious orthogonal digital signal synthesis technology in the step 3 is a synthesis technology that can generate two digital signals with orthogonal phases and no phase truncation spur.
[0051] Step 4: The multi-segment decimation and filtering is performed on a mixed signal to filter out a high-frequency component in the mixed signal. Decimation can reduce a data rate after the mixing, and can reduce a quantity of operation points required for the power spectrum estimation in a step 8, thus achieving phase noise measurement with millimeter-level frequency resolution.
[0052] Step 5: After a quotient of data output from upper and lower channels (in other words, | and Q channels) is obtained, the four-quadrant arctangent is performed to obtain an initial phase difference sequence.
[0053] Step 6: The phase unwrapping is performed on the initial phase difference sequence to recover a jump of the phase difference sequence at 1.
[0054] Step 7: A frequency offset elimination algorithm is applied to a phase difference sequence obtained after the phase unwrapping to eliminate an impact from a slight frequency difference, such that a low-error phase difference sequence is obtained.
[0055] The phase unwrapping and the frequency offset elimination that are performed for the initial phase difference sequence the steps 6 and 7 work 7906246 together to reduce a measurement error caused by the slight frequency difference.
[0056] Step 8: The power spectrum estimation is performed on a phase difference sequence obtained after the frequency offset elimination to obtain a phase fluctuation spectral density, and then a single-sideband phase noise spectral density of the measured signal is further obtained. A single-sideband phase noise spectral density of each segment in a multi-segment decimation filter can be spliced to obtain a complete phase noise measurement curve.
[0057] To achieve a trade-off between a quantity of Fast Fourier Transform (FFT) points and resolution (higher resolution leads to a larger quantity of FFT points required), a frequency range segmentation method is adopted, and different sampling frequencies are selected for different frequency bands (a higher sampling multiple leads to a lower sampling rate) to achieve the power spectrum estimation for the frequency bands. For example, for an extremely close frequency band (a frequency band close to an origin of a coordinate system) that requires millisecond-level resolution (frequency band | shown in FIG. 5), a decimation factor is high, a sampling rate is low, and the quantity of FFT points is small, making it easy to implement.
[0058] It can be understood that when the multi-segment decimation and filtering is performed in the step 4, due to a plurality of decimation factors, the steps 5to 7 are separately performed under each decimation factor, that is, after the steps 5 to 7, a phase difference sequence obtained after the frequency offset elimination under each decimation factor is obtained. For example, when the decimation factors are 16, 64, 256, and 2048, phase difference sequences that are obtained after the frequency offset elimination and respectively correspond to the decimation factors 16, 64, 256, and 2048 are obtained in the step 7. Accordingly, in the step 8, single-sideband phase noise spectral densities of phase difference sequences obtained after the frequency offset elimination under the decimation factors are spliced.
[0059] For example, when the measured signal is a sine wave, the measured signal
LU506246 sampled by the ADC is as shown in formula (1):
V(n)= (4, + en) xsin(w,n + Ap(n)) (1)
[0060] In the above formula, w, =27f,. A, W, and f, respectively represent an ideal amplitude, angular frequency, and frequency of the signal, and (7) and
Ap(n) respectively represent a random amplitude fluctuation and a phase fluctuation of the signal.
[0061] A principle of the equal-precision frequency measurement is shown in FIG. 3.
Two counters are used to count the measured signal and a standard signal respectively. When a rising edge of a first measured signal is detected after a preset gate is a high level, an actual gate is enabled and the two counters start counting. After a period of time, the preset gate is lowered. When the measured signal is detected to rise, the actual gate is set to a low level, and the counters stop counting. Count values of the standard signal and the measured signal are respectively denoted as N, and N,.
[0062] A principle of the low-spurious orthogonal digital signal synthesis is shown in
FIG. 4. N represents a bit width of a phase accumulator, and the frequency control word K is as shown in formula (2):
AN
K=2"N_/N, (2)
[0063] A reference frequency source uses an internal clock with a frequency of fx.
Therefore, a relationship between generated signal frequency f. andthe fx is as follows:
K
Jo = Feu IN
[0064] The two signals generated by the low-spurious orthogonal digital signal synthesis technology are respectively as shown in formulas (3) and (4).
V,(n)= An sin (wn +P + Ap, (n)) (3)
Vio(n)=4, cos(w,N+ @, + Ap,,(n)) (4)
[0065] Ideally, amplitudes, angular frequencies, initial phases, and random phase fluctuations of the two signals satisfy A, = 42, War = War, Pn=Pn, and
Ap, (n) = Ap,,(n) respectively.
[0066] After the mixing and the decimation and filtering, the high-frequency component in the mixing can be filtered out to obtain signals | and Q as shown in
LU506246 formulas (5) and (6).
I(n)= (4, + e(n)) An cos((w, WI D + M — Ap, (n) + Agp(n)) (5)
O(n) = (A, + e(n)) Ay sin((w, —W2)N— Pia + M = A2) + Ag(n)) (6)
[0067] The division and the four-quadrant arctangent are performed on the O(n) and the /(7) to obtain initial phase difference sequence @(n) between the measured signal and each of the two orthogonal signals, as shown in formula (7).
O(n) = (Wo —w,, )n—@,, +, — An 01) + Ap(n) (7)
[0068] There is a slight frequency difference between each of the two digital signals
V,(n) and V,,(n) generated by a numerically controlled oscillator (NCO) and the measured signal, namely, w, #w,,. In addition, because the four-quadrant arctangent limits the 007) to be within [-7,7], the 9(n) is in a sawtooth shape, which will distort phase noise when the power spectrum estimation is directly performed on the 9(n).
[0069] Therefore, before the frequency offset elimination is performed on the (7), it is necessary to perform the phase unwrapping on the (n). Specific steps are as follows:
[0070] 1) Starting from a second phase difference sequence, a phase difference between current data and previous data is calculated.
[0071] 2) If the phase difference is greater than TT or less than —TT, phase difference data between the point and each piece of subsequent data is added or subtracted by 2T.
[0072] 3) The step 2) is repeated until all data is traversed.
[0073] A simple linear regression model is used to fit a linear trend error after the phase unwrapping.
[0074] A fitted linear error is subtracted from the phase difference sequence after the phase unwrapping to obtain the phase difference sequence obtained after the frequency offset elimination. The power spectrum estimation is performed on the phase difference sequence, and the single-sideband phase noise spectral density is calculated.
[0075] Phase noise of a 500 kHz signal is measured by using the method provided in the present disclosure. In the multi-segment decimation and filtering, a 7906246 four-segment decimation filter with decimation factors of 16, 64, 256, and 2048 is used. A quantity of operation points, frequency resolution, and a splicing range for each segment are shown in Table 1. For each frequency band, data with different sampling frequencies (the sampling multiple is inversely proportional to the sampling frequency, and an original sampling frequency/the decimation factor= current sampling frequency) are selected to achieve the power spectrum estimation. For example, for a frequency band with a frequency range of 10k Hz to 100k Hz offset from the carrier frequency, data with a decimation factor of 16 is selected, and 2’? is used as the quantity of operation points for the power spectrum estimation. Similarly, for a frequency band with a frequency range of 1k
Hz to 10k Hz offset from the carrier frequency, data with a decimation factor of 64 is selected, and 2' is used as the quantity of operation points for the power spectrum estimation.
Table 1 Segmental decimation and splicing range of a 500 kHz signal oo Frequency range Quantity of
Decimation offset from the Frequency Splicing range factor carrier frequency operation resolution [Hz] [Hz] [Hz] points ma mm
[0076]
[0077] FIG. 5 shows a phase noise test result of splicing in this example. A curve labeled "A" shows a data measurement result of the algorithm in the present 7906246 disclosure, and a curve labeled "B" shows a measurement result of a commercial instrument. As shown in FIG. 5, compared with the commercial instrument, the present disclosure can measure phase noise within [0.0006 Hz, 1 Hz], and achieve an extremely close frequency resolution of 0.0006 Hz. FIG. 6 is a schematic diagram of the splicing in the example in FIG. 5 Target phase difference sequences ai, ai, an, and ay, and frequency bands 1, Il, Ill, and IV are included. Corresponding decimation factors of the frequency bands |, II, lll, and IV are 2048, 256, 64, and 16 respectively. For the extremely close frequency band, namely, the frequency band |, to save computational resources, five segments are obtained through segmentation based on the quantity of FFT points, and then phase noise power spectral densities are spliced to obtain a phase noise power spectral density with a frequency range of [0.0006 Hz, 100 Hz] offset from the carrier frequency. Finally, an image obtained by splicing phase noise power spectral densities with a frequency range of [0.0006 Hz,100k Hz] offset from the carrier frequency is obtained.
[0078] An embodiment of the present disclosure provides a device for fully-digital phase noise measurement with millinertz-level measurement frequency resolution, including:
[0079] a first processing module configured to convert a measured signal into a digital signal, and obtain a frequency control word of the digital signal,
[0080] a second processing module configured to generate two digital signals Vas and Vn2 with orthogonal phases and no phase truncation spur based on the frequency control word, and mix the Vhs and the Viz with the digital signal to obtain
Vhs and Vp2 respectively;
[0081] a third processing module configured to: separately perform multi-segment decimation and filtering on the Vm and the Vp2' to obtain a filtered signal of the
Var and a filtered signal of the V,2" under each decimation factor; and under each decimation factor, perform division and four-quadrant arctangent on the filtered signal of the Vhs and the filtered signal of the Vx." to obtain a corresponding initial phase difference sequence, and then perform phase unwrapping and frequency offset elimination on the initial phase difference sequence in turn to obtain a corresponding target phase difference sequence; and
[0082] a fourth processing module configured to perform power spectrum estimation on the target phase difference sequence under each decimation factor to obtain a 7906246 single-sideband phase noise spectral density of the target phase difference sequence, and splice the single-sideband phase noise spectral density of the target phase difference sequence under each decimation factor to obtain a complete phase noise measurement curve.
[0083] An embodiment of the present disclosure provides a system for fully-digital phase noise measurement with millinertz-level measurement frequency resolution, including a computer-readable storage medium and a processor.
[0084] The computer-readable storage medium is configured to store an executable instruction.
[0085] The processor is configured to read the executable instruction stored in the computer-readable storage medium to execute the method described in any one of the above embodiments.
[0086] An embodiment of the present disclosure provides a computer-readable storage medium. The computer-readable storage medium stores a computer instruction, and the computer instruction is configured to enable a processor to perform the method described in any one of the above embodiments.
[0087] It is easy for those skilled in the art to understand that the above-mentioned contents are merely the preferred embodiments of the present disclosure, and are not intended to limit the present disclosure. Any modifications, equivalent substitution, and improvements made within the spirit and principles of the present disclosure should fall within the protection scope of the present disclosure.
Claims (7)
1. A method for fully-digital phase noise measurement with millihertz-level measurement frequency resolution, comprising: S1: converting a measured signal into a digital signal, and obtaining a frequency control word of the digital signal: S2: generating two digital signals Vhs and Vh2 with orthogonal phases and no phase truncation spur based on the frequency control word, and mixing the Vhs and the Vr2 with the digital signal to obtain V»7 and Vaz respectively; S3: separately performing multi-segment decimation and filtering on the V»7 and the Vh2' to obtain a filtered signal of the Vhs and a filtered signal of the Vk2' under each decimation factor; and under each decimation factor, performing division and four-quadrant arctangent on the filtered signal of the V;7° and the filtered signal of the Vh2' to obtain a corresponding initial phase difference sequence, and then performing phase unwrapping and frequency offset elimination on the initial phase difference sequence in turn to obtain a corresponding target phase difference sequence; and S4: performing power spectrum estimation on the target phase difference sequence under each decimation factor to obtain a single-sideband phase noise spectral density of the target phase difference sequence, and splicing the single-sideband phase noise spectral density of the target phase difference sequence under each decimation factor to obtain a complete phase noise measurement curve.
2. The method according to claim 1, wherein in the step S1, the frequency control word of the digital signal is obtained by using an equal-precision frequency measurement method, a direct measurement method, or a periodic measurement method.
3. The method according to claim 1, wherein in the step S2, the Vhs and the Vr2 are generated by using a phase truncation-free frequency synthesis method, a ROM table compression frequency synthesis method, or a coordinate rotation digital computer (CORDIC) frequency synthesis method.
4. The method according to claim 1, wherein the step S4 comprises: segmenting a frequency range offset from the carrier frequency, and selecting different sampling 7506246 frequencies for different frequency bands to achieve the power spectrum estimation for the frequency bands.
5. A device for fully-digital phase noise measurement with millihertz-level measurement frequency resolution, comprising: a first processing module configured to convert a measured signal into a digital signal, and obtain a frequency control word of the digital signal; a second processing module configured to generate two digital signals V»7 and Vaz with orthogonal phases and no phase truncation spur based on the frequency control word, and mix the Vm and the Vr2 with the digital signal to obtain Vhs and Vn2 respectively; a third processing module configured to: separately perform multi-segment decimation and filtering on the Vhs and the V2" to obtain a filtered signal of the Var and a filtered signal of the Vh2 under each decimation factor, and under each decimation factor, perform division and four-quadrant arctangent on the filtered signal of the Vis and the filtered signal of the Vn2 to obtain a corresponding initial phase difference sequence, and then perform phase unwrapping and frequency offset elimination on the initial phase difference sequence in turn to obtain a corresponding target phase difference sequence; and a fourth processing module configured to perform power spectrum estimation on the target phase difference sequence under each decimation factor to obtain a single-sideband phase noise spectral density of the target phase difference sequence, and splice the single-sideband phase noise spectral density of the target phase difference sequence under each decimation factor to obtain a complete phase noise measurement curve.
6. A system for fully-digital phase noise measurement with millihertz-level measurement frequency resolution, comprising a computer-readable storage medium 7506246 and a processor, wherein the computer-readable storage medium is configured to store an executable instruction; and the processor is configured to read the executable instruction stored in the computer-readable storage medium to execute the method according to any one of claims 1 to 4.
7. A computer-readable storage medium, wherein the computer-readable storage medium stores a computer instruction, and the computer instruction is configured to enable a processor to execute the method according to any one of claims 1 to 4.
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