KR880001593B1 - Ac motor speed control system - Google Patents

Abstract

The speed of an induction motor is controlled by an AC voltage based on a speed command signal (VCMD) in the form of an analog voltage. The actual value of the rotational speed (TSA) is fed back to the input side of the control network. The VCMD and TSA signals feed into an adder subtractor to produce an error signal ER indicative of the difference and representing the motor slip. The error signal ER is applied to a proportional integrating circuit, a polarity discriminating circuit and also applied to an absolute value circuit with the output going to a voltage-to-frequency converter producing a pulse train in which the frequency is proportional to the error signal magnitude.

Description

AC motor speed controller

1 to 3 are explanatory diagrams for explaining a synchronous motor control method.

4 is a block diagram of a driving circuit of a synchronous motor.

5 is an explanatory diagram of a resolver.

6 is an explanatory diagram of an output waveform of a resolver.

7 is a circuit diagram of a two-phase three-phase conversion circuit.

8 and 9 are explanatory diagrams of a speed control method.

10 and 11 are explanatory diagrams for explaining the speed detection method of the previous proposal.

12 to 14 are explanatory views of the first embodiment of the present invention, FIG. 12 is a block diagram, FIG. 13 is a time chart, and FIG. 14 is a detection speed explanatory diagram.

15 and 16 are explanatory views for explaining vector control.

Fig. 17 is a block diagram for realizing vector control.

18 and 19 are explanatory views for explaining a conventional speed detection method.

20 to 22 are explanatory views of the second embodiment of the present invention, in which FIG. 20 is a block diagram, FIG. 21 is a time chart, and FIG. 22 is a detection speed explanatory diagram.

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a speed detection device for an AC motor such as a synchronous motor or an induction motor, and more particularly to a speed detection device for an AC motor suitable for use in detecting a speed of an AC motor at low speed rotation.

Rotating field type synchronous motors have the armature as the stator and the magnetic pole as the rotor. The rotating magnetic field is generated through three-phase alternating current through the stator winding (armature winding), and the magnetic field magnetic pole is attracted to the rotating magnetic field. Rotate at the same speed as the electromagnetic field. As a control method of such a synchronous motor, a method has been developed in which torque generation equivalent to that of a conventional motor can be performed by instantaneously controlling a stator current (armature current). 1 to 4 are explanatory views for explaining such a synchronous motor control method.

In general, the torque generating mechanism of the decentralized DC motor performs the current switching operation by the commutator so that the armature current Ia is always orthogonal to the main magnetic flux ψ as illustrated in FIGS. 1a and b. The generated torque T is represented by the following equation. If the main magnetic flux ψ is constant, the torque T is proportional to the armature current I a.

T = K ψ Ia (I)

(Where K is an integer)

1, FM is field stimulation, AM is armature, and AW is armature winding.

However, if the above relation is applied to the rotating field type synchronous motor shown in Fig. 2, ψ can correspond to the main magnetic flux vector ψs of the field magnetic pole PM, and Ia can correspond to the current vector Is of the armature winding SW, respectively. Generated torque T is

T = K 'ψsIs cosT (2)

은 동기 전동기의 등가 회로인 제 3 도를 참조하면 전기자 전류 I s와 유전 기전압 Eo의 위상차이다. 그리고 제 3 도에 있어서 ra는 전기자 권선 SW의 저항이며, Xs은 전기자 반작용 및 전기자 누설 자속을 고려한 동기 리액턴스이다.(Where K 'is an integer). here

Referring to FIG. 3, which is an equivalent circuit of the synchronous motor, is a phase difference between the armature current Is and the dielectric voltage Eo. In Fig. 3, ra is a resistance of the armature winding SW, and Xs is a synchronous reactance in consideration of the armature reaction and the armature leakage magnetic flux.

Therefore, if the phase of the induced electromotive voltage Eo and the armature current Is is in phase, in other words, if the main magnetic flux ψs and the armature current Is are controlled to be orthogonal, the torque T given by the equation (2) is

T = K 'ψsIs = K _T T _n (3)

Thus, the synchronous motor can be driven completely equivalent to the torque generation of the direct current motor. Provided that K _T is a conversion constant.

Therefore, in the proposed method, the phase of the induced electromotive voltage Eo is detected, and a current command having the phase is generated, and this current command is applied to the armature winding of the synchronous motor so that the synchronous motor is equivalent to the DC motor. Drive control. Since the phase of the induced electromotive voltage Eo and the parking speed is delayed by 90 degrees, the phase of the induced electromotive voltage Eo can be detected from the main magnetic flux, in other words, the position of the field magnetic pole. 4 is a circuit block diagram for realizing such a synchronous motor control method.

In the drawing, reference numeral 101 denotes a synchronous motor of a rotating field type, and reference numeral 102 denotes a resolver connected to an axis of the motor and detects the position of the magnetic pole of the synchronous motor. This resolver generates carrier waves of two stator windings 102c, 102d and sin ωt disposed with the rotor 102a and the rotor winding 102b in phase with each other as shown in FIG. Has a carrier generation circuit 102e.

Here, assuming that the rotor 102a is at the position of the angle θ, the voltages shown in the following equations from the stator windings 102c and 102d, respectively,

ea = sinθsin ωt (4)

eb = cosθsin ωt (5)

Is output. That is, the sine and voltage ea and cosine wave voltage eb corresponding to the position θ of the field stimulus of the synchronous motor 101 are output from the resolver 102. Reference numeral 103 denotes a synchronous rectification circuit. The sinusoidal rectification circuit synchronously rectifies the sinusoidal voltage ea and the cosine wave voltage eb to output sinθ and cosθ (Fig. 6). Reference numeral 104a is a quadruple circuit that converts the sinusoidal signals ea and eb into a pulse train and quadruples its frequency. In addition, the quadruple circuit 104a monitors the phases of the sinusoidal signals ea and eb of the two phases, and performs a forward rotation pulse pn with a frequency of four times on line 11 when the motor rotates forward. Outputs the reverse rotation pulse pr of frequency.

Reference numeral 104a denotes a frequency voltage converter (referred to as an F / V converter) for converting the frequency of the forward or reverse pulses Pn and Pr into voltage, and reference numeral 105 denotes a speed command voltage commanded from a speed command circuit (not shown). An operation circuit that calculates the difference between VCMD and the actual speed voltage TSA (hereinafter referred to as speed error) ER. Reference numeral 106 denotes an error amplifier that amplifies the speed error ER and outputs the amplitude Is of the armature current. Reference numerals 107 and 108 As a multiplication circuit, the outputs of the error amplifier and the outputs cos θ and sin θ of the synchronous rectification circuit 103 are raised to output two-phase current commands I 1a (= Is · sinθ) and I 1b (Is · xosθ), respectively.

Reference numeral 109 denotes a two-phase three-phase conversion arc, and has a circuit configuration as shown in FIG. That is, the two-phase three-phase conversion circuit has two operational amplifiers OA1 and OA2, R1 to R4 of 10KΩ, resistor R5 of 5.78K, and resistor R6 above 5KΩ. However, when the values of the resistors R1 to R6 are determined as described above and connected as shown in the figure, the terminals Tu, Tv, and Tw are respectively

Is output. These Iu, Iv, and Iw each have a phase value of 2π / 3, and are each of an induced electromotive voltage Eo and a three-phase current command in phase.

Reference numerals 110V, 110V, and 110W are calculation circuits set for each phase, respectively, and are calculation circuits for calculating the difference between the command currents Iu, Iv, Iw and the phase currents Iau, Iav, Iw of the snow, and 111 is for the addition of Iav and Iaw. Computation circuit that outputs the phase current Iau of the U phase, 112V and 112W are the current transformers for detecting the phase currents Iav and Iaw of the V phase and the W phase, respectively, and 113U, 113V, and 113W are set for each phase to amplify the phase difference between each phase. 114 is a pulse width modulation circuit, 115 is an inverter controlled by an output signal of an output signal of a pulse width modulation circuit, 116 is a three-phase AC power supply, 117 is a rectified three-phase alternating current. As a known rectifier circuit, a diode group 117a and a capacitor 117b are included.

The above is the details of the synchronous control, but in order to accurately control such a synchronous motor, the rotational speed must be detected with high precision at both the low speed and the high speed. In the above description, the resolver is used as the position detector. In this case, a tachogenerator is used as the speed detector. However, this method is not widely used because the cost is higher than that of using a pulse generator as a position detector to perform speed detection therefrom. Therefore, there is a need for a method of performing speed detection with high accuracy from the pulse generator.

However, in the conventional speed detection method, as shown in FIG. 4, the two-phase signals ea and eb of the frequency f which are mutually π / 2 phase difference and are proportional to the rotational speed of the motor are generated from the pulse generator 102. The voltage in proportion to the rotational speed is then converted from the frequency voltage converter 104b, which converts the two-phase signal into a frequency 401 signal through the quadruple circuit 104a, and finally generates a voltage proportional to the frequency 4f. ) Was printed.

However, in this type of industry, when the motor speed becomes low, the output voltage value of the frequency voltage converter is not proportional to the rotation speed, and the precision rapidly decreases. In addition, it is preferable in the LIS to read a microcomputer (hereinafter abbreviated as microcomputer) by using the pulse signal as a digital amount rather than frequency / voltage conversion of the pulse signal generated from the pulse generator 102. Therefore, various methods of reading the speed digitally have been proposed. 8 is a conventional explanatory diagram in which pulses generated from the pulse generator 211 are counted for a predetermined time and outputted to the microcomputer. In this method, the pulse generator 211 is set, and whenever the motor rotates a small amount, the pulse generator 211 generates a pulse and the pulse is counted in the counter 212 so that the contents of the counter are set at predetermined times. The microcomputer 214 transfers the data to the register 213 and resets it, and then reads the contents of the register into the microcomputer 214 at actual speed. Then, the above operation is repeated to take out the actual speed by digital. In this method, however, the speed is detected with high accuracy at high speed, but the pulse period generated from the pulse generator 211 becomes large at low speed, so that the speed cannot be detected with high precision. However, in order to increase the resolution at low speed, it is necessary to increase the number of pulses per revolution generated from the pulse generator 211 or to increase the read period. However, the number of pulses generated from the pulse generator 211 is limited to 10,000 pulses per revolution, and the resolution cannot be improved at this degree, and therefore the resolution intended by the former means cannot be increased.

On the other hand, adopting the latter means of lengthening the read cycle results in poor control response. That is, the read period (sampling period) of the register 213 by the microprocessor takes into account the responsiveness of the speed control system and does not improve the responsiveness to about 1 ms. Then, the sampling period is 1 m sec. The case where the output pulse Pc generated from 211 is set to 10,000 (pulse / rotation) and the motor rotates at an extremely low speed of 1 rpm will be described with reference to FIG.

In FIG. 9, SPi is a sampling pulse, Pc is a pulse generated from the pulse generator 211, and [CN] is a count value of the counter 212. In FIG. Here, the pulse Pc from the pulse generator 211 is counted by the counter 212 as described above, and the contents of the counter are reset at the same time as the satin 213 is synchronized with the sampling pulse SPi, and then The contents of the register are read by the microcomputer 214. However, since the sampling period is 1 msec and the pulse Pc is 6 m sec, the count value [CN] of the counter 212 becomes 1 at the time of the first sampling pulse SPi, but when the subsequent sampling pulses SP2 to SP6 occur, [CN] Is 0. In other words, data intermittent at 1,0,0,0,0,0 is input to the microcomputer as speed data. Therefore, it is not possible to carry out fast and accurate speed control. Therefore, the present inventor has proposed a speed detection method (Japanese Patent Application No. 56-74677) in which speed detection can be performed accurately without speed data being interrupted. Next, the method of this proposal is explained in full detail.

In the method of the present proposal, the period T of the pulse Pc generated from the pulse generator at the time of continuous rotation is counted, and the period T is used to detect the row speed.

Here, if the number of clock pulses CP generated in the period of the pulse Pc is N and the period of the clock pulse is (T (= 0.125 μs), the frequency f of the pulse Pc generated from the pulse generator is

f = 1 / T

= 1 / N∇∇ (Hz / μ sec)

= 106 / N ・ ∇T (Hz / sec)

= 60 x 106 / N / T (Hz / min)

If you substitute T = 0.125

f = 480 × 106 / N (Hz / min) (7)

Becomes If the pulse generator makes 10,000 pulses per revolution, the rotation speed n is

n = 48,000 / N (rpm) (8)

Becomes Therefore, in the proposed scheme, the period T of the pulse Pc generated from the pulse generator at low speed, in other words, the number N of clockples CP generated within one period of the pulse Pc, is rotated from the equation (7) or (8). Speed is being detected.

10 is a block diagram illustrating this approach.

In the figure, reference numeral 301 denotes a pulse generator which outputs two pulse signals Pc and Pc ', each of which is delayed by π / 2 phase, each time the motor (not shown) rotates a predetermined amount, and 302 denotes a rotation direction discrimination cycle. Which phase of the signals Pc and Pc 'is advanced is output, and the rotation direction signal SGN is output. True signal 303 is a counter and has a clear terminal CLR, a count enable terminal EN, a clock terminal CLK, and a carry pulse generation terminal TC. The pulse signal Pc is input to the clear terminal CLR, and "1" is always input to the counter enable terminal EN.

Therefore, the counter 303 clears the count value every time the pulse signal Pc occurs, and counts the number of clock pulses CP until the next pulse Pc occurs.

Reference numeral 304 is a flip flop which is set whenever a carry pulse (place pulse) OFP occurs and is reset whenever a pulse signal Pc occurs. That is, the counter 303 and the flip flop 304 count the number of clock pulses generated within one period of the pulse signal Pc. And when the capacity of the counter 303 is large enough, it is not necessary to set the flip flop 304. Reference numeral 305 denotes an AND gate, 306 denotes an n-bit register, and each time the output of the AND gate 305 becomes "1". In other words, the count value of the counter 303 is set whenever the pulse signal Pc occurs. Reference numeral 307 denotes a 2-bit register. When the output of the AND gate 305 becomes " 1 ", the set output of the flip-flop 304 and the forward direction signal SGN (" 1 " In the case of 0 ", reverse rotation) is set.

Reference numeral 308 denotes a division unit, which calculates the rotational speed n by inputting the contents of the registers 306 and 307 to perform the calculation (8). In addition, the division unit 308 can be configured using the microcomputer division function for digitally controlling the speed control. Particularly in recent microcomputers, a microcomputer capable of performing 16-bit division at 10 μs or less has emerged. Thus, the use of such a microcomputer makes it possible to easily configure a division unit. In FIG. 10, each time a pulse signal Pc is generated from the pulse generator 301, the counter 303 and the flip flop 304 are reset, and then the counter 303 and the flip flop 304 are next pulsed. The number N of clock pulse CPs is counted until the signal Pc occurs.

Thereby, the period of the pulse signal Pc is measured.

When the next pulse signal Pc is generated, the contents of the counter 303 and the flip flop 304 and the rotation direction signal SGN are set in the registers 306 and 307. At the same time, the counter 303 and the flip flop 304 are reset again to start counting the clock pulse CP. By the way, the number N of clock pulses CP stored in the registers 306 and 307 is properly read by the division unit 308. The division unit of formula (8) is executed by the division unit to obtain the rotation speed n. .

Thus, according to the method shown in FIG. 10, as shown in FIG. 11, even if the rotation speed n is very low speed, a detection value is not intermittent, and it can detect a high accuracy speed compared with the conventional method.

However, in the scheme of this proposal, the detection speed becomes stepped. In other words, in one method, the actual speed is annualized every time a pulse is generated from the pulse generator, and the detection speed is increased to be equal to the actual speed, and the detected value is maintained until the next pulse occurs.

Therefore, there is no problem when the speed is constant or when it is changing very slowly, but when the speed is changing relatively rapidly, the step width is increased so that the exact speed is not shown between the pulses. This means that a delay element is introduced into the speed detector, and this does not allow the gain of the speed control loop to be held high so that rapid control cannot be realized.

This problem also arises not only in the speed control system of a synchronous motor, but also in the speed control system of an induction conductor.

It is a first object of the present invention to provide a novel speed detection device capable of detecting the actual speed of an AC motor with a higher accuracy than before.

It is a second object of the present invention to provide a novel speed detecting apparatus capable of detecting the actual speed of an AC synchronous electric road with a higher accuracy than before.

It is a third object of the present invention to provide a novel speed detection device capable of detecting the actual speed of an AC induction motor with a higher accuracy than before.

Other objects are clarified by the following detailed description.

In the present invention, the speed is calculated by the method shown in FIG. 10 when a pulse pulse is generated by the pulse generator attached to the shaft of the synchronous motor. Whenever the enable signal occurs for a certain period between pulses generated by the pulse generator, the actual actual speed is predicted using the current current value and the past detection speed according to the characteristics of the synchronous motor. Speed is regarded as detection speed.

Next, a description will be given of the calculation of the actual speed.

When the torque T of the synchronous motor is regarded as the motor inertia of the motor, the inertia of the Jm (kgcmsec2) load is set to the JL [kgcmsec2] rotational speed as the vload torque To [kgcmcm].

+To …………………………………(9)T = (J _M + J _L )

+ To… … … … … … … … … … … … … (9)

From (3), (9)

+To ……………………………(10)KTIS = (J _M + J _L )

+ To… … … … … … … … … … … 10

This holds true. If we transform (10)

Becomes

dv = v (k + 1) = v (k)

This holds true. The equation (11) is corrected by the discrete scale of the sampling period Tsm`.

This holds true. However, it is also possible to assume that the load torque To does not change between samplings, so that it can be regarded as ▷ To = 0, so that Equation (12)

Becomes And if we transform (13) again

Becomes

Therefore, if the detection speed v (k-1) before 1 sampling and the current command value I _S (k-2) before 2 sampling are known, a rotation speed can be estimated from Formula (14).

FIG. 12 is a block diagram of an embodiment of the present invention, FIG. 13 is the same time chart, and FIG. 14 is an explanatory diagram of detection speed. In addition, the same code | symbol is attached | subjected to the same part as the conventional apparatus of FIG.

In the figure, reference numeral 401 has a configuration shown in FIG. 10 (except for the pulse generator 301 and the division unit 308) as a speed detector. Reference numeral 402 is set by the pulse Pa generated from the pulse generator 102 as a flip flop and reset by the speed calculation end signal OPEN generated from the judging device 113.

Reference numeral 403 is a speed calculation of the predetermined period interrupt signal (speed operation enable signal) when an interrupt pulse ITP of a predetermined period generator for generating a ITP from the interrupt pulse generating circuit 403 (the start t _2), processing device (113 First detects whether flip flop (FF) 420 is set. Since the FF 402 is set at this time, the processing apparatus 113 performs division operation of Formula (8) using the period AV of the pulse Pa input from the speed detection part 401, and calculate | requires a rotation speed. After completion of the calculation (time t ₃ ), the processing apparatus outputs the speed calculation end signal OPEN to reset the FF 402 to the initial state. Then, the processing apparatus 113 calculates a speed deviation, and outputs the primary current gun signal I _{S according} to the speed deviation. Thereby, the primary current is commanded to the synchronous motor by the function of the subsequent circuit.

The above is the case where the FF 402 is set, but when the FF 402 is set, the prediction operation of the rotational speed is executed. In other words, when the enable pulse ITP is issued at time t ₄ , the processing apparatus 113 detects the set state of the FF 402. If the FF 402 is not set, the processing apparatus 113 uses the past detection speed v (k-1) and the past current command I _S (k-2) stored in the data memory 113c ( 14) is executed to estimate the rotational speed v (k). Thus, the difference between the command speed VD is obtained by using the rotational speed v (k) as the actual speed, and the primary current command is output as described above.

Thereafter, the FF 402 is not set also at the times t ₅ and t ₆ , so that the processing apparatus 113 executes the calculation (14) and predicts the actual speed. At the time t ₇ , pulse Pa is generated from the pulse generator 102 so that the FF 402 is set, and the rotational speed processing using the pulse period AV as described above is executed.

As a result, if the actual speed Va of the synchronous motor is converted as shown in Fig. 14 (a), the rotation speed detected by the present invention is changed as shown by the solid line of Fig. 14 (b). On the contrary, the stem width is much smaller, so that high-speed detection is possible.

As a driving method of an induction motor, a vector control method having a torque generating mechanism equivalent to a DC motor is known. 15 and 16 are explanatory views for explaining such a vector control method. FIG. 15 is an equivalent circuit of an induction motor in vector control, and FIG. 16 is an excitation current Io and a secondary current I _{2 in} vector control. 1 m is excitation reactance, r ₂ is equivalent resistance, and S is slip. Considering the equivalent circuit of such an induction motor, the generated torque T is

Becomes Where ωs is the slip angle frequency. If I ₂ and S · ωs are proportional to each other, the torque T will have a torque generating mechanism such as a direct current motor in proportion to the secondary current.

Since this holds true, the excitation current Io must be made constant in order to proportion I ₂ to S · ωs.

From the above, as shown in FIG. 16, the vector control changes the excitation current Io and the secondary current I ₂ while maintaining the excitation current Io constant and only the secondary current I ₂ in proportion to the load torque. Control method. In actual vector control, the deviation (speed deviation) ER between the command speed and the actual speed is regarded as the torque command, and the primary current I1 is determined according to the speed deviation ER.

I ₁ = Io + j I ₂

= Io + jk ER (17)

Is determined to satisfy. Where k is the ratio

17 is a block diagram for realizing vector control. In the figure, reference numeral 1101 denotes a three-phase induction motor, and 1102 denotes a pulse generator such as a resolver, and generates two sinusoidal wave signals Pa and Pb delayed by 90 ° in proportion to the rotational speed. Reference numeral 1105 denotes a quadruple circuit that converts the sinusoidal signals Pa and Pb into pulse trains and quadruples the frequency.

In addition, the quadruple circuit 1105 monitors the phases of the two-phase sine wave signals Pa and Pb, outputs the rotation direction signal RDS, and forwards the forward rotation pulse Pn of four times the frequency to the line 1 ₁ , When reverse rotation, the reverse rotation pulse Pr of 4 times the frequency is output to the line 12, respectively.

Reference numeral 1106 denotes a frequency voltage converter (referred to as an F / V converter) for converting the frequency of the forward or reverse pulses Pn and Pr into voltage, and 1108 denotes a speed command voltage VCMD and a real speed commanded from a speed command circuit (not shown). An arithmetic circuit that calculates the difference ε R (hereinafter referred to as speed deviation) of the voltage TSA, 1109 is a proportional integration circuit that proportionally integrates the speed deviation ε γ, and 1110 is an absolute value circuit that absoluteizes the speed deviation ε γ, 1112 is a voltage frequency converter (hereinafter referred to as a V / F converter), and outputs a pulse string Pe of a frequency multiplied by the size of the ER.

Reference numeral 1113 denotes a microcomputer, which has a processing unit 1113a, a control program memory 1113b, and a data memory 1113c. The data memory 1113c digitally stores the torque large amplitude characteristic (TI ₁ characteristic), the horn full-band sinusoidal characteristic (sin pattern), and the like as a function table.

The processing unit 1113a counts the pulse string Pe generated from the V / F converter 1112 for a predetermined time under the control of the control program, and outputs the digital current amplitude I ₁ using the count value N and the TI ₁ characteristic. In other words, the count value N is regarded as a torque command, and I ₁ is obtained from the TI ₁ characteristic and output. And this current command I ₁ becomes the amplitude of the primary current shown by Formula (17). In addition, the processor 1113a uses a pulse train Pn or Pr having a frequency ωn proportional to the rotational speed of the induction motor 1101, a constant phase difference ψ, or the like.

sin (ωnt + ωst + ψ) (18)

sin (ωnt + ωst + ψ + 2π / 3) (19)

Outputs by digital. Ωs is the slip angular frequency, and ψ is the positional phase. Reference numerals 1116 and 1117 are odds meetings.

I1sin (ωnt + ωst + ψ) (20)

I1sin (ωnt + ωst + ψ + 2π / 3) (21)

Is a QA converter that converts it to analog and outputs the analog current commands iu and iv of the U and V phases.

iu + iv ──── iw (22)

The calculation circuit for outputting the current command iw of the W phase by performing the addition calculation of

iua + iva ─────iwa (23)

Is a calculation circuit that outputs the phase current iwa flowing through the W phase by performing the addition operation, and 1122U, 1122V, and 1122W are installed in each phase, respectively, and the current difference (iu-iua), (iv-iva), and (iw-iwa). Is a current suppression circuit that calculates and amplifies, and 1123 is a pulse width modulation circuit, and has three pulse width modulation circuits (1123U), (1123V), and (1123W) respectively provided for each phase, and pulse width modulation of each of the current differences. do.

Reference numeral 1124 denotes an inverter circuit as a transistor, and 1125 denotes a rectifier for converting three-phase alternating current into direct current.

However, in the scheme of the above-described proposal, the period T of the pulse Pe generated from the pulse generator at the low speed rotation is displayed, and the speed detection is performed using the period T. When the number is N, the period of the pulse pulse is / T (= 0.125 μs), the frequency f of the pulse Pc generated from the pulse generator is

If you substitute T = 0125

f = 480 x 10 ⁶ / N (Hz / min) (24). If the pulse generator generates 10,000 pulses per revolution, the rotation speed n

n = 48,000 / N (rpm) (25)

Becomes Therefore, in the proposal, the period T of the pulse Pc from the pulse generator at low speed, in other words, the number N of clock pulses CP generated in one period of the pulse Pc is obtained, and the rotational speed is detected from the equation (24) or (25). Doing.

18 is a block diagram illustrating this approach.

In the figure, reference numeral 1301 denotes a pulse generator for outputting two pulse signals Pc and Pc 'delayed by π / 2 phase every time a motor (not shown) is diffracted by a predetermined amount, and 1302 is a rotation direction discrimination circuit. It determines which phase is ahead of Pc 'and outputs the rotation direction signal SNG.

Reference numeral 1303 denotes a counter, which has a clear terminal CLR, a counter enable terminal EN, a clock terminal CLK, a carry pulse generation terminal, and a TC. The pulse signal Pc is input to the clear terminal CLR, and "1" is always input to the count enable terminal EN. Therefore, the counter 1303 clears the count value every time the pulse signal Pc occurs, and counts the number of clock pulses Pc until the next pulse Pc occurs. Reference numeral 13.4 denotes a flip flop, which is set whenever a carry pulse (Placement Pulse) OFP occurs, and is reset whenever a pulse signal Pc occurs. In other words, the counter 1303 and the flip flop 1304 indicate the number of clock pulses generated in one cycle of the pulse signal Pc. If the capacity of the counter 1303 is sufficiently large, it is not necessary to set the flip flop 1304. Reference numeral 1305 denotes an AND gate, 1306 denotes an n-bit register, and each time the output of the AND gate 1305 becomes "1". In other words, the count value of the counter 1303 is set whenever the pulse signal Pc occurs.

Reference numeral 1307 denotes a two-bit register. Whenever the output of the AND gate 1305 becomes "1", the set output of the flip-flop 1304 and the rotation direction signal SGN ("1" are "0" during forward rotation. In reverse rotation) is set. Reference numeral 1308 denotes a division unit. The rotational speed N is calculated by inputting the contents of the registers 1306 and 1307 and performing the calculation (25). In addition, the division unit 1308 can be configured using the division function of Micom which digitally processes speed control.

In particular, in recent microcomputers, a microcomputer capable of dividing 16 bits to 10 microseconds or less has emerged. Thus, using such a microcomputer, the division unit can be easily configured. In FIG. 18, whenever the pulse signal Pc is generated from the pulse generator 1301, the counter 1303 and the flip-flop 1304 are reset, and then the counter 1303 and the flip-flop 1304 are next pulse signals. The number N of clock pulses CP is counted until Pc occurs. Thereby, the periodic yarn measurement of the pulse signal Pc is measured.

When the next pulse signal Pc is generated, the contents of the counter 1303 and the flip flop 1304 and the rotation direction signal SGN are set in the registers 1306 and 1307. At the same time, the counter 1303 and the flip flop 1304 are reset again to start counting the clock pulse CP. However, the number N of clock pulses CP stored in the registers 1306 and 1307 is appropriately read by the division unit 1308, and the division unit of formula (25) is executed by the division unit to obtain the rotation speed n. . Thus, according to the method shown in FIG. 18, as shown in FIG. 19, even if the rotation speed N is extremely low speed, the detection speed does not become intermittent, but the conventional method of counting the number of pulses emitted from the pulse register connected to the axis of the induction motor Compared with this, speed detection can be performed with high accuracy. However, in the present invention, when a pulse is generated from the pulse generator, the speed is calculated by the method of the proposed method shown in FIG. Whenever an interrupt signal (enable signal) occurs for a certain period between pulses generated from the pulse generator and the pulse, the current is set by using the current current value and the past detection degree according to the characteristic expression of the induction motor. It estimates the actual speed of and considers the predicted speed as a detection speed.

Next, the calculation of the real speed prediction will be described.

In the formula (15), when 3r ₂ I ₂ / S · ωs = K _T (K _T is constant and the conversion constant), the generated torque T is

T = KTI2 (26)

Can be expressed as On the other hand, if the generated torque T is the motor inertia J _M (kgcm sce2), the load inertia is JL (kg cm sce2), the rotational speed is V, and the load torque is To (kg cm)

+To (27)T = (J _M + J _L ) +

+ To (27)

From (26), (27)

+To (28)K _T I _S = (J _M + J _L ) +

+ To (28)

This holds true. If we transform (28)

Becomes dv = v (k + 1) -v (k)

This holds true. (29) is solved by the discrete value of sampling period Ts

This holds true. However, it is safe to assume that the load torque To does not change between samplings, so ∇ To = 0

Becomes And if you transform (30) again

Becomes

Therefore, if the detection speed v (k-1) before 1 sampling and the current command value I ₂ (k-2) before 2 sampling are known, the rotation speed can be estimated from the equation (31).

FIG. 20 is a block diagram of an embodiment of the present invention, FIG. 21 is the same time chart, and FIG. 22 is an explanatory diagram of detection speed. The same reference numerals are given to the same parts as in the conventional apparatus of FIG.

In the figure, reference numeral 1401 has a configuration shown in FIG. 18 (except for the pulse generator 1301 and the division unit 1308) as the speed detector. Reference numeral 1402 denotes a flip flop, reset by the pulse Pa generated from the pulse generator 1102, and reset by the speed calculation end signal OPEN generated from the processing apparatus 1113.

Reference numeral 1403 denotes an interrupt pulse generation circuit that generates a constant period operation interrupt signal (speed operation enable signal) ITP. VD is a digital command speed, and AV is a digital value output from the registers 1306 and 1307 shown in FIG.

Next, the operation of speed detection will be described.

Flip pulse 1402 is set (RGS = " 1 ") when pulse Pa occurs at start t ₁ (FIG. 21) from pulse generator 1102. FIG. In this state, when an interrupt pulse ITP of a predetermined period is generated from the interrupt pulse generating circuit 1403 (time t ₂ ), the processing unit 1113 first detects whether the flip flop (FF) 1402 is set. .

At this time, since the FF 1402 is set, the processing unit 1113 is input from the speed detection unit 1401 to perform a division operation of the equation (25) using the period AN of the pulse Pa to obtain the rotational speed. After the completion of the operation (time t ₃ ), the processing apparatus outputs the speed calculation end signal OPEN and resets the FF 1402 to the initial state. Thereafter, the processing apparatus 1113 obtains the speed deviation, and outputs the phase signals having the phases shown in the primary current amplitude signals I ₁ , (20) and (21) according to the speed deviation, respectively. Thereby, the primary current is commanded to the induction motor by the function of the subsequent circuit.

The above is the case where the FF 1402 is set. However, when the FF 1402 is set, the prediction operation of the rotational speed is executed.

That is, when an interrupt pulse ITP occurs at time t ₄ , the processing unit 1113 senses the set state of the FF 1402. If the FF 1402 is not set, the processing unit 1113 uses the past detection speed v (k-1) and past current command I ₂ (k-2) stored in the data memory 113c ( 31) Calculate the speed v (k) by performing the calculation. Thus, the difference between the command speed VD is obtained by using this rotation speed v (k) as the actual speed, and the primary command is output as described above.

Subsequently, since the FF 1402 is not set also at the times t ₅ and t ₆ , the processing apparatus 1113 performs calculation of the equation (31) to predict the actual speed. When time t _{7 is} rolled, pulse Pa is generated from the pulse generator 1102, the FF 1402 is set, and the rotational speed calculation process using the pulse period AV as described above is executed.

As a result, if the actual speed va of the induction motor is changed as shown in Fig. 22A, the rotation speed detected by the present invention is changed as the solid line of Fig. 22B, and is represented by the dotted line. Compared with the proposed one, the stem width is much smaller, which enables highly accurate detection.

In addition, although the case where the present invention is applied to the speed detection of the induction motor controlled by the vector control has been described, the present invention can also be applied to the case where it is driven by another control method. In addition, although the current command value of the past was used in the calculation of (31), it may be the actual current value of the past.

As described above, according to the present invention, the rotational speed can be detected with high accuracy and the individual can be held high, thereby enabling the induction motor to be excellent in quick response.

In addition, this invention is not limited to a present Example and can be variously modified in the range which does not exceed the summary of a Claim.

Claims (7)

In a speed detection device of an AC motor driven by a current command outputted at a constant cycle due to a speed deviation between a speed command and a real figure, pulses for generating a number of pulses proportional to the rotation amount of the AC motor or the movement amount of the machine moving part. And a generator for counting the period of the pulse and a speed calculating means for calculating the actual speed of the AC motor whenever an enable signal of a certain period is generated. The speed calculating means generates a pulse from the pulse generator. In this case, the actual speed is calculated using the above period in synchronization with the enable signal, and in other cases, the current stall speed is generated using the current output value or the actual current value and the previous actual speed in synchronization with the enable signal. An apparatus for detecting a speed of an alternating-current motor, comprising predicting and calculating degrees. The speed detection device of an AC motor according to claim 1, wherein a synchronous motor is used as the AC motor. The speed control apparatus for an AC motor according to claim 1, wherein an induction motor is used as the AC motor. In the speed detection device of an AC motor driven by a current command outputted at regular intervals due to a speed deviation between a speed command and a real speed, a pulse for generating a number of pulses proportional to the rotation amount of the AC motor or the movement amount of the moving part of the machine. And a generator for counting the period of the pulse and a speed calculating means for calculating the actual speed of the AC motor whenever an enable signal of a certain period is generated. The speed calculating means generates a pulse from the pulse generator. In this case, the real speed is calculated using the period in synchronization with the enable signal, and in the interval between the pulse-a pulses emitted from the pulse generator and the plurality of enable signals generated from the interrupt pulse generation circuit, respectively. Use current output current setpoint or actual current value and past real speed The detecting device of the alternating current motor, characterized in that the prediction operation over a substantial degree of the current. And a plurality of enable signals generated from the interrupt pulse generating circuit are equally spaced. 5. The speed detection device of an AC motor according to claim 4, wherein a synchronous motor is used as the AC motor. 5. An apparatus for controlling the speed of an AC motor according to claim 4, wherein an induction motor is used as the AC motor.

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