JP2003274688A - Method and apparatus for controlling inverter - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To reduce noise and vibration that are generated during operation of a commutatorless motor when achieving rotation control of the commutatorless motor by adjusting inverter output, and to achieve a low-cost inverter controlling apparatus having a simple structure. <P>SOLUTION: The inverter controlling apparatus 1 generates a switching signal in an inverter 2 for driving the commutatorless motor M from an armature current. The inverter controlling apparatus 1 has a retention means 11 for retaining the phase of the switching signal when the armature current is measured, an estimation means 12 for estimating the rotor position of the commutatorless motor from the armature current, an operation means 13 for generating a phase correction value according to the retained phase of the switching signal and the estimated phase of the rotor position, and an output means 14 for continuously outputting a signal string of the switching signal that is being outputted until the phase correction value is generated, and correcting the phase of the switching signal that is being outputted when the phase correction value is generated to output the signal string of a new switching signal. <P>COPYRIGHT: (C)2003,JPO

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an inverter control device for an inverter that controls rotation of a commutatorless motor.

[0002]

2. Description of the Related Art In recent years, various types of motors have been realized. Among them, the non-commutator motor is a motor whose rotation speed is controlled by specifying the position of the rotor of the synchronous motor and controlling the switching timing of the inverter. Therefore, its basic characteristic is similar to that of a DC motor, and the rotation speed can be controlled by adjusting the armature voltage.

Brushless D, which is a kind of commutatorless motor
The C motor is an ordinary DC motor in which mechanical contacts such as brushes and commutators are removed and electronically replaced. That is, the rotor is provided with a permanent magnet, and each stator is provided with an armature winding.
Since the rotation of the rotor is controlled by sequentially switching each energization state of the power supplied to the armature winding of the stator with an electronic switch in a timely manner, it is important how to detect the rotor position information. Come on.

In controlling a brushless DC motor, it is general to provide a sensor for detecting the position of the rotor and switch the energized state of the armature using the sensor output as a criterion.

However, depending on the environment in which it is used,
It may be difficult to attach a sensor that detects the position of the rotor. In such a case, a sensorless arithmetic control called a 120-degree conduction method of driving a motor with a rectangular wave current or a 180-degree conduction method (continuous conduction method) of driving a motor with a sine wave current is used.

FIG. 4 is a diagram for explaining a 120-degree energization method according to a conventional example in the rotation control of a brushless DC motor. FIG. 4A is a schematic view of armature windings provided on the stator of the brushless DC motor. 4B is a wiring diagram, and FIG. 6B is a timing chart of the output current waveform of each phase.

According to the conventional 120-degree energization method, the current flowing into the armature winding of each phase is 120 during rotation of the motor.
It is sequentially switched to “high (high current)”, “low (low current)” and “off (non-energized)” at a commutation timing of a cycle. In the example shown in FIG. 4A, the brushless DC motor has U, V, and W three-phase armature windings.
As shown in, the energization state of the armature winding of each phase is "high", "low", or "off" in any cycle.

For example, when the phase is between 0 and 120 degrees, the U phase is high, the V phase is low, and the W phase is off. At this time, the drive current flows from the U-phase armature winding through the neutral point N into the V-phase armature winding. On the other hand, in the W-phase armature winding in which no current flows, that is, in the non-energized state, the counter electromotive force is induced by the change of the interlinkage magnetic flux due to the rotation of the rotor including the permanent magnet. The position of the rotor can be estimated by detecting this back electromotive force. The commutation timing is measured based on the position information of the rotor thus obtained, and the electronic switch that switches the energization state of each phase armature winding is switched.

FIG. 5 is a timing chart of the output current waveform of each phase in the conventional 180-degree energization method in the rotation control of the brushless DC motor.

In the 180-degree energization method, as shown in FIG. 5, the sine wave current, which is 120 degrees out of phase with each other in the armature winding, is used as a drive current to rotate the rotor. The position of the rotor can be estimated by measuring a current flowing in each armature winding, a voltage generated in each armature winding, and substituting this into a predetermined arithmetic expression. The obtained rotor position information is used as one of the parameters for generating the switching signal of the PWM inverter that supplies the drive current to the armature winding.
There are several methods of calculation for estimating the rotor position information.

[0011]

In the rotation control by the above 120-degree energization method, the energization state of the armature winding of each phase is switched every 120 degrees by the phase angle, and the driving current flowing through the armature winding is It becomes a square wave. The steep edges of the square wave cause electrical and mechanical noise and vibration in the commutatorless motor. Further, since the non-energization period is provided in one of the armature windings, the efficiency is low.

On the other hand, the rotation control by the 180-degree energization method is
Since the motor is driven by the sine wave current output from the PWM inverter, there is little electrical or mechanical noise and vibration, and it is ideal in terms of efficiency. However, a large amount of arithmetic processing is required to calculate the position of the rotor, and it is essential to install a high-performance arithmetic processing device in order to cope with such arithmetic. This can be an obstacle, for example, in introducing the product in the popular price range.

Therefore, an object of the present invention is to solve the above problems.
An object of the present invention is to provide an inverter control device having a simple structure and a low cost, which reduces noise and vibration generated during operation of a non-commutator motor when the rotation control of the non-commutator motor is realized by adjusting the inverter output. .

[0014]

In order to achieve the above object, according to the present invention, a switching signal for controlling switching of an inverter so as to drive a non-commutator electric motor in a sine wave is used as an armature current of the non-commutator electric motor. In the inverter control method to generate using, a holding step of holding the phase of the switching signal at the time of measuring the armature current, an estimating step of estimating the rotor position of the non-rectifier motor from the armature current, The step of calculating the error between the phase of the held switching signal and the estimated phase of the rotor position to generate the phase correction value, and the phase of the switching signal currently being output until the phase correction value is generated. The signal sequence is continuously output, and when the phase correction value is generated, the phase of the switching signal currently being output is calculated based on the phase correction value. And an output step of outputting a signal sequence of the new switching signals correctly.

FIG. 1 is a functional block diagram of an inverter control device according to the present invention.

The inverter control device 1 for generating a switching signal for controlling the switching of the inverter 2 so as to drive the non-rectifier motor M by a sine wave, using the armature current of the non-rectifier motor M, measures the armature current. Holding means 11 for holding the phase of the switching signal at that time, estimating means 12 for estimating the rotor position of the non-commutator motor from the armature current, the phase of the held switching signal, and the estimated rotor A calculation unit 13 that calculates an error from the phase of the position to generate a phase correction value, and a signal sequence of the switching signal that is currently being output is continuously output until the phase correction value is generated. When generated, the output means 14 corrects the phase of the switching signal currently being output based on the phase correction value and outputs a signal sequence of a new switching signal. , Comprising a. The estimating means 12 estimates the phase of the rotor position from the armature current using the equation described later.

According to the present invention, since the commutatorless electric motor is driven by a sine wave, electrical or mechanical noise and vibration are smaller and more efficient than the rectangular wave driving, and a microcomputer for executing arithmetic processing is also used. Alternatively, since control is possible according to the performance of the DSP, it is possible to easily drive the commutatorless motor by a low-performance inexpensive microcomputer or a computing unit such as a DSP.

[0018]

BEST MODE FOR CARRYING OUT THE INVENTION In describing an inverter control device according to the present invention, first, an observer (hereinafter referred to as an observer) used for estimating a rotor position of a brushless DC motor will be described.
It is called "extended magnetic flux observer". ) Will be described.
(Extended magnetic flux observer theory: Proceedings of the 2000 IEEJ Industrial Application Division Conference (IV-p981 to p984), Hanamoto et al., Published August 9, 2000) Fig. 2 is an analysis model for brushless DC motors. FIG.

In this figure, d and q are rotor angular velocities ω
dq coordinate axes rotated by _e , α and β are stationary αβ coordinate axes,
u, v, and w are three-phase coordinates, and θ _e is an electrical angle from the α axis (u axis). Also, v is the armature voltage, i is the armature current, and R
Is an armature resistance, L is an armature inductance, p is a differential operator (= d / dt), subscripts _{d 1} and _q are d-axis components and q-axis components, and subscripts _{α 1} and _{β 2} are α-axis components and β, respectively.
The axis component is shown.

The voltage equation and the number of magnetic flux linkages on the dq coordinate axes of the brushless DC motor are as shown in equation (1).

[0021]

[Equation 7]

When vector control is realized by decoupling the d-axis component and the q-axis component in the speed control system of the brushless DC motor, the equation (1) is converted into the αβ coordinate and the equation (2) is obtained. It is expressed as. Here, ψ _α and ψ _β are αβ axis magnetic flux linkage numbers.

[0023]

[Equation 8]

The flux linkage numbers ψ _α and ψ _β in the equation (2) are expressed as in the equation (3).

[0025]

[Equation 9]

[0026] However, Ψ ₌ a _{(L d -L q) i d} .

Equation (4) is obtained by differentiating equation (3) assuming that the time change of Ψ is gentler than the control period and can be assumed to be a constant.

[0028]

[Equation 10]

Here, e _α = −Ψω _e sin θ _e , e _β
= Ψω _e cos θ _e , equation (2) becomes equation (5)
It is expressed as.

[0030]

[Equation 11]

Hereinafter, only the α phase will be described, and the β phase will be the same, so the description thereof will be omitted. Here, equation (6) is defined by modeling an equivalent induced voltage including a harmonic component and a DC component of a brushless DC motor rotating at a constant electric angular velocity ω ₁ .

[0032]

[Equation 12]

Rewriting the equation (6) into the fundamental wave component and the other components, that is, the sum of the direct current component e _αdc and the harmonic component, the equation (7) is obtained.

[0034]

[Equation 13]

E_αd, The harmonic components have small amplitude
Since it can be ignored, the DC component e _αdcOnly consider
, D (e_αd) / Dt = 0. Also, a₁Oh
And b₁Is a constant value. Here, from the formula (3), the formula
(8) is defined.

[0036]

[Equation 14]

From this,

[0038]

[Equation 15]

[0039]

[Equation 16]

Therefore, equations (2), (9) and (1
From 0), taking the α axis as an example,

[0041]

[Equation 17]

Is obtained.

In the actual calculation of the equation (11), i _α
Is the measured value of the armature current. Further, it is preferable to use the measured armature voltage value for v _α as well, but a value that can be mathematically derived from the above armature current measurement value by a microcomputer may be used. Further, the command value of the angular velocity is substituted for the rotor angular velocity ω _e in the equation (11).

The above also applies to the β phase. Expression (1
If the same dimension observer is applied to 1), the flux linkage number ψ
It is possible to estimate _α ′ and the fundamental wave component e _α1 and the direct current component e _αdc of the equivalent induced voltage. Since part of the system matrix and the observer gain change depending on ω _e , these values are calculated in advance as a function of the speed command value and used.

Using the estimated flux linkage numbers ψ _α 'and ψ _β ', the position of the rotor can be estimated from the following equations (12) and (13).

[0046]

[Equation 18]

[0047]

[Formula 19]

When controlling the rotational speed of a motor driven by an inverter, it is preferable to execute feedback control as often as possible. However, the above-mentioned arithmetic processing takes a very long time, and the feedback control cannot be applied during the period from the measurement of the armature current of the motor to the completion of the estimation of the rotor position. A microcomputer capable of high-speed processing may be used to shorten the processing time of this calculation, but this leads to an increase in cost.

Therefore, in the inverter control device according to the present invention, the signal sequence of the switching signal currently being output is continuously output during the arithmetic processing. On the other hand, after the arithmetic processing is completed, feedback control is performed to correct the phase of the switching signal that is currently being output and output a new switching signal sequence. As described above, according to the present invention, the control cycle is determined according to the processing capability of the microcomputer, so that even a microcomputer having a low-speed processing capability can drive the brushless DC motor in a sine wave by the 180-degree energization method. .

FIG. 3 is a block diagram of an inverter control device according to an embodiment of the present invention.

Generally, a motor drive device using an inverter control system includes a set of an inverter and a converter (rectifying device). In the embodiment shown in FIG. 3, the AC commercial power supply 4 is converted into DC by the converter 3 and is used as the DC input of the inverter 2 to output the PW of the inverter 2.
A case of driving the brushless DC motor M with M output will be described. The inverter 2 is switching-controlled based on the switching signal mentioned later, and outputs a pseudo 3-phase alternating current (U, V, W). The AC output of the inverter 2 is supplied to the armature winding provided on the stator of the brushless DC motor M.

The inverter control device 1 in the embodiment of the present invention generates a switching signal for driving the brushless DC motor M with a sine wave current.

The measuring means 15 of the inverter control device 1 is
The current value of the armature current of the brushless DC motor output from the inverter 2 is measured. In the embodiment shown in FIG. 3, two phases out of three phases (U, V, W) are detected, and the remaining one phase is calculated by regarding the three phases (U, V, W) as balanced. To do.

The holding means 11 holds the phase of the switching signal output from the output means 14 at the time when the armature current is measured until the estimation of the rotor position by the estimation means 12 described later is completed.

The armature current measured by the measuring means 15 is also input to the estimating means 12.

The estimating means 12 estimates and calculates the rotor position of the brushless DC motor M from the armature current using the above-mentioned extended magnetic flux observer theory.

The calculating means 13 holds the phase of the switching signal held in the holding means 11 at the time when the armature current is measured, and the estimated brushless DC.
The phase difference between the phase of the rotor position of the motor M, that is, the error, is calculated to generate the phase correction value. in this way,
Since the phase of the switching signal held at the time when the armature current is measured is compared with the phase of the rotor position by using the armature current at this time, the temporal timing can be the same.

The phase correction value is added to the switching signal described later to adjust the output of the inverter 2. If the phase of the rotor position is behind the phase of the switching signal, a phase correction value that raises the output voltage of the inverter 2 is generated. If the phase of the rotor position leads the phase of the switching signal, a phase correction value that causes the output voltage of the inverter 2 to drop is generated. For example, when the error between the phase of the switching signal held in the holding means 11 and the phase of the rotor position estimated by the estimating means 12 is -1 degree, the phase correction value is +1 degree.

The output means 14 is a PWM signal generator which generates a switching signal for controlling the switching of the inverter 2 based on the frequency command value input from the outside. The output unit 14 in the present embodiment does not feed back the information obtained from the armature current until the above-described phase correction value is generated, and continuously outputs the signal sequence of the switching signal currently being output. On the other hand, when the phase correction value is generated, the signal sequence of the new switching signal in which the phase of the switching signal currently being output is corrected based on the phase correction value is output. As a result, the output of the inverter 2 is adjusted.

After the armature current is measured, the inverter 2
The time required to generate a switching signal for PWM control of the
It depends on the WM signal generation capability. For example, in a commonly used microcomputer, it takes 100 to 200 μs only to obtain a current measurement value and then use only the current measurement value to generate a PWM signal.

According to the present invention, the feedback control based on the measured armature current is completed in consideration of the calculation processing capability of the microcomputer, the PWM signal generation capability, etc. regarding the interval (sampling interval) for measuring the armature current. It is set to be longer than the time required for. That is, in the present invention, since the feedback control cycle is determined by setting the interval for measuring the armature current, it is possible to realize a system configuration that makes the best use of the performance of the microcomputer provided in the inverter control device. is there. For example,
A low-cost microcomputer used in the conventional 120-degree energization system can be sufficiently realized.

Next, the lead angle control in the inverter device according to the present invention will be described.

Generally, the higher the rotation speed of the motor,
Motor efficiency decreases. In the lead angle control, a reduction in motor efficiency is prevented by advancing the power supply timing to the motor.

For example, in the lead angle control in the conventional 120-degree energization method, it is realized by advancing the timing of commutation (low to high, high to off, and off to low) of each phase by the control amount relative to the actual position. is doing.

On the other hand, in the present invention, when the phase correction value is generated by the calculating means 13 described above, the phase of the switching signal currently being output is added to the phase correction value and further to the lead angle control amount. It is shifted to form a new switching signal sequence. In FIG. 3, the lead angle control means is shown as reference numeral 16.

For example, consider a case where a lead angle control amount of 30 degrees is set. When the calculating unit 13 calculates that the error between the phase of the switching signal held in the holding unit 11 and the phase of the rotor position estimated by the estimating unit 12 is −1 degree, the phase correction value is +1. Therefore, the phase of the switching signal currently being output is shifted by 31 ° in combination with the advance angle control amount of 30 °, and the signal is output as a new switching signal sequence.

With the configuration described above, when the rotational speed of the brushless DC motor changes due to load fluctuations and the phase shift occurs, the sine wave output from the inverter 2 is temporarily discontinued, but the rotation is stable. If it does, it will return to a continuous sine wave.

In FIG. 3, the portion indicated by reference numeral 1 can be realized by various arithmetic units such as a microcomputer or DSP.

Note that it is necessary to carefully set the cycle of the feedback control because the control cannot be performed sufficiently depending on the rotation speed of the brushless DC motor.

In describing the present invention, a brushless DC motor is taken up as a non-commutator motor in the present specification, but it is also classified as a non-commutator motor and is a structurally similar to a brushless DC motor. It can also be applied to reluctance motors.

[0071]

As described above, according to the present invention,
Since the commutatorless motor is driven by a sine wave, it has less electrical or mechanical noise and vibration than the 120-degree energization method, is more efficient, and can be controlled according to the performance of the microcomputer or DSP that executes arithmetic processing. Because it is possible
Even a low-performance inexpensive microcomputer or a computing unit such as a DSP can easily control the rotation of the commutatorless motor by the 180-degree energization method.

Brushless D, which is a kind of commutatorless motor
C motors are widely used because of their extremely simple structure and high efficiency, but they are also classified as non-commutator motors, and the present invention can also be applied to synchronous reluctance motors that are structurally similar to brushless DC motors. Is.

[Brief description of drawings]

FIG. 1 is a functional block diagram of an inverter control device according to the present invention.

FIG. 2 is a diagram showing an analytical model of a brushless DC motor.

FIG. 3 is a block diagram of an inverter control device according to an embodiment of the present invention.

FIG. 4 is a diagram illustrating a conventional 120-degree energization method in rotation control of a brushless DC motor,
(A) is a schematic connection diagram of an armature winding provided in a stator of a brushless DC motor, and (b) is a timing chart of output current waveforms of respective phases.

FIG. 5 is a timing chart of output current waveforms of respective phases in a 180-degree conduction system according to a conventional example in rotation control of a brushless DC motor.

[Explanation of symbols]

1 ... Inverter control device 2 ... Inverter 3 ... Converter 4 ... Commercial AC power supply 11 ... Holding means 12 ... Estimating means 13 ... Calculation means 14 ... Output means 15 ... Measuring means 16 ... Lead angle control means M ... Brushless DC motor

Continued front page    F-term (reference) 5H550 BB05 CC05 DD08 HB07 JJ03                       JJ04 JJ25 LL14 LL22 LL29                       LL35                 5H560 BB04 BB12 BB18 DA12 DB12                       DC12 EB01 RR01 SS07 TT15                       UA07 XA12 XA15

Claims (6)

[Claims] 1. An inverter control method for generating a switching signal for controlling switching of an inverter so as to drive a non-rectifier motor by a sine wave, using an armature current of the non-rectifier motor, wherein the armature current The holding step of holding the phase of the switching signal at the time of measuring, the estimating step of estimating the rotor position of the non-commutator motor from the armature current, the phase of the held switching signal, and the estimated An operation step of calculating an error from the phase of the rotor position to generate a phase correction value, and continuously outputting the signal sequence of the switching signal currently being output until the phase correction value is generated, When the phase correction value is generated, the phase of the switching signal currently being output is corrected based on the phase correction value, and a new switching signal is generated. An output step of outputting a signal sequence,
An inverter control method comprising: 2. In the estimating step, the armature resistance is R, the armature inductance on the dq coordinate axes is L _d and L _q , the rotor angular velocity on the dq coordinate axes is ω _e ,
The armature currents on the αβ axis are i _α and i _β , the armature voltages on the αβ axis are v _α and v _β, and the fundamental wave components of the equivalent induced voltage on the αβ axis are e _α1 and e _β1 , αβ. The DC component of the equivalent induced voltage on the axis is e _αdc and e _βdc , and the number of interlinkage magnetic fluxes on the axis αβ is ψ _α 'and ψ _β '
When, To apply the same dimension observer the flux linkage number [psi _alpha '
And ψ _β ', the fundamental wave components e _α1 and e _{β1 of} the equivalent induced voltage, and the DC component e of the equivalent induced voltage.
_When α _dc and e β _dc are estimated, and the electrical angle indicating the rotor position with respect to the α axis or the β axis is θ _e , the estimated interlinkage magnetic flux numbers ψ _α 'and ψ _β ' , [Equation 2] [Equation 3] The inverter control method according to claim 1, wherein the phase of the rotor position is estimated by substituting into the. 3. The inverter control method according to claim 1, further comprising a lead angle control step of further adding a lead angle control amount to the new switching signal when the phase correction value is generated. 4. An inverter control device for generating a switching signal for controlling switching of an inverter so as to drive a non-rectifier motor by a sine wave, using an armature current of the non-rectifier motor, wherein the armature current is The holding means for holding the phase of the switching signal at the time of measuring, the estimating means for estimating the rotor position of the non-commutator motor from the armature current, the phase of the held switching signal, and the estimated A calculating unit that calculates an error from the phase of the rotor position to generate a phase correction value, and continuously outputs the signal sequence of the switching signal that is currently being output until the phase correction value is generated, When the phase correction value is generated, the phase of the switching signal currently being output is corrected based on the phase correction value, and the signal sequence of the new switching signal is corrected. Inverter control apparatus comprising: a, and output means for outputting. 5. The estimating means comprises: armature resistance R; armature inductance L _d and L _{q on} the dq coordinate axes; rotor angular velocity ω _{e on} the dq coordinate axes;
The armature currents on the αβ axis are i _α and i _β , the armature voltages on the αβ axis are v _α and v _β, and the fundamental wave components of the equivalent induced voltage on the αβ axis are e _α1 and e _β1 , αβ. The DC component of the equivalent induced voltage on the axis is e _αdc and e _βdc , and the number of interlinkage magnetic fluxes on the axis αβ is ψ _α 'and ψ _β '
When, To apply the same dimension observer the flux linkage number [psi _alpha '
And ψ _β ', the fundamental wave components e _α1 and e _{β1 of} the equivalent induced voltage, and the DC component e of the equivalent induced voltage.
_When α _dc and e β _dc are estimated, and the electrical angle indicating the rotor position with respect to the α axis or the β axis is θ _e , the estimated interlinkage magnetic flux numbers ψ _α 'and ψ _β ' , [Equation 5] [Equation 6] The inverter control device according to claim 4, wherein the phase of the rotor position is estimated by substituting into 6. The inverter control device according to claim 4, further comprising a lead angle control means for further adding a lead angle control amount to the new switching signal when the phase correction value is generated.