KR20180122917A - Method and apparatus for transmitting and receiving a signal for low peak-to-average power ratio signal in wireless communications systems - Google Patents

Abstract

The present disclosure relates to a communication technique for fusing IoT technology with a 5G communication system for supporting a higher data transmission rate than a 4G system, and a system thereof. The present disclosure can be applied to intelligent services (for example, smart home, smart building, smart city, smart car or connected car, healthcare, digital education, retail business, security- and safety-related services, etc.) based on 5G communication technology and IoT-related technology. The present invention can reduce a peak-to-average power ratio (PAPR) by performing time domain circulation filtering. Also, a data transmission rate or coverage can be improved by selecting and transmitting a transmission waveform from CP-OFDM or DFT-s-OFDM.

Description

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a method and apparatus for reducing peak-to-average power ratio in a wireless communication system,

This disclosure relates generally to wireless communication systems, and more specifically to an apparatus and method for reducing peak-to-average power ratio (PAPR) in a wireless communication system.

Efforts are underway to develop an improved 5G or pre-5G communication system to meet the growing demand for wireless data traffic after commercialization of the 4G communication system. For this reason, a 5G communication system or a pre-5G communication system is called a system after a 4G network (Beyond 4G network) communication system or after a LTE system (Post LTE). To achieve a high data rate, 5G communication systems are being considered for implementation in very high frequency (mmWave) bands (e.g., 60 gigahertz (60GHz) bands). In order to mitigate the path loss of the radio wave in the very high frequency band and to increase the propagation distance of the radio wave, in the 5G communication system, beamforming, massive MIMO, full-dimension MIMO (FD-MIMO ), Array antennas, analog beam-forming, and large scale antenna technologies are being discussed. In order to improve the network of the system, the 5G communication system has developed an advanced small cell, an advanced small cell, a cloud radio access network (cloud RAN), an ultra-dense network, (D2D), a wireless backhaul, a moving network, cooperative communication, Coordinated Multi-Points (CoMP), and interference cancellation Have been developed. In addition, in the 5G system, the Advanced Coding Modulation (ACM) scheme, Hybrid FSK and QAM Modulation (FQAM) and Sliding Window Superposition Coding (SWSC), the advanced connection technology, Filter Bank Multi Carrier (FBMC) (non-orthogonal multiple access), and SCMA (sparse code multiple access).

On the other hand, the Internet is evolving into an Internet of Things (IoT) network in which information is exchanged between distributed components such as objects in a human-centered connection network where humans generate and consume information. IoE (Internet of Everything) technology, which combines IoT technology with big data processing technology through connection with cloud servers, is also emerging. In order to implement IoT, technology elements such as sensing technology, wired / wireless communication, network infrastructure, service interface technology and security technology are required. In recent years, sensor network, machine to machine , M2M), and MTC (Machine Type Communication). In the IoT environment, an intelligent IT (Internet Technology) service can be provided that collects and analyzes data generated from connected objects to create new value in human life. IoT is a field of smart home, smart building, smart city, smart car or connected car, smart grid, health care, smart home appliance, and advanced medical service through fusion of existing information technology . ≪ / RTI >

Accordingly, various attempts have been made to apply the 5G communication system to the IoT network. For example, technologies such as a sensor network, a machine to machine (M2M), and a machine type communication (MTC) are implemented by techniques such as beamforming, MIMO, and array antennas It is. The application of the cloud RAN as the big data processing technology described above is an example of the convergence of 5G technology and IoT technology.

Efforts are underway to develop improved 5G (5th generation) communication systems or pre-5G communication systems to meet the growing demand for wireless data traffic after commercialization of 4G (4th generation) communication systems. For this reason, a 5G communication system or a pre-5G communication system is referred to as a 4G network (Beyond 4G Network) communication system or a LTE (Long Term Evolution) system (Post LTE) system.

To achieve a high data rate, 5G communication systems are being considered for implementation in very high frequency (mmWave) bands (e.g., 60 gigahertz (60GHz) bands). In the 5G communication system, beamforming, massive MIMO, full-dimensional MIMO, and FD-MIMO are used in order to mitigate the path loss of the radio wave in the very high frequency band and to increase the propagation distance of the radio wave. ), Array antennas, analog beam-forming, and large scale antenna technologies are being discussed.

In addition, in order to improve the network of the system, the 5G communication system has developed an advanced small cell, an advanced small cell, a cloud radio access network (cloud RAN), an ultra-dense network, (D2D), a wireless backhaul, a moving network, cooperative communication, Coordinated Multi-Points (CoMP), and interference cancellation Have been developed.

In addition, in the 5G system, the Advanced Coding Modulation (ACM) scheme, Hybrid Frequency Shift Keying and Quadrature Amplitude Modulation (FQAM) and Sliding Window Superposition Coding (SWSC), and the Advanced Connection Technology (FBMC) ), Non-Orthogonal Multiple Access (NOMA), and Sparse Code Multiple Access (SCMA).

The communication technology required for the next generation Internet of things (IoT) technology is the Internet of things, which is very different from the existing cellular communication system. Particularly, in a conventional cellular communication system, if data rate and quality of service (QoS) are measures of communication quality, it is necessary to guarantee a very large number of connectivity in the IoT environment, The peak-to-average power ratio (PAPR), which can reduce the driving power due to the limitation of the battery, is important. In order to guarantee the performance of the cell edge user and the coverage lighthouse of the next generation mobile communication, a certain amount of power boost is required. However, the amount of power boost is closely related to the PAPR due to the nonlinearity of the power amplifier, Lowering PAPR is directly related to coverage enhancement performance.

Recently, as a result of the development of LTE (Long Term Evolution) and LTE-Advanced, a method and an apparatus for reducing a peak-to-average power ratio (PAPR) in a wireless communication system are needed.

Based on the foregoing discussion, the disclosure provides a method and apparatus for effectively reducing the peak-to-average power ratio (PAPR) of a signal in a wireless communication system.

The present disclosure also provides a method and apparatus for reducing the PAPR of a signal transmitted at a transmitting end of a wireless communication system.

The present disclosure also provides a method and apparatus for selectively applying spectral shaping and DFT spreading in a transmitter of a wireless communication system.

The present disclosure also provides a method and apparatus for increasing the data rate while reducing the PAPR of a signal transmitted at a transmitting end of a wireless communication system.

The present disclosure also provides a method and apparatus for increasing the reception ratio of a transmitted signal by reducing the PAPR at the receiving end of a wireless communication system.

According to various embodiments of the present disclosure, a method of operating a transmitting end in a wireless communication system includes: identifying resources allocated to transmit data to a receiving end; applying a filter to the data; Mapping the mapped data to a time domain, and transmitting the data converted to the time domain, wherein the filter is configured to determine, based on the allocated resource, And can be shared by the transmitting end and the receiving end.

The apparatus and method according to various embodiments of the present disclosure may reduce PAPR by performing time domain cyclic filtering or performing filtering that weighting on frequency domain subcarriers. Also, transmission data can be improved or coverage can be improved by selecting a transmission waveform from CP-OFDM or DFT-s-OFDM.

The effects obtainable from the present disclosure are not limited to the effects mentioned above, and other effects not mentioned can be clearly understood by those skilled in the art from the description below will be.

1 illustrates an example of a wireless communication system in accordance with various embodiments of the present disclosure.
2 illustrates an example of a configuration of an apparatus in a wireless communication system according to various embodiments of the present disclosure.
3 shows an example of the configuration of a communication section of a transmitting end in a wireless communication system according to various embodiments of the present disclosure.
4 shows an example of the configuration of a communication unit of a terminal transmitting terminal in a cellular system according to various embodiments of the present disclosure.
Figure 5 illustrates the effect according to various embodiments of the present disclosure.

The terms used in this disclosure are used only to describe certain embodiments and may not be intended to limit the scope of other embodiments. The singular expressions may include plural expressions unless the context clearly dictates otherwise. Terms used herein, including technical or scientific terms, may have the same meaning as commonly understood by one of ordinary skill in the art. The general predefined terms used in this disclosure may be interpreted as having the same or similar meaning as the contextual meanings of the related art and, unless explicitly defined in the present disclosure, include ideally or in an excessively formal sense . In some cases, the terms defined in this disclosure can not be construed to exclude embodiments of the present disclosure.

In the various embodiments of the present disclosure described below, a hardware approach is illustrated by way of example. However, the various embodiments of the present disclosure do not exclude a software-based approach, since various embodiments of the present disclosure include techniques that use both hardware and software.

The present disclosure relates to an apparatus and method for transmitting and receiving signals in a wireless communication system. In particular, this disclosure describes various embodiments for reducing peak-to-average power ratio (PAPR) in a wireless communication system.

A term referring to network entities (e.g., a transmitting end, a receiving end), a term referring to signal processing means (e.g., a filter), a term referring to a component of the apparatus And the like are illustrated for convenience of explanation. Accordingly, the present disclosure is not limited to the following terms, and other terms having equivalent technical meanings can be used.

1 illustrates a wireless communication system in accordance with various embodiments of the present disclosure. 1 illustrates a transmitting end 110 and a receiving end 120 as a part of nodes using a wireless channel in a wireless communication system. 1 shows one transmitting terminal 110 and one receiving terminal 120, but may include a plurality of transmitting terminals or a plurality of receiving terminals. For the sake of convenience of explanation, in this document, the transmitting terminal 110 and the receiving terminal 120 are described as separate entities, but the functions of the transmitting terminal 110 and the receiving terminal 120 may be mutually changed. For example, in an uplink of a cellular communication system, a transmitter 110 may be a terminal, a receiver 120 may be a base station, and a transmitter 110 may be a base station and a receiver 120 may be a terminal in a downlink. The PAPR reduction scheme according to various embodiments of the present disclosure can be applied not only to the uplink but also to the downlink.

The transmitting end 110 and the receiving end 120 can transmit and receive wireless signals in the millimeter wave band (e.g., 28 GHz, 30 GHz, 38 GHz, and 60 GHz). At this time, in order to improve the channel gain, the transmitter 110 and the receiver 120 may perform beamforming. Here, beamforming includes transmit beamforming and receive beamforming. That is, the transmitting end 110 and the receiving end 120 may assign a directivity to a transmission signal or a reception signal. To this end, the transmitting end 110 and the receiving end 120 may select at least one serving beam through a beam search procedure. According to various embodiments of the present disclosure, due to the high frequency characteristics of the millimeter waveband, excellent PAPR reduction performance may be required.

In addition, a wireless communication system according to various embodiments of the present disclosure can provide services requiring connection of a large-scale terminal. Here, the service may be referred to as a massive machine type communication (mMTC) service. For example, the mMTC service can be used for the internet of things (IoT) technology. The mMTC service may require a large coverage to provide services for a large number of terminals. According to various embodiments of the present disclosure, the transmitter 110 and the receiver 120 are devices for mMTC services, and may require excellent PAPR reduction performance due to miniaturization or battery limitation.

PAPR is an important measure for determining the driving power of a mobile device. When PAPR is high, serious distortion occurs when passing through a nonlinear device such as an analog-to-digital converter (ADC), a digital-to-analog converter (DAC), or a power amplifier. Therefore, high-performance devices with high linearity are required, Even if a high-performance semiconductor device satisfying linearity is used, a power source voltage must be sufficiently high when a FET (field effect transistor) device such as a complementary metal oxide semiconductor (CMOS) device is used. Therefore, when the PAPR of the transmission signal is high, there is a high possibility of generating a large problem in the uplink in which the power and the battery are limited. In particular, in a mMTC environment of 5G (5th generation) communication in which a large number of low- It is essential to lower the PAPR. In 4G (4th generation) communication (eg LTE (Long Term Evolution), DFT (spread-spectrum-OFDM) with a single carrier property instead of orthogonal frequency division multiplexing (OFDM) The DFT-S-OFDM scheme is based on the PAPR scheme, which is based on the PAPR scheme, and the PAPR scheme is used in the next generation mobile communication system. It is not enough to boost the power when the performance and amplifier performance are taken into consideration. [0004] Techniques have been proposed in the prior art to reduce the PAPR by cutting out signals exceeding the maximum value or adding additional frequency resources, In the beginning, the DFT-S-OFDM is modified to lower the PAPR without additional frequency resources. In order to minimize the reception performance degradation by changing the receiver structure All.

In accordance with various embodiments of the present disclosure, transmission and reception of binary phase shift keying (BPSK) symbols with improper properties is described as an example. However, the scope of the present disclosure is not limited to transmission and reception of BPSK symbols, and it is also applicable to transmission and reception of symbols or other types of symbols having improper properties. When the pseudo covariance matrix of the symbol vector is a zero matrix, the symbol vector is defined as proper. If the pseudo-covariance matrix of the symbol vector is a matrix other than a zero matrix, the symbol vector is defined as improper. The PAPR reduction scheme according to various embodiments of the present disclosure may employ a widely linear receiver that utilizes the impropriety nature of BPSK symbols.

2 illustrates a configuration of an apparatus in a wireless communication system according to various embodiments of the present disclosure. That is, the configuration illustrated in FIG. 2 can be understood as a configuration of the transmitting end 110 or the receiving end 120. [ Used below '... Wealth, '... Quot; and the like denote a unit for processing at least one function or operation, and may be implemented by hardware, software, or a combination of hardware and software.

Referring to FIG. 2, the apparatus may include a control unit 210, a communication unit 220, and a storage unit 230.

The control unit 210 can control the overall operations of the apparatus. For example, the control unit 210 can transmit and receive a signal through the communication unit 220. In addition, the control unit 210 can write or read data in the storage unit 230. [ To this end, the control unit 330 may include at least one processor or a microprocessor, or may be part of a processor. In addition, a part of the communication unit 220 and the control unit 210 may be referred to as a communication processor (CP). In particular, according to various embodiments, the control unit 210 controls the communication unit 220 to perform an operation for PAPR reduction in modulating or demodulating a signal. In other words, the control unit 210 can control the operation of each of the components included in the communication unit 220. [ For example, the control unit 210 may control the transmitting end 110 or the receiving end 120 to perform operations according to various embodiments described below.

The communication unit 220 may perform functions for transmitting and receiving signals through a wireless channel. For example, the communication unit 220 may perform a function of converting a baseband signal and a bit string according to a physical layer standard of the system. For example, at the time of data transmission, the communication unit 220 can generate complex symbols by encoding and modulating transmission bit streams. Also, upon receiving data, the communication unit 220 can recover the received bit stream through demodulation and decoding of the baseband signal. In addition, the communication unit 220 up-converts the baseband signal to a radio frequency (RF) band signal and transmits the RF band signal through the antenna, down-converts the RF band signal received through the antenna to a baseband signal, conversion.

To this end, the communication unit 220 may include a transmission filter, a reception filter, an amplifier, a mixer, an oscillator, a digital to analog converter (DAC), an analog to digital converter (ADC) In addition, the communication unit 220 may include a plurality of transmission / reception paths. Further, the communication unit 220 may include at least one antenna array composed of a plurality of antenna elements. In terms of hardware, the communication unit 210 may be composed of a digital unit and an analog unit, and the analog unit may be divided into a plurality of sub-units according to operating power, Lt; / RTI >

The communication unit 220 transmits and receives signals as described above. Accordingly, the communication unit 220 may be referred to as a 'transmission unit', a 'reception unit', or a 'transmission / reception unit'. Also, in the following description, transmission and reception performed through the wireless channel are used to mean that the processing as described above is performed by the communication unit 220. [ The communication unit 220 may include a backhaul communication unit for communicating with another network entity connected through a backhaul network.

The storage unit 230 may store data such as a basic program, an application program, and setting information for operation of the apparatus. The storage unit 230 may be composed of a volatile memory, a nonvolatile memory, or a combination of a volatile memory and a nonvolatile memory. The storage unit 230 may provide the stored data according to the request of the control unit 210.

3 shows an example of the configuration of the communication unit of the transmitting end 110 in the wireless communication system according to various embodiments of the present disclosure. The configuration illustrated in FIG. 3 can be understood as a part of the configuration of the communication section 220. 3 includes a modulation block 305, a spectrum shaping block 310, a discrete Fourier transform (DFT) block 315, a reference signal (RS) generating block 320, a resource mapping resource mapping block 325, an inverse fast Fourier transform (IFFT) block 330, and a cyclic prefix (CP) insertion block 335.

The modulation block 305 receives a bit string encoded through an error correction encoder such as LDPC, Polar, and the like and a scrambler, and can generate a complex symbol corresponding to a given modulation scheme. According to various embodiments of the present disclosure, the complex symbol may be a data symbol modulated with? / 2-BPSK,? / 4-QPSK, BPSK, QPSK, 16-QAM, 64-QAM or 256-QAM. In particular, the π / 2-BPSK and π / 4-QPSK modulation schemes reduce the probability of signal reinforcement due to in-phase in time, further reducing the PAPR of the signal to be transmitted relative to the BPSK and QPSK modulation schemes, respectively.

이다. 즉, 심볼의 파워가 평균으로 정규화(normalized) 되어 있다고 할 때, 입력 비트열의 인덱스의 짝/홀 여부에 따라 해당 비트의 성상도가 각각

또는

로 변경됨을 의미한다. 아울러 본 개시에서 입력 비트 0/1에 대한 출력 심볼 매핑 규칙은 일대일대응 관계이면 충분하며, <표 1>의 대응 관계로 한정하지 않는다. 즉, <표 1>에서 일대일대응 관계를 바꾸어 1과 0이 각각

로 매핑될 수도 있다. 또한 성상도(constellation)의 경우에도 초기 위상값을 <표 1>과 같이 45°로 한정하지 않는다. 다만 입력 비트의 인덱스가 홀/짝으로 변경될 때 각각의 성상도 간 위상 차가 180°이면 충분하다. 일례로, <표 1>과 같은

또는

를 사용하는 성상도 대신

또는

를 성상도로 사용할 수 있다.For example, Table 1 shows the output complex symbol mapping rules corresponding to the input bits when using π / 2-BPSK modulation in the modulation block 305. The modulation block output complex-valued modulation symbols for a single bit b (2i) or b (2i + 1)

to be. That is, assuming that the power of the symbol is normalized to the average, the constellation of the corresponding bit depends on whether the index of the input bit string is mapped /

or

. &Lt; / RTI > In addition, in the present disclosure, the output symbol mapping rule for input bit 0/1 is sufficient if a one-to-one correspondence is satisfied, and is not limited to the correspondence in Table 1. In other words, we change the one-to-one correspondence in Table 1,

Lt; / RTI &gt; Also, in case of constellation, the initial phase value is not limited to 45 ° as shown in <Table 1>. However, when the index of the input bit is changed to a hole / pair, the phase difference between each constellation is preferably 180 degrees. For example, as shown in Table 1

or

Instead of using

or

Can be used as a constellation.

<Table 1> π / 2-BPSK modulation mapping

이다. 즉, 심볼의 파워가 평균으로 정규화(normalized) 되어 있다고 할 때, 입력 비트열의 2비트 쌍(pair) 순서에 따라 해당 비트 쌍의 성상도가 각각

또는

로 번갈아 사용됨을 의미한다. 아울러 본 개시에서 입력 비트 쌍 00/01/10/11에 대한 출력 심볼 매핑 규칙은 상기 π/2-BPSK의 경우와 같이 일대일대응 관계이면 충분하며, <표 2>의 대응 관계로 한정하지 않는다. 또한 성상도(constellation)의 경우에도 초기 위상값을 <표 2>와 같이 45°로 한정하지 않는다. 다만 연속되는 입력 비트 쌍의 각각의 성상도 간 위상 차가 90°이면 충분하다. 일례로, <표 2>와 같은

또는

를 사용하는 성상도 중 후자(초기 위상값 0°)를 초기 성상도로 사용할 수도 있다.As another example, Table 2 shows an output complex symbol mapping rule corresponding to an input bit when? / 4-QPSK modulation is used in a modulation block 305. Single bit b (4i), b (4i + 1), b (4i + 2), b for modulating (4i + 3) (modulation) block outputs complex values (complex-valued) modulation symbol

to be. That is, assuming that the power of the symbol is normalized to an average, the constellation of the corresponding bit pair is

or

. &Lt; / RTI > In addition, in the present disclosure, the output symbol mapping rule for the input bit pair 00/01/10/11 is sufficient as a one-to-one correspondence relationship as in the case of the? / 2-BPSK, and is not limited to the correspondence in Table 2. Also, in the case of constellation, the initial phase value is not limited to 45 ° as shown in Table 2. However, a phase difference of 90 degrees between constellations of successive input bit pairs is sufficient. For example, as shown in Table 2

or

(Initial phase value of 0 deg.) Among the constellation diagrams using the initial phase.

<Table 2> π / 4-QPSK modulation mapping

The spectral shaping block 310 may perform time domain spectrum shaping on the data symbols provided from the modulation block 305. 3, the spectrum shaping block 310 receives L complex data symbols from the modulation block 305, and outputs L complex data symbols that have undergone circular convolution with a given spectral shaping filter coefficient to a DFT block 315 Can be output. In other words, when spectrum shaping is applied for PAPR reduction, L filtered data equal to the number L of inputted data symbols is output without using additional frequency resources and transmitted to the DFT block 315.

The DFT block 315 may transform the data symbols in the frequency domain by performing a DFT on the data symbols provided from the spectral shaping block 310. Referring to FIG. 3, the DFT block 315 receives L complex symbols from the spectrum shaping block 310, converts the complex symbols into L frequency-domain symbols, and outputs the L frequency-domain symbols to the resource mapping block 325.

A reference signal (RS) generation block 320 may generate a promised signal between a transmitter and a receiver so that the receiver can use the receiver to perform channel estimation to smoothly demodulate data. That is, the signal generated in the reference signal generation block 320 may be a demodulation reference signal (DMRS), and the signal generation method may include a system frame structure, a size of allocated time / frequency resources, a DFT length, ID, and so on. According to various embodiments of the present disclosure, the reference signal generation block 320 may generate the DMRS of the size of the frequency resource allocated by the base station, i.e., the DFT length L, and output it to the resource mapping block 325.

The resource mapping block 325 maps the symbols provided from the DFT block 315 and the reference signal generation block 320 to the promised OFDM subcarrier and symbol positions between the transmitter and receiver and outputs the symbols to the corresponding IFFT input positions. According to various embodiments of the present disclosure, the L data symbols received from the DFT block 315 and the L DMRSs received from the reference signal generation block 320 maintain the single carrier nature of the data and are time divided into OFDM symbols to obtain low PAPR And can be mapped by time division multiplexing. For example, when there is a slot composed of 14 OFDM symbols, the first and second OFDM symbols are the uplink control channel (PUCCH), the third OFDM symbol is the DMRS, and the remaining 11 If the symbols are composed of a physical uplink shared channel (PUSCH), the L DMRSs input to the reference signal generation block 320 output the third OFDM symbol to the IFFT block 330, The L received data symbols are output to the IFFT block 330 when one OFDM symbol of the remaining 11 symbols is generated.

The IFFT block 330 may convert the signal provided from the resource mapping block 325 into a time domain. Referring to FIG. 3, the IFFT block 330 may convert N time-domain symbols by applying an inverse fast Fourier transform (IFFT) of size N to the symbols provided from the resource mapping block 325. At this time, the L symbols provided from the resource mapping block 325 are mapped to L subcarrier indexes corresponding to a predetermined frequency region between the transmitter and the receiver in N input subcarrier indexes, and input to the remaining (NL) And performs IFFT. The time domain data transformed by the IFFT block 330 may be provided to the CP insertion block 335.

The CP insertion block 335 may insert the CP into the time domain data. 3, the CP insertion block 335 inserts a symbol corresponding to the N-Nc + 1 to N index of the N symbols converted into the time domain by the IFFT block 330 in front of the index 1 symbol (Nc + N ) Symbols can be output after Parallel-to-Serial (P2S) conversion. The symbols output from the CP insertion block 335 may be up-converted through the DAC and the RF module and transmitted to the receiver 120 through the antenna.

According to various embodiments of the present disclosure, the spectral shaping filter coefficients may be selected by using one of a finite number of filters previously agreed upon between the transmitting and receiving ends, or by using a one-to-one correspondence filter coefficient according to the resource allocation size, Or use only one filter coefficient regardless of the DFT length.

According to various embodiments of the present disclosure, the spectrum shaping block 310 may be applied only to a particular modulation level and transmission waveform type. For example, in the uplink transmission of the cellular system, the base station may select one of CP-OFDM or DFT-s-OFDM waveforms to be transmitted to the UE, and spectrum shaping may be performed using π / 2-BPSK DFT- It can be applied only at the time of transmission.

In addition, the spectrum shaping is regarded as a capability of the UE, not a mandatory requirement. In the process of capability negotiation in which the UE transmits its capability to the BS during initial connection, the BS informs the UE of the spectrum shaping capability, And modulation scheme, it is possible to control the application of spectral shaping in uplink data transmission. For example, the base station can instruct the terminal to apply spectral shaping to the UE through the DCI or the RRC during the PUSCH transmission in the π / 2-BPSK DFT-s-OFDM with respect to the UE having the spectrum shaping capability.

Additionally, spectrum shaping may be capability negotiation associated with the power class of the terminal. For example, the capability information can be transmitted to the base station by mapping spectrum shaping applicability according to the power class of the UE. For example, spectrum shaping is not possible for a 23dBm terminal and spectral shaping is possible for a 26dBm terminal, so that a base station can implicitly determine whether spectrum shaping is applied to the UE when the corresponding power class information of the UE is obtained. In this case, spectrum shaping can always be applied to a terminal capable of spectral shaping by using a combination of a specific waveform and a modulation scheme during uplink data transmission (a form in which the base station and the terminal promise) It is also possible to change the application of spectral shaping by means of signaling.

Now, a description will be made in detail of a method for generating an uplink transmission signal of a UE of the present disclosure when transmitting a Physical Uplink Shared Channel (PUSCH) of a cellular system.

4 shows an uplink PUSCH transmission structure of a terminal of a cellular system as an example of a configuration of a communication unit of a transmitter 110 in a wireless communication system according to various embodiments of the present disclosure. The configuration illustrated in Fig. 4 can be understood as a part of the configuration of the communication unit 220. [ 4 includes a scrambling block 405, a modulation block 410, a layer mapping block 415, a spectrum shaping block 420, a DFT (or transform precoder) block 425 A precoding block 430, a resource mapping or resource element mapper block 435, and a CP-OFDM signal generation block 440.

이라 하면, 도 4의 변조(modulation) 블록 410은 비트열

을 입력 받아 기지국이 지시한 변조 방식에 따라 변조를 수행한 후 변조된 복소(complex-valued) 심볼

을 출력할 수 있다. 복소 변조 심볼들

은 레이어 매핑 (layer mapping) 블록 415를 거쳐 하나 이상의 엔트리(entry)를 갖는 레이어 심볼 벡터

,

에 매핑될 수 있다. 여기서 v는 레이어 수를 의미하며,

는 레이어 당 변조 심볼 수를 의미한다.The UE may generate one or more codewords through error correction coding using the promised rules with the base station. For each codeword q, scrambling may be performed in the scrambling block 405 of FIG. 4 as a promising rule with the base station. The output bit stream of the scrambling block 405

4, the modulation block 410 includes a bit string

And performs modulation according to the modulation scheme indicated by the base station. The modulated complex-valued symbol

Can be output. The complex modulation symbols

Via a layer mapping block 415, a layer symbol vector having one or more entries,

,

Lt; / RTI &gt; Where v is the number of layers,

Denotes the number of modulation symbols per layer.

In the 5G cellular system, CP-OFDM or DFT-s-OFDM can be used as the uplink waveform. To improve the data rate, CP-OFDM is used for MIMO transmission and a single layer (SISO) DFT-s-OFDM can be used for transmission. In other words, the terminal may apply the spectral shaping block 420 and the DFT block 425 according to the selection of the base station, or by-pass the spectrum shaping block 420 and the DFT block 425.

In addition, according to the selection of the base station, the terminal may bypass by only the spectrum shaping block 420 and apply the DFT block 425. This is so that spectrum shaping can be selectively transmitted according to the base station implementation.

이며,

관계가 성립하고, 도 4에서의 스펙트럼 셰이핑 블록 420의 입력은

이 된다.In particular, DFT-s-OFDM can be transmitted on a single layer, layer 0, where the role of the layer mapping block is to pass the input symbol to the output symbol without any processing. This can be expressed by the following equation

Lt;

And the input of the spectrum shaping block 420 in FIG. 4 is

.

를 추출해낼 수 있다. 이 값은 수학식

으로 표현될 수 있으며, 여기서

는 자원 할당의 단위인 리소스 블록 (resource block, RB) 개수로 표현되는 PUSCH 대역폭을 의미하며,

는 하나의 리소스 블록을 구성하는 OFDM 부반송파(subcarrier) 개수를 의미한다. 구체적으로, 5G 셀룰러 시스템에서

=12 일 수 있고,

는 단말의 DFT 구현 복잡도 감소를 위해 <수학식 1>을 만족시키는 유한한 자연수(natural number) 값들로 한정될 수 있다.The MS transmits the PUSCH resource allocation information received from the BS through a physical downlink control channel (PDCCH)

Can be extracted. This value can be calculated using Equation

Lt; RTI ID = 0.0 &gt;

Denotes a PUSCH bandwidth expressed by the number of resource blocks (RBs), which is a unit of resource allocation,

Denotes the number of OFDM subcarriers constituting one resource block. Specifically, in a 5G cellular system

= 12,

May be limited to finite natural number values that satisfy Equation (1) to reduce the DFT implementation complexity of the terminal.

&Quot; (1) &quot;

is a set of non-negative integers and

is the maximum number of subcarriers allocated for PUSCH in the system.where

is a set of non-negative integers and

is the maximum number of subcarriers allocated for PUSCH in the system.

The UE can be instructed to apply DFT-s-OFDM and spectral shaping when transmitting a PUSCH from a base station through a PDCCH or an RRC message. In particular, spectrum shaping applications can be applied in conjunction with specific modulation schemes. For example, a UE can apply spectrum shaping only to a limited number of time slots for transmission of π / 2-BPSK DFT-s-OFDM transmission or π / 4-QPSK DFT-s-OFDM transmission.

The concrete operation of the spectrum shaping block 420 and the DFT block 425 will now be described when the terminal applies spectrum shaping and DFT spreading in the PUSCH transmission.

In FIG. 4, the spectrum shaping block 420 performs spectral shaping before the DFT block 425, that is, in the time domain. This operation mathematically refers to a circular convolution of a given DFT block length input symbol vector and a given spectral shaping filter.

The spectrum shaping filter can be applied using the predetermined coefficients between the base station and the terminal. In this case, when the spectral shaping filter coefficient is defined in the frequency domain, that is, when the DFT spreading is defined as the symbol multiplication factor after the DFT spreading, the coefficients must be defined according to the DFT size type. To solve this problem, one frequency domain spectral shaping function is defined in the real area [0, 2π) or [-π, π), and the corresponding area is sampled uniformly according to the DFT size and then normalized . In this case, mathematically equivalent operations can be performed before DFT spreading, that is, by simply defining a time domain spectral shaping filter.

Equation (2) is an example of the definition of the time-domain spectrum shaping filter h _ss [i] of the present disclosure.

&Quot; (2) &quot;

is the delta function in discrete-time domain defined as follows:where

is the delta function in discrete-time domain defined as follows:

Equation (3) is another example of the definition of the time-domain spectral shaping filter of the present disclosure.

&Quot; (3) &quot;

is the delta function in discrete-time domain defined as follows:where

is the delta function in discrete-time domain defined as follows:

에 대해

개수만큼 분리되어 각

개 심볼에 대해 스펙트럼 셰이핑 필터 h_ss[i] 와 순환 컨벌루션(circular convolution) 연산을 수행하여

을 출력하며, 이는 <수학식 4>로 표현할 수 있다.The spectral shaping block 420 includes an input

About

As many as the number

The spectral shaping filter h _ss [i] is subjected to a circular convolution operation

, Which can be expressed by Equation (4).

&Quot; (4) &quot;

) and

is modulo operation with

. For example,

.where L _h is number of filter taps of h _ss [i] (

) and

is modulo operation with

. For example,

.

개 복소 심볼로 구성된

개 블록에 대해 각각 DFT를 수행하여 출력하며, 이는 <수학식 5>로 표현할 수 있다.The DFT block 425

Composed of complex symbols

DFT is performed for each block, and this can be expressed by Equation (5).

Equation (5)

,

이라 하면, 프리코딩 블록 420은 이를 입력받아 프리코딩 적용 후

,

을 각 안테나 포트 별 리소스 매핑 블록 435로 출력한다. 이때, 레이어 0에 해당하는 출력은 스펙트럼 셰이핑 블록 420, DFT 블록 425를 by-pass하고 바로 프리코더 블록 430의 입력으로 보내질 수 있다. CP-OFDM 전송에 해당하는 프리코딩에 대한 일반적인 수학식은 <수학식 6>과 같이 표현될 수 있다.The precoding block 430 applies spatial precoding to each layer symbol in SISO or MIMO transmission in a CP-OFDM waveform in the uplink. At this time, the precoder applied according to the transmission mode may be different. The output of the layer mapping block 415

,

, The precoding block 420 receives it and applies precoding

,

To the resource mapping block 435 for each antenna port. At this time, the output corresponding to the layer 0 can be bypassed by the spectrum shaping block 420 and the DFT block 425, and then directly sent to the input of the precoder block 430. A general equation for precoding corresponding to the CP-OFDM transmission can be expressed as Equation (6).

&Quot; (6) &quot;

In the uplink, when the UE transmits a single layer in a DFT-s-OFDM waveform, the output of the layer 0 can be transmitted to the input of the precoding block 430 through the spectrum shaping block 420 and the DFT block 425. At this time, The role is by-passing the corresponding signal, which can be expressed by Equation (7).

&Quot; (7) &quot;

4, DMRS (Demodulation RS) for channel estimation in the resource mapping block 435 and the CP-OFDM signal generation block 440 are inserted into the TDM or FDM form together with the PUSCH data symbols to generate a final CP-OFDM symbol .

In a single stream transmission in the DFT-s-OFDM waveform, the baseband signal can generally be expressed as Equation (8).

&Quot; (8) &quot;

는 기지국이 단말에게 할당해준 주파수 자원에 해당하는 부반송파 인덱스이며, 리소스 매핑 규칙에 따라 해당 인덱스는 달라질 수 있다.

는 OFDM의 부반송파 이격(subcarrier spacing)을, N_{CP ,l}은 l번째 OFDM 심볼의 CP 길이를 기준 시간 T_s로 나눈 정수 샘플 수를,

은 안테나 포트 p의 (k,l)번째 리소스 엘리먼트 (resource element)에 매핑될 송신 심볼을 각각 의미한다. (k는 주파수 축 부반송파 인덱스이며, l은 OFDM 심볼 인덱스를 의미한다.)here

Is a subcarrier index corresponding to a frequency resource allocated to the UE by the BS, and the corresponding index may be changed according to a resource mapping rule.

Denotes the number of integer samples obtained by dividing the CP length of the l &lt; th & gt _; OFDM symbol by the reference time T _s , N _CP denotes a subcarrier spacing of OFDM,

Denotes a transmission symbol to be mapped to a (k, l) -th resource element of the antenna port p, respectively. (k is a frequency axis subcarrier index, and 1 is an OFDM symbol index).

The spectral shaping coefficient of Equation (4) can be determined in consideration of performance, implementation complexity, etc. such as PAPR of the uplink transmission signal and EVM / BLER of the reception end. In terms of reduction of the implementation complexity, it may be unified by one coefficient such as Equation (2) or Equation (3), but the degree of freedom of implementation of the terminal may be increased by specifying specific conditions. For example, in the case of time domain spectrum shaping (TDSS) as shown in Equation (4), the following conditions can be specified instead of specifying fixed coefficients.

● Power normalization:

● The maximum number of filter tabs: For example, L _h ≤3 or L _h = 2

● Characteristics of filter coefficients: real-symmetric, conjugate-symmetric, equal-gain, low-pass, high-pass

● Dependency on DFT size: For example, whether the coefficient is function of L

The above conditions may be all defined at the same time, and some conditions may be selectively adopted. For example, the filter coefficient of Equation (2) satisfies the conditions of power normalization, maximum number of filter taps 2, real-symmetric, equal-gain, high-pass, and no dependence on the DFT size. The filter coefficient of Equation (3) satisfies the conditions of power normalization, the number of maximum filter taps 2, equal-gain, high-pass, and dependence on DFT size among the above conditions. In addition, the filter coefficient should be selected so that the base station can satisfy the EVM requirement even when the UE uses an unspecified coefficient.

The maximum number of filter taps among the above conditions is closely related to the delay and the implementation complexity of the filter. Since the time domain spectral shaping is performed by circular convolution operation, the complexity increases as the number of non-zero taps increases as the number of second-order / third-order polynomials increases. . Also, if there is no non-zero tap from h _ss [0] and thereafter, the order of data symbols before DFT is different from the order of data symbols after IDFT in transmission due to delay in characteristics of filters, . Accordingly, in order to prevent this, it is possible to prevent the circular movement phenomenon due to the filter delay of the data symbol by allowing the base station and the terminal to know each other by specifying the filter coefficient or by specifying the condition without specifying the filter coefficient.

The reason for the low-pass / high-pass distinction among the above conditions is that, when defining the uplink OFDM signal generation method regardless of the waveform type such as CP-OFDM and DFT-s- Pass filter in order to map the DFT output sequentially to the IFFT without generating further rules at the resource mapping block 435. [ If a low-pass filter is also defined, the mapping of the DFT output to the IFFT requires circular mapping of the half of the DFT size so that energy can be concentrated at the center of the band to which the transmission spectrum is allocated.

The weight symmetry condition for the coefficients among the above conditions can keep the group delay of the filter constant regardless of the frequency (that is, a linear phase filter becomes possible) There is an advantage that the complexity structure can be considered.

Equation (2) differs depending on whether the filter coefficients of Equation (2) and Equation (3) are dependent on the DFT size. In terms of energy symmetry of the transmission spectrum, Equation 3> is more preferable and it is easy to satisfy the ACLR condition and the like. In the frequency response of Equation (2), H [0] is always 0 and the remaining H [1], ... , H [L-1] are symmetrical with respect to H [L / 2], so that there is no energy symmetry with respect to the entire DFT size, and the shape slightly biased to the right (high frequency side) And the filter coefficient of Equation (3) plays a role of correcting it.

에 RRC의 roll-off 값 β(0≤β≤1)에 해당하는 톤 수 즉,

이하로 한정할 수 있다. Spectrum flatness 영역을 늘리려면 0에 가까운 β값을 설정하면 되고,

는 floor 함수를 의미한다 (.을 넘지않는 최대 정수)On the other hand, instead of the condition of the time-domain spectrum shaping filter, the terminal may be given an implementation degree of freedom under the condition of a frequency domain spectrum shaping (FDSS) filter (which may be called as window in the form of an operation). Since the FDSS coefficient extraction may vary according to the DFT size, in order to generate the coefficients in real time according to the DFT size, one of the well-known window functions is selected and the FDSS coefficients are sampled by sampling according to the DFT- . A window function such as Root Raised Cosine is typical. Since there is excess bandwidth corresponding to roll-off, we can consider truncated RRC window in which the coefficients of the roll-off region are all set to 0 for use without bandwidth extension only in DFT size . In this case, the region where the weights of the FDSS coefficients on the central portion of the transmission band are equal to each other is referred to as a 'spectrum flatness region', and the FDSS coefficients region affected by the roll- flatness' region, it is possible to maintain the spectrum flatness by limiting the number of tones by which the FDSS coefficient is multiplied by the non-flatness coefficient to a predetermined value, thereby reducing the influence even if the base station receiver does not know the accurate spectrum shaping coefficient You can do it. However, this method may need to optimally compromise the short / long term because the PAPR reduction capability of the transmission signal due to the spectrum shaping is degraded. Therefore, when the FDSS coefficient condition is to be defined as the tone number of the 'spectrum non-flatness' region, the corresponding tone number may be determined as a fixed value without depending on the allocated DFT size, or the DFT size As shown in FIG. For example, when the number of subcarriers in the physical resource block (PRB) is 12, the number of FDSS non-flatness tones may be limited to 2 or 4 or less considering the minimum resource allocation unit, As noted, the various DFT sizes

The number of tones corresponding to the roll-off value? (0??? 1) of the RRC,

Or less. To increase the spectral flatness area, set a value close to 0,

Means the floor function (the maximum integer not exceeding.)

If the BS does not use the promised spectral shaping coefficients and the UE performs the uplink transmission using the spectral coefficients within the limited conditions as described above, even if the BS does not know the spectrum shaping coefficient used by the UE A transmission method of a terminal for maintaining reception performance without deterioration will be described.

The transmission structure shown in FIG. 3 and FIG. 4 is a case where when the UE transmits data using the promised spectrum shaping coefficient, the BS and the MS transmit the DMRS symbol for the PUSCH without applying spectrum shaping, After demodulating and equalizing the channel through the DMRS symbol, the data symbols of the UE can be demodulated through appropriate post-processing (e.g., matched filtering) of the promised spectrum shaping. However, if the base station does not know the spectrum shaping coefficient used by the UE, the performance may deteriorate due to an effective channel change due to spectral shaping when using this scheme. In this case, if the terminal applies spectral shaping to the DMRS symbol, the base station considers the spectrum shaping as an effective channel combined with the actual channel, and performs channel estimation, equalization, and demodulation of the data symbols.

When transmitting PUSCH with DFT-s-OFDM, the DMRS can use sequences and patterns designed independently of the modulation scheme, as in the conventional LTE. For example, if there are π / 2-BPSK, QPSK, 16-QAM, 64-QAM, and 256-QAM modulation schemes available for PUSCH transmission in DFT-s-OFDM, spectral shaping is π / 2-BPSK . Therefore, spectrum shaping can be applied to the DMRS only in this case. However, since the DMRS is designed in the frequency domain without directly applying the transform precoding (DFT) and is directly mapped to the IFFT, the circular convolution filter defined in the time domain can not be immediately applied. In this case, it is possible to generate a DMRS to be used in the π / 2-BPSK DFT-s-OFDM transmission in which spectral shaping is applied in the following two methods (or devices).

1) Generate frequency domain coefficients by applying DFT to the time domain filter coefficients used for data symbol transmission, and then multiply the DMRS sequence by tone-by-tone, apply spectral shaping to DMRS, and map to IFFT

② Generate time domain sequence by applying IDFT to the designed DMRS in the frequency domain, and then perform time domain filter coefficient and cyclic convolution to use it for data symbol transmission and then map the DFT to IFFT again

을 곱하였다), DMRS에 곱할 주파수 영역 스펙트럼 셰이핑 (frequency domain spectrum shaping, FDSS) 계수는 다음과 같이 도출할 수 있다.When the method 1 is used, the corresponding process can be performed using the filter coefficient in the spectrum shaping block 310 in the RS generation block 320 of FIG. There is a need for a method / apparatus that allows the power of the signal to be normalized. The sum of the coefficient powers is normalized to 1 as in Equation (2) or Equation (3), and the DFT operation is normalized so that the input and output powers do not change as in Equation (5) When assuming that

, The frequency domain spectrum shaping (FDSS) coefficients to be multiplied by the DMRS can be derived as follows.

&Quot; (9) &quot;

이 삭제된 이유는 FDSS의 경우 tone-by-tone 형태로 이후 곱셈이 이루어질 것이기 때문에 할당된 톤에 적용될 FDSS 계수 파워 합이 1이 될 경우 곱셈 이후 신호 파워의 합이 1로 정규화 되어버려 원래 전송 데이터 심볼의 파워의 합 (보통 평균적으로 DFT 크기인 )보다 줄어들게 된다. 따라서 이를 방지하기 위해 FDSS 계수 H_ss[k]는 그 파워의 합이 1이 아닌 DFT 크기, 즉,

이 되어야 하며, <수학식 9>가 이를 반영한 생성 식이다.Unlike Equation (5), the normalization coefficient

The reason for the deletion is that if the sum of the power of the FDSS coefficients to be applied to the allocated tone is 1 since the multiplication will be performed after the tone-by-tone type in case of the FDSS, the sum of the signal powers after the multiplication is normalized to 1, The sum of the powers of the symbols (usually the average DFT size ). Therefore, in order to prevent this, the FDSS coefficient H _ss [k] is the DFT size in which the sum of the powers is not 1,

And Equation (9) is a generation formula that reflects this.

If the implementation of the DFT processing of the filter coefficient becomes burdensome each time, the output coefficients for each DFT size for the corresponding time domain filter coefficient are calculated and stored using a large amount of memory, The value may be read and multiplication may be performed with the DMRS sequence.

In case of uplink transmission through a precoder with two or more logical antenna ports, even though it is a single layer transmission, the DMRS symbols that are TDM and TDM symbols can be orthogonalized so that a DMRS resource (RE, resource element) interleaved and transmitted. For example, a DMRS of port 0 may be mapped to an even subcarrier index among allocated subcarrier resources, and a DMRS of port 1 may be mapped to an odd subcarrier index. At this time, spectrum shaping can be applied to the DMRS of each port as follows.

&Quot; (10) &quot;

That is, the DMRS sequence of each port is generated by half the allocated DFT size, and is multiplied by the DMRS of the port 0 using the spectrum shaping coefficient of the even index tone, and is mapped to the IFFT of the port 0 (zero insertion ) Can be multiplied by the DMRS of port 1 using the spectral shaping factor of the odd index tone and then mapped to the IFFT of port 1 (zero insertion in even index tones). In the same way, the spectrum shaping rules for the DMRS can be extended when transmitting to three or more multi-antenna ports. In this case, if the transmission power satisfies the transmission power normalization rules of Equations (2), (5), and (9) as described above, the same transmission power is maintained .

개의 FDSS 계수들 중에서 다시 해당 포트에 약속된 RE 자원에 해당하는 계수들만으로 그 파워 합이

이 되도록 한번 더 정규화할 수도 있고, FDSS 계수에 대한 재 정규화없이 DMRS에 power control 파라미터 (계수)를 적용하여 interleaving 간격 만큼 해당 비율로 데이터 심볼 전력 대비 줄어드는 전력을 보상시킬 수도 있고, 상기 2가지 방식을 결합시켜 정규화 과정을 진행할 수도 있다.On the other hand, when the terminal is designed to perform single port data transmission and the DMRS for the corresponding port has already been designed to interleave at a uniform interval for multi-port transmission, when spectrum shaping is applied to the corresponding DMRS, The FDSS coefficient is multiplied only by the RE promised by the DMRS in the resource and the FDSS coefficient is multiplied by 0 in the remaining RE promised for the other port so that the normalization may not be easy even if the FDSS coefficient is made as shown in Equation (9) . In this case, the equation

Of the FDSS coefficients, only the coefficients corresponding to the RE resources promised to the corresponding port are summed

Or power control parameters (coefficients) may be applied to the DMRS without re-normalization of the FDSS coefficients to compensate for the power that is reduced by the ratio of the data symbols to the data symbol power by the interleaving interval. The normalization process may be performed.

In the case of using the method 2, the DMRS symbol can be generated by applying the IDFT to the frequency domain DMRS and using the data symbol generating block of FIG. 3 as it is, instead of outputting the modulation block 305, by inputting the corresponding output as the input of the spectrum shaping block 310. If the application of the DMRS IDFT in real time is burdensome, the IDFT output values for the DMRS in the frequency domain are calculated and stored according to the frequency resource area location and the DFT size using a large amount of memory, and the resource scheduling The DMRS symbol can be generated by reading the value according to the instruction and inputting it into the spectrum shaping block 310.

On the other hand, the PAPR (about 2 to 5 dB in LTE) of the DMRS symbol for DFT-s-OFDM is applied to the π / 2-BPSK DFT- When the PAPR (about 1 to 3 dB level) of the s-OFDM data symbol is large, when the UE performs uplink transmission in the vicinity of the power amplifier threshold, the DMRS symbol having a relatively higher PAPR due to the nonlinearity of the power amplifier May be more degraded than the data symbols, resulting in deterioration in channel estimation performance at the time of base station reception. Therefore, in this case, similarly to the π / 2-BPSK DFT-s-OFDM data symbol, the spectral shaping is applied to the DMRS symbol to lower the PAPR, thereby improving the base station reception performance. In this case, since the base station knows the spectrum shaping coefficient in advance, the base station can effectively obtain the channel from which the spectral shaping effect applied to the DMRS is removed through various methods, or equalize using the effective channel combining the spectrum shaping as in the case where the spectral shaping coefficient is unknown.

When π / 2-BPSK modulation is performed with DMRS with PAPR with π / 2-BPSK DFT-s-OFDM reception performance using spectral shaping using DMRS designed irrespective of modulation scheme, In addition to the DMRS used for high-order modulation transmission such as QPSK and 16QAM, a DMRS specialized for PAPR reduction can be designed and operated.

It should be noted that the embodiments of the present invention disclosed in the present specification and drawings are only illustrative of the present invention in order to facilitate the understanding of the present invention and are not intended to limit the scope of the present invention. That is, it will be apparent to those skilled in the art that other modifications based on the technical idea of the present invention are possible. Further, each of the above embodiments can be combined with each other as needed.

Claims (1)

A method for processing a control signal in a wireless communication system,
Receiving a first control signal transmitted from a base station;
Processing the received first control signal; And
And transmitting the second control signal generated based on the process to the base station.

Images