KR102224679B1 - Apparatus and method for reducing peak-to-average power ratio in wireless communication system - Google Patents

Abstract

This disclosure relates to LTE (Long Term Evolution) and 4G (4 ^th generation) 5G to support higher data rates than the subsequent communication system (5 ^th generation) or pre-5G communication system such. The operating method of the transmitter in a wireless communication system includes a process of applying a filter to data, a process of mapping the filter-applied data to at least one subcarrier, and a process of transmitting the mapped data to a receiver, wherein the The filter is determined based on the allocated resources.

Description

A device and method for reducing the peak-to-average power ratio in a wireless communication system {APPARATUS AND METHOD FOR REDUCING PEAK-TO-AVERAGE POWER RATIO IN WIRELESS COMMUNICATION SYSTEM}

The present disclosure generally relates to a wireless communication system, and more particularly, to an apparatus and method for reducing a peak-to-average power ratio (PAPR) in a wireless communication system.

4G (4 ^th generation) to meet the traffic demand in the radio data communication system increases since the commercialization trend, efforts to develop improved 5G (5 ^th generation) communication system, or pre-5G communication system have been made. For this reason, the 5G communication system or the pre-5G communication system is called a Beyond 4G Network communication system or a Long Term Evolution (LTE) system (Post LTE) system.

In order to achieve a high data rate, 5G communication systems are being considered for implementation in an ultra-high frequency (mmWave) band (eg, such as a 60 gigabyte (60 GHz) band). In order to mitigate the path loss of radio waves in the ultra-high frequency band and increase the propagation distance of radio waves, in 5G communication systems, beamforming, massive MIMO, and Full Dimensional MIMO (FD-MIMO) ), array antenna, analog beam-forming, and large scale antenna technologies are being discussed.

In addition, in order to improve the network of the system, in 5G communication system, advanced small cell, advanced small cell, cloud radio access network (cloud RAN), ultra-dense network , Device to Device communication (D2D), wireless backhaul, moving network, cooperative communication, CoMP (Coordinated Multi-Points), and interference cancellation And other technologies are being developed.

In addition, in the 5G system, the advanced coding modulation (Advanced Coding Modulation, ACM) method of FQAM (Hybrid Frequency Shift Keying and Quadrature Amplitude Modulation) and SWSC (Sliding Window Superposition Coding), and advanced access technology, FBMC (Filter Bank Multi Carrier). ), NOMA (Non Orthogonal Multiple Access), and SCMA (Sparse Code Multiple Access) are being developed.

Communication technologies required for next-generation Internet of Things (IoT) technologies include the Internet of Things, and their characteristics are very different from existing cellular communication systems. In particular, in the existing cellular communication system, if data rate and quality of service (QoS) were measures of communication quality, in the IoT environment, a very large number of connectivity must be guaranteed, and the miniaturization of mobile devices and The peak-to-average power ratio (PAPR), which can reduce driving power, is important due to limitations of the battery and the like. In addition, a certain amount of power boost is required to ensure the performance of the next-generation mobile communication coverage lighthouse and cell edge users, but the amount of power boost available is closely related to PAPR due to the nonlinearity of the power amplifier. Lowering the PAPR is directly related to the coverage enhancement performance.

Based on the above discussion, the present disclosure provides an apparatus and method for effectively reducing a peak-to-average power ratio (PAPR) of a signal in a wireless communication system.

In addition, the present disclosure provides an apparatus and method for reducing PAPR of a signal transmitted at a transmitting end of a wireless communication system.

In addition, the present disclosure provides an apparatus and method for increasing a data transmission rate while reducing PAPR of a signal transmitted from a transmitting end of a wireless communication system.

In addition, the present disclosure provides an apparatus and method for reducing PAPR of a signal received at a receiving end of a wireless communication system.

In addition, the present disclosure provides an apparatus and method for increasing a data reception rate while reducing a PAPR of a signal received at a receiving end of a wireless communication system.

According to various embodiments of the present disclosure, a method of operating a transmitter in a wireless communication system includes a process of checking an allocated resource for transmitting data to a receiver, a process of applying a filter to the data, and the data to which the filter is applied. Mapping to one or more subcarriers, converting the mapped data to a time domain, and transmitting data converted to the time domain, wherein the filter is determined based on the allocated resources And may be shared by the transmitting end and the receiving end.

In addition, according to various embodiments of the present disclosure, a method of operating a transmitter in a wireless communication system includes a process of applying a filter to data, a process of mapping the filter-applied data to at least one subcarrier, and the mapping to a receiver. Transmitting the data, and the filter is determined based on the allocated resources.

In addition, according to various embodiments of the present disclosure, a method of operating a receiver in a wireless communication system includes a process of receiving data from a transmitter, a process of applying a conversion matrix to the data, and a process of decoding the data to which the conversion matrix is applied. It may include. The transformation matrix may be determined based on a filter shared by the transmitting end and the receiving end. The filter may be determined based on receiver characteristics and reception performance of the receiver.

In addition, according to various embodiments of the present disclosure, in a wireless communication system, an apparatus of a transmitting end may include a transmitting/receiving unit and at least one processor coupled to the transmitting/receiving unit. The at least one processor may apply a filter to data and map the filter-applied data to at least one subcarrier. The transceiving unit may transmit the mapped data to a receiving end. The filter may be determined based on receiver characteristics and reception performance of the receiver.

In addition, according to various embodiments of the present disclosure, in a wireless communication system, an apparatus of a receiving end may include a transmitting/receiving unit and at least one processor coupled to the transmitting/receiving unit. The transceiving unit may receive data from a transmitting end. The at least one processor may apply the transformation matrix and decode data to which the transformation matrix is applied. The transformation matrix may be determined based on a filter shared by the transmitting end and the receiving end. The filter may be determined based on receiver characteristics and reception performance of the receiver.

The apparatus and method according to various embodiments of the present disclosure may reduce PAPR by performing filtering set in consideration of a reception performance gain of a receiving end.

The effects obtainable in the present disclosure are not limited to the effects mentioned above, and other effects not mentioned may be clearly understood by those of ordinary skill in the technical field to which the present disclosure belongs from the following description. will be.

1 illustrates an example of a wireless communication system according to various embodiments of the present disclosure.
2 illustrates an example of a configuration of an apparatus in a wireless communication system according to various embodiments of the present disclosure.
3 illustrates an example of a configuration of a communication unit of a transmitting end in a wireless communication system according to various embodiments of the present disclosure.
4 illustrates an example of input and output of a phase rotation block at a transmitting end of a wireless communication system according to various embodiments of the present disclosure.
5A illustrates an example of input and output of a filter block at a transmitting end of a wireless communication system according to various embodiments of the present disclosure.
5B is a diagram illustrating an example of a filter for a case in which an additional frequency resource is not allocated at a transmitting end of a wireless communication system according to various embodiments of the present disclosure.
5C is a diagram illustrating an example of a filter when an additional frequency resource is allocated at a transmitting end of a wireless communication system according to various embodiments of the present disclosure.
6A illustrates an example of a polynomial for determining filter coefficients at a transmitting end of a wireless communication system according to various embodiments of the present disclosure.
6B illustrates an example of filter coefficient sampling when the number of symbols is 12 at the transmitting end of the wireless communication system according to various embodiments of the present disclosure.
6C illustrates an example of filter coefficient sampling when the number of symbols is 32 at the transmitting end of the wireless communication system according to various embodiments of the present disclosure.
7A illustrates an operating method for determining filter coefficients at a transmitting end of a wireless communication system according to various embodiments of the present disclosure.
7B illustrates an example of an operating method for determining filter coefficients at a transmitting end of a wireless communication system according to various embodiments of the present disclosure.
8 illustrates an operating method for determining a circular convolution filter in a wireless communication system according to various embodiments of the present disclosure.
9 illustrates an example of a configuration of a communication unit of a receiving end of a wireless communication system according to various embodiments of the present disclosure.
10 illustrates another example of a configuration of a communication unit of a receiving end of a wireless communication system according to various embodiments of the present disclosure.
11A illustrates an example of a process of applying a WLMMSE matrix at a receiving end of a wireless communication system according to various embodiments of the present disclosure.
11B illustrates an example of a process of applying reverse symbol phase rotation at a receiving end of a wireless communication system according to various embodiments of the present disclosure.
12 illustrates a method of operating a transmitter for reducing PAPR in a wireless communication system according to various embodiments of the present disclosure.
13 illustrates a method of operating a receiver for reducing PAPR in a wireless communication system according to various embodiments of the present disclosure.
14 illustrates PAPR performance when there is no allocation of additional frequency resources in a wireless communication system according to various embodiments of the present disclosure.
15 illustrates PAPR performance when an additional frequency resource is allocated in a wireless communication system according to various embodiments of the present disclosure.
16 illustrates PAPR performance when a circular convolution filter is used in a wireless communication system according to various embodiments of the present disclosure.
17 illustrates another method of operating a transmitter for reducing PAPR in a wireless communication system according to various embodiments of the present disclosure.
18 illustrates another method of operation of a receiving end for PAPR reduction in a wireless communication system according to various embodiments of the present disclosure.

Terms used in the present disclosure are only used to describe a specific embodiment, and may not be intended to limit the scope of other embodiments. Singular expressions may include plural expressions unless the context clearly indicates otherwise. Terms used herein, including technical or scientific terms, may have the same meaning as commonly understood by a person of ordinary skill in the technical field described in the present disclosure. Among the terms used in the present disclosure, terms defined in a general dictionary may be interpreted as having the same or similar meanings as those in the context of the related technology, and unless explicitly defined in the present disclosure, an ideal or excessively formal meaning Is not interpreted as. In some cases, even terms defined in the present disclosure cannot be interpreted to exclude embodiments of the present disclosure.

In various embodiments of the present disclosure described below, a hardware approach is described as an example. However, since various embodiments of the present disclosure include technology using both hardware and software, various embodiments of the present disclosure do not exclude a software-based approach.

Hereinafter, the present disclosure relates to an apparatus and method for transmitting and receiving signals in a wireless communication system. Specifically, the present disclosure describes various embodiments for reducing a peak-to-average power ratio (PAPR) in a wireless communication system.

A term referring to network entities (e.g., a transmitting end, a receiving end), a term referring to a signal processing means (e.g., a filter), and a term referring to a component of a device (e.g.: Communication unit, control unit) and the like are illustrated for convenience of description. Accordingly, the present disclosure is not limited to terms to be described later, and other terms having an equivalent technical meaning may be used.

1 illustrates a wireless communication system according to various embodiments of the present disclosure. 1 is a part of nodes using a radio channel in a wireless communication system, and illustrates a transmitting end 110 and a receiving end 120. Referring to FIG. 1 illustrates one transmitting terminal 110 and one receiving terminal 120, but may include a plurality of transmitting terminals or a plurality of receiving terminals. Also, for convenience of explanation, in this document, the transmitting end 110 and the receiving end 120 are described as being separate entities, but the functions of the transmitting end 110 and the receiving end 120 may be interchanged with each other. For example, in the uplink of a cellular communication system, the transmitting end 110 may be a terminal, the receiving end 120 may be a base station, and in the downlink, the transmitting end 110 may be a base station, and the receiving end 120 may be a terminal. The PAPR reduction technique according to various embodiments of the present disclosure may be applied not only to uplink but also to downlink.

The transmitter 110 and the receiver 120 may transmit and receive radio signals in a millimeter wave (mmWave) band (eg, 28 GHz, 30 GHz, 38 GHz, 60 GHz). In this case, in order to improve the channel gain, the transmitting terminal 110 and the receiving terminal 120 may perform beamforming. Here, beamforming includes transmission beamforming and reception beamforming. That is, the transmitting end 110 and the receiving end 120 may impart directivity to a transmit signal or a receive signal. To this end, the transmitter 110 and the receiver 120 may select at least one serving beam through a beam search procedure. According to various embodiments of the present disclosure, excellent PAPR reduction performance may be required due to high frequency characteristics in a millimeter wave band.

In addition, a wireless communication system according to various embodiments of the present disclosure may provide a service requiring access to a large-scale terminal. Here, the service may be referred to as an mMTC (massive machine type communication) service. For example, the mMTC service can be used for Internet of Things (IoT) technology. The mMTC service may require wide coverage in order to provide a service to a large terminal. According to various embodiments of the present disclosure, the transmitting terminal 110 and the receiving terminal 120 are devices for an mMTC service, and excellent PAPR reduction performance may be required due to miniaturization or limitation of a battery.

PAPR is an important measure to determine the driving power of a mobile device. When PAPR is high, serious distortion occurs when passing through nonlinear devices such as ADC (analog-to-digital converter), DAC (digital-to-analog converter), and power amplifier. Therefore, high-performance devices with high linearity are required, resulting in higher cost. In the case of using a field effect transistor (FET) series device such as a complementary metal oxide semiconductor (CMOS), even if a high-performance device satisfies linearity, power consumption increases because the power supply voltage must be sufficiently high. Therefore, if the PAPR of the transmission signal is high, there is a high possibility of causing a big problem in the uplink where power and battery are limited, and especially in the mMTC environment of 5G (5th generation) communication where a very large number of low-power devices are expected to be connected. It is essential to lower the PAPR. In 4G (4th generation) communication (e.g., Long Term Evolution (LTE)), a discrete Fourier transform (DFT)-S-OFDM (DFT-spread-OFDM) having a single carrier property is used instead of orthogonal frequency division multiplexing (OFDM) with high PAPR. As the next-generation mobile communication requires wider coverage, technologies to improve the performance of cell edge users through power boost are being considered. Considering the performance and amplifier performance, it is not enough to boost the power. In the disclosure, the DFT-S-OFDM is modified to lower the PAPR without additional frequency resources, and the receiver structure is changed to minimize the reception performance degradation.

According to various embodiments of the present disclosure, PAPR can be reduced by designing both a transmitting end and a receiving end. In particular, by designing a receiver (eg, a base station) that is relatively free in terms of complexity in uplink as well as a transmitter to reduce PAPR, it is possible to improve signal reception performance while reducing PAPR of a signal processed at the receiver. In addition, additional frequency resources are not required, and PAPR can be further reduced by allocating additional frequency resources when necessary. In addition, PAPR performance can be further reduced by slightly sacrificing reception performance if necessary.

In particular, according to various embodiments of the present disclosure, transmission and reception of a binary phase shift keying (BPSK) symbol having an improper property will be described as an example. However, the scope of the present disclosure is not limited to transmission and reception of BPSK symbols, and may be applied to transmission and reception of symbols having inappropriate properties or other types of symbols. When the pseudo covariance matrix of the symbol vector is zero matrix (0), the symbol vector is defined as being appropriate. If the pseudo covariance matrix of the symbol vector is a matrix other than the zero matrix, the symbol vector is defined as inappropriate. The PAPR reduction technique according to various embodiments of the present disclosure may convert a reception performance gain obtained by applying a widely linear receiver using the impropriety property of a BPSK symbol into a PAPR gain. In order to convert the reception performance gain into PAPR, the transmitting end or the receiving end according to various embodiments of the present disclosure may determine a transmitter filter and a phase rotation amount according to a situation (eg, an allocated frequency resource).

2 is a diagram illustrating a configuration of an apparatus in a wireless communication system according to various embodiments of the present disclosure. That is, the configuration illustrated in FIG. 2 may be understood as the configuration of the transmitting end 110 or the receiving end 120. Used below'… Wealth','… A term such as'group' refers to a unit that processes at least one function or operation, which may be implemented by hardware or software, or a combination of hardware and software.

Referring to FIG. 2, the device may include a control unit 210, a communication unit 220, and a storage unit 230.

The controller 210 may control overall operations of the device. For example, the controller 210 may transmit and receive signals through the communication unit 220. Also, the control unit 210 may write or read data in the storage unit 230. To this end, the controller 330 may include at least one processor or a micro processor, or may be a part of a processor. In addition, a part of the communication unit 220 and the control unit 210 may be referred to as a communication processor (CP). In particular, according to various embodiments, the controller 210 controls the communication unit 220 to perform an operation for reducing PAPR in modulation or demodulation of a signal. In other words, the control unit 210 may control the operation of each component included in the communication unit 220. For example, the controller 210 may control the transmitting end 110 or the receiving end 120 to perform operations according to various embodiments described later.

The communication unit 220 may perform functions for transmitting and receiving signals through a wireless channel. For example, the communication unit 220 may perform a conversion function between a baseband signal and a bit stream according to the physical layer standard of the system. For example, when transmitting data, the communication unit 220 may generate complex symbols by encoding and modulating a transmission bit stream. In addition, when receiving data, the communication unit 220 may restore a received bit stream through demodulation and decoding of the baseband signal. In addition, the communication unit 220 may up-convert the baseband signal into a radio frequency (RF) band signal, transmit through an antenna, and down-convert an RF band signal received through the antenna into a baseband signal.

To this end, the communication unit 220 may include a transmission filter, a reception filter, an amplifier, a mixer, an oscillator, a digital to analog convertor (DAC), an analog to digital convertor (ADC), and the like. In addition, the communication unit 220 may include a plurality of transmission/reception paths. Furthermore, the communication unit 220 may include at least one antenna array composed of a plurality of antenna elements. In terms of hardware, the communication unit 210 may be composed of a digital unit and an analog unit, and the analog unit is composed of a plurality of sub-units according to operation power, operation frequency, etc. Can be configured.

The communication unit 220 transmits and receives signals as described above. Accordingly, the communication unit 220 may be referred to as a'transmitting unit', a'receiving unit', or a'transmitting/receiving unit'. In addition, in the following description, transmission and reception performed through a wireless channel are used in a sense including the processing as described above is performed by the communication unit 220. In addition, the communication unit 220 may include a backhaul communication unit for communication with another network entity connected through a backhaul network.

The storage unit 230 may store data such as a basic program, an application program, and setting information for the operation of the device. The storage unit 230 may be formed of a volatile memory, a nonvolatile memory, or a combination of a volatile memory and a nonvolatile memory. In addition, the storage unit 230 may provide stored data according to the request of the control unit 210.

3 illustrates an example of a configuration of a communication unit of a transmitting terminal 110 in a wireless communication system according to various embodiments of the present disclosure. The configuration illustrated in FIG. 3 may be understood as a partial configuration of the communication unit 220. The communication unit of the transmitter 110 illustrated in FIG. 3 is a symbol phase rotation block 305, a discrete Fourier transform (DFT) block 310, a spectrum shaping block 315, an inverse DFT (IDFT) block 320, and a cyclic prefix (CP) insertion block. 325 may be included.

은 한 OFDM 심볼 내 송신할 데이터 심볼들의 개수, 즉, DFT 크기에 기반하여 결정될 수 있다. 위상 회전 블록 305에 의해 전송 심볼의 위상을 회전시킴으로써, 시간 상에서 동위상 보간 간섭 확률을 감소시킬 수 있다. 시간상에서 동위상 보간 간섭 확률이 감소됨에 따라, 송신될 신호의 PAPR이 저감될 수 있다. 도 3을 참고하면, L개의 데이터 심볼에 대하여

만큼의 위상 회전이 수행될 수 있다. 다시 말해, 위상 회전 블록 305는 L개의 데이터 심볼을 입력받고, 각 데이터 심볼의 위상을 변환하고, 위상이 변환된 L개의 데이터 심볼을 DFT 블록 310으로 출력할 수 있다.The symbol phase rotation block 305 may rotate the phase of a transmission symbol. According to various embodiments of the present disclosure, the amount of phase rotation

May be determined based on the number of data symbols to be transmitted in one OFDM symbol, that is, the DFT size. By rotating the phase of the transmission symbol by the phase rotation block 305, it is possible to reduce the probability of in-phase interpolation interference in time. As the probability of in-phase interpolation interference in time is reduced, the PAPR of the signal to be transmitted can be reduced. Referring to FIG. 3, for L data symbols

As many as phase rotation can be performed. In other words, the phase rotation block 305 may receive L data symbols, transform the phase of each data symbol, and output the L data symbols whose phase is transformed to the DFT block 310.

The DFT block 310 may convert data in the frequency domain by performing DFT on the data symbols provided from the phase rotation block 305. Referring to FIG. 3, the DFT block 310 may receive L data symbols from the phase rotation block 305, convert it into L frequency domain data, and output it to the spectrum shaping block 315.

The spectrum shaping block 315 may perform filtering on data in the frequency domain transformed by the DFT block 310. According to various embodiments of the present disclosure, the filter of the spectrum shaping block 315 may be configured to reduce the PAPR of a signal modulated by the transmitting terminal 110 while satisfying a certain reception performance of the receiving terminal 120. In other words, the spectrum shaping block 315 may receive data in the frequency domain from the DFT block 310, and may output data filtered using the filtering matrix to the IDFT block 320. FIG. 3 shows an example in which there is no allocation of additional frequency resources for PAPR reduction, and the spectrum shaping block 315 outputs L filtered data equal to the number L of input data. According to other embodiments of the present disclosure, when additional frequency resources are allocated for PAPR reduction, the spectrum shaping block 315 may convert L input data into K data greater than L.

In FIG. 3, the DFT block 310 and the spectrum shaping block 315 are described as separate blocks, but the DFT block 310 and the spectrum shaping block 315 may be configured as one block. For example, a DFT operation and filtering may be simultaneously performed on L pieces of data provided from the phase rotation block 305 using one filtering matrix. Here, the filtering may be defined as K-point circular filtering.

The IDFT block 320 may convert data provided from the spectrum shaping block 315 into the time domain. Referring to FIG. 3, the IDFT block 320 may be converted into N time domain data by applying _{an IDFT matrix W N} ^{H to data provided from the spectrum shaping block 315.} The time domain data transformed by the IDFT block 320 may be provided to the CP insertion block 325.

을 출력할 수 있다. 송신 신호 벡터

는 RF 모듈(미도시)에서 상향 변환되어 안테나를 통해 수신단 120으로 송신될 수 있다. The CP insertion block 325 may insert a CP into time domain data. Referring to FIG. 3, the CP insertion block 325 is a transmission signal vector having (N+N _{C ) entries by adding N C} CPs in addition to the N symbols converted to the time domain by IDFT 320.

Can be printed. Transmit signal vector

May be up-converted in an RF module (not shown) and transmitted to the receiving end 120 through an antenna.

According to various embodiments of the present disclosure, the phase rotation value and the filter coefficient may be determined based on the PAPR performance of the transmitter 110 and the reception performance of the receiver 120. More specifically, the phase rotation value and the filter coefficient may be determined as a value that satisfies an orthogonality condition of a transmission symbol while minimizing the probability of occurrence of a case in which the PAPR at the transmitting terminal 110 is greater than a predetermined value.

4 illustrates an example of input and output of a phase rotation block at a transmitter 110 of a wireless communication system according to various embodiments of the present disclosure. FIG. 4 shows an example of the operation of the phase rotation block 305 of FIG. 3.

은 심볼 위상 회전 동작 400를 통해 위상이 회전된 데이터 심볼 벡터

로 변환될 수 있다. 보다 구체적으로, 도 4에 도시된 바와 같이, 위상 회전 블록 305는 BPSK 데이터 심볼 벡터

을 입력 받고, 아래의 <수학식 1>을 통해 위상이 회전된 데이터 심볼 벡터

를 출력할 수 있다.Referring to FIG. 4, a BPSK data symbol vector consisting of L entries

Silver symbol phase rotated data symbol vector through phase rotation operation 400

Can be converted to More specifically, as shown in Figure 4, the phase rotation block 305 is a BPSK data symbol vector

Is input, and the phase is rotated through <Equation 1> below

Can be printed.

은 BPSK 데이터 심볼 벡터,

는 위상 회전량,

는 위상이 회전된 데이터 심볼 벡터이다. 여기서, 위상 회전 값

는 아래의 <수학식 2>와 같이 데이터 심볼의 수 L에 파라미터화(parameterized) 될 수 있다.In <Equation 1>,

Is a BPSK data symbol vector,

Is the amount of phase rotation,

Is a data symbol vector whose phase is rotated. Where, the phase rotation value

May be parameterized to the number L of data symbols as shown in Equation 2 below.

는 위상 회전량이고, L은 한 OFDM 심볼 내 송신할 데이터 심볼의 수이다. 다시 말해, 위상 회전 값

는 OFDM 내 송신할 데이터의 심볼의 수 L에 기초하여 결정될 수 있다.In <Equation 2>,

Is the amount of phase rotation, and L is the number of data symbols to be transmitted in one OFDM symbol. In other words, the phase rotation value

May be determined based on the number L of symbols of data to be transmitted in OFDM.

5A illustrates an example of input and output of a filter block at a transmitting end of a wireless communication system according to various embodiments of the present disclosure. 5A shows an example of operation of the spectral shaping block 315 of FIG. 3.

에 L-포인트 DFT 행렬

이 곱해짐으로써 주파수 도메인으로 변환된 데이터 벡터

이 스펙트럼 셰이핑 블록 315로 입력된다. 주파수 도메인으로 변환된 데이터 벡터

은 스펙트럼 스펙트럼 셰이핑 동작 500에 의해 스펙트럼 셰이핑 벡터

가 적용된 후, 출력된다. 스펙트럼 셰이핑 블록 315에 의해 출력되는 벡터는 아래의 <수학식 3>과 같이 결정될 수 있다.Referring to Figure 5a, the phase is rotated data symbol vector

L-point DFT matrix in

Data vector converted to frequency domain by multiplying by

This is fed into the spectral shaping block 315. Data vector converted to frequency domain

Spectral shaping vector by spectral spectral shaping motion 500

After is applied, it is output. The vector output by the spectrum shaping block 315 may be determined as shown in Equation 3 below.

는 주파수 도메인으로 변환된 데이터 벡터,

는 스펙트럼 셰이핑 벡터,

는 스펙트럼 셰이핑 벡터를 구성하는 스펙트럼 셰이핑 계수(spectrum shaping coefficient) 또는 필터 계수이다. 스펙트럼 셰이핑 벡터가 적용된 벡터

는 N-포인트 IDFT 블록(예: 도 3의 IDFT 블록 320)로 출력될 수 있다.In <Equation 3>,

Is the data vector transformed into the frequency domain,

Is the spectral shaping vector,

Is a spectrum shaping coefficient or filter coefficient constituting a spectral shaping vector. Vector with Spectral Shaping Vector Applied

May be output as an N-point IDFT block (eg, IDFT block 320 of FIG. 3).

또는

로 정의될 수 있다. 필터 행렬은 아래의 <수학식 4>와 같이 정의될 수 있다.In the embodiments of FIGS. 3 and 5A, it is described that the DFT matrix and the spectral shaping vector are separately applied, but according to another embodiment, a matrix in which the DFT matrix and the spectral shaping vector are combined may be applied. Here, the matrix in which the DFT matrix and the spectral shaping vector are combined is a filter matrix

or

Can be defined as The filter matrix may be defined as in Equation 4 below.

는 필터 계수,

는 주파수 인덱스,

는 심볼 인덱스, L은 한 OFDM 심볼 내 송신할 데이터 심볼의 수이다. 본 개시의 다양한 실시 예들에 따르면, 필터 행렬은 원형(circular) 필터의 형태로 구성될 수 있다. 필터 행렬을 구성하는 계수들

는 이하 도 5b 또는 도 5c와 같이 정의될 수 있다.In <Equation 4>,

Is the filter coefficient,

Is the frequency index,

Is the symbol index, and L is the number of data symbols to be transmitted in one OFDM symbol. According to various embodiments of the present disclosure, the filter matrix may be configured in the form of a circular filter. Coefficients that make up the filter matrix

May be defined as shown in FIG. 5B or 5C below.

의 절대값의 제곱에 해당한다. 도 5b의 예는 추가 주파수 자원의 할당이 없는 경우에 해당하므로(

), 데이터 심볼의 전송을 위해 할당된 주파수(K)는 OFDM 심볼 내 송신할 데이터 심볼의 수 L과 동일하다. 도 5b에서, 주파수 인덱스 i는 1 이상이고 OFDM 심볼 내 송신할 데이터 심볼의 수 L 이하의 정수일 수 있다.5B is a diagram illustrating an example of a filter for a case in which an additional frequency resource is not allocated at a transmitting end of a wireless communication system according to various embodiments of the present disclosure. In FIG. 5B, the horizontal axis is a normalized frequency value, and the vertical axis is a filter coefficient.

It corresponds to the square of the absolute value of Since the example of FIG. 5B corresponds to the case where there is no allocation of additional frequency resources (

), the frequency (K) allocated for transmission of data symbols is equal to the number L of data symbols to be transmitted in the OFDM symbol. In FIG. 5B, the frequency index i may be equal to or greater than 1 and may be an integer equal to or less than the number L of data symbols to be transmitted in the OFDM symbol.

의 절대값의 제곱에 해당한다. 도 5c의 예는 추가 주파수 자원의 할당이 존재하는 경우에 해당하므로(

), 데이터 심볼의 전송을 위해 할당된 주파수 K는 OFDM 심볼 내 송신할 데이터 심볼의 수 L 보다 크다. 도 5b에서, 주파수 인덱스 i는 1 이상이고 데이터 심볼의 전송을 위해 할당된 주파수 K 이하의 정수일 수 있다.5C is a diagram illustrating an example of a filter when an additional frequency resource is allocated at a transmitting end of a wireless communication system according to various embodiments of the present disclosure. In Fig. 5c, the horizontal axis is a normalized frequency value, and the vertical axis is a filter coefficient.

It corresponds to the square of the absolute value of Since the example of FIG. 5C corresponds to the case where the allocation of additional frequency resources exists (

), the frequency K allocated for data symbol transmission is greater than the number L of data symbols to be transmitted in the OFDM symbol. In FIG. 5B, the frequency index i may be equal to or greater than 1 and may be an integer equal to or less than the frequency K allocated for transmission of data symbols.

에 해당한다. 추가 주파수 자원이 없는 경우, 즉, 데이터 심볼의 전송을 위해 할당된 주파수 K가 OFDM 심볼 내 송신할 데이터 심볼의 수 L과 동일할 때(K=L), 필터 계수

는 아래의 <수학식 5>와 같이 정의될 있다. 6A illustrates an example of a polynomial for determining filter coefficients at a transmitting end of a wireless communication system according to various embodiments of the present disclosure. In Fig. 6A, the horizontal axis is a normalized frequency value, and the vertical axis is a filter coefficient.

Corresponds to. When there is no additional frequency resource, that is, when the frequency K allocated for data symbol transmission is equal to the number L of data symbols to be transmitted in the OFDM symbol (K=L), the filter coefficient

Can be defined as in <Equation 5> below.

는 아래 <수학식 6>과 같이 정의될 수 있다.In <Equation 5>, L corresponds to the number of data symbols to be transmitted in an OFDM symbol, and a polynomial for determining filter coefficients

Can be defined as in Equation 6 below.

일 수 있다.In <Equation 6>, the range of x is

Can be

는 송신 신호의 PAPR을 최소화하기 위한 조건과 수신단 120에서의 수신 성능을 보장하기 위한 조건을 만족하는 다항식일 수 있다. 또한,

는 송신 신호의 PAPR을 최소화하기 위한 조건과 수신단 120에서의 수신 성능을 보장하기 위한 조건을 만족하는 최적화된 필터 계수를 연속적인 함수로 근사화한 다항식일 수 있다. 보다 구체적으로,

는 PAPR이 일정한 값 이상인 신호가 발생할 확률을 최소화하면서, 특정 변조 방식(예: BPSK, PAM(pulse amplitude modulation))을 사용한 경우, 수신단 120의 수신기(예: WLMMSE(widely linear minimum mean square error) 수신기, MMSE 수신기, 또는 MF(matched filter) 수신기)에서 일정 성능 조건(예: MSE(mean square error)가 최소화되는 조건)을 만족하도록 설정된 최적의 함수일 수 있다. 다시 말해,

는 수신단 120의 수신 특성 및 수신 성능에 기반하여 결정된 함수일 수 있다.here,

May be a polynomial that satisfies the condition for minimizing the PAPR of the transmission signal and the condition for guaranteeing the reception performance at the receiver 120. Also,

May be a polynomial approximating an optimized filter coefficient that satisfies the condition for minimizing the PAPR of the transmission signal and the condition for ensuring the reception performance at the receiving end 120 with a continuous function. More specifically,

When using a specific modulation method (e.g. BPSK, PAM (pulse amplitude modulation)) while minimizing the probability of occurrence of a signal with a PAPR equal to or greater than a certain value, the receiver at the receiving end 120 (e.g., a widely linear minimum mean square error (WLMMSE) , MMSE receiver, or matched filter (MF) receiver) may be an optimal function set to satisfy a certain performance condition (eg, a condition in which a mean square error (MSE) is minimized). In other words,

May be a function determined based on reception characteristics and reception performance of the reception terminal 120.

로서 <수학식 6>과 같은 다항식을 이용하는 경우의 예를 설명하나, <수학식 6>과 유사한 다른 다항식이 이용될 수 있다. 즉, 필터 계수의 결정을 위한 다항식은, 송신 신호의 PAPR을 최소화하기 위한 조건과 수신단 120에서의 수신 성능을 보장하기 위한 조건을 만족하는 어느 다항식일 수 있다.The present disclosure is a polynomial equation for the determination of filter coefficients

An example of using a polynomial such as <Equation 6> will be described, but other polynomials similar to <Equation 6> may be used. That is, the polynomial for determining the filter coefficient may be any polynomial that satisfies a condition for minimizing PAPR of a transmission signal and a condition for guaranteeing reception performance at the receiving end 120.

는 <수학식 6>의 다항식

의 샘플링 값으로 정의될 수 있으며, <수학식 5>의 필터 계수의 결정을 위한 다항식

의 샘플링 인덱스

는 아래의 <수학식 7>과 같이 정의될 수 있다.That is, the filter coefficient of <Equation 5>

Is the polynomial in Equation 6

It can be defined as a sampling value of, and a polynomial for determining the filter coefficient of Equation 5

Sampling index of

Can be defined as in Equation 7 below.

는 필터 계수의 결정을 위한 다항식

의 샘플링 인덱스, L은 OFDM 심볼 내 송신할 데이터 심볼의 수에 해당한다. 즉, 다항식

의 샘플링 인덱스는 OFDM 심볼 내 송신할 데이터 심볼의 수 L에 기초하여 결정될 수 있다.In <Equation 7>,

Is a polynomial for the determination of filter coefficients

Sampling index, L, corresponds to the number of data symbols to be transmitted in the OFDM symbol. I.e. polynomial

The sampling index of may be determined based on the number L of data symbols to be transmitted in the OFDM symbol.

및 샘플링된 필터 계수의 제곱

에 해당한다. 도 6b에서 연속적인 실선은 필터 계수의 결정을 위한 다항식

에 해당하며, 다항식

에 대하여 샘플링된 불연속적인 값이 필터 계수의 제곱

에 해당한다. 6B illustrates an example of filter coefficient sampling when the number of data symbols is 12 at the transmitting end of the wireless communication system according to various embodiments of the present disclosure. In the graph of FIG. 6B, the horizontal axis represents a normalized frequency value, and the vertical axis represents a polynomial for determining filter coefficients.

And the square of the sampled filter coefficients

Corresponds to. In Figure 6b, the continuous solid line is a polynomial for the determination of the filter coefficient.

Corresponds to, and polynomial

The discrete values sampled for are the square of the filter coefficients

Corresponds to.

의 샘플링 인덱스

는 아래의 <수학식 8>과 같이 정의될 수 있다.The graph of FIG. 6B shows that when there is no additional frequency resource, that is, when the frequency K allocated for data symbol transmission is equal to the number L of data symbols to be transmitted in the OFDM symbol (K=L), transmission in the OFDM symbol An example in which the number L of data symbols to be performed is 12 is shown. That is, since K=L=12, the polynomial for the determination of the filter coefficient

Sampling index of

Can be defined as in Equation 8 below.

를 샘플링 함으로써, 아래 <수학식 9>와 같이 필터 계수가 결정될 수 있다.As shown in Equation 8 above, a polynomial for determining the filter coefficient

By sampling, the filter coefficient can be determined as shown in Equation 9 below.

는 필터 계수, i는 주파수 인덱스에 해당한다. 도 6b 및 <수학식 9>는 주파수 인덱스 i가 1에서 6인 경우의 필터 계수를 나타내며, 나머지 주파수 인덱스가 7 내지 12인 경우의 필터 계수는 도 6b의 필터 계수와 y축에 대하여 대칭되는 그래프와 같이 설정될 수 있다. In <Equation 9>,

Is the filter coefficient and i is the frequency index. 6B and <Equation 9> show filter coefficients when the frequency index i is 1 to 6, and the filter coefficients when the remaining frequency indexes are 7 to 12 are graphs symmetrical with respect to the filter coefficients of FIG. 6B and the y-axis. It can be set like this.

및 샘플링된 필터 계수

에 해당한다. 도 6c에서 연속적인 실선은 필터 계수의 결정을 위한 다항식

에 해당하며, 다항식

에 대하여 샘플링된 불연속적인 값이 필터 계수

에 해당한다. 6C illustrates an example of filter coefficient sampling when the number of symbols is 32 at the transmitting end of the wireless communication system according to various embodiments of the present disclosure. In the graph of Fig. 6c, the horizontal axis represents a normalized frequency value, and the vertical axis represents a polynomial for determining filter coefficients.

And sampled filter coefficients

Corresponds to. In Fig. 6c, a continuous solid line is a polynomial for the determination of filter coefficients.

Corresponds to, and polynomial

The discrete values sampled for are the filter coefficients

Corresponds to.

의 샘플링 인덱스

는 아래의 <수학식 10>과 같이 정의될 수 있다.The graph of FIG. 6D shows that when there is no additional frequency resource, that is, when the frequency K allocated for data symbol transmission is equal to the number L of data symbols to be transmitted in the OFDM symbol (K=L), transmission in the OFDM symbol An example in which the number L of data symbols to be performed is 32 is shown. That is, since K=L=32, the polynomial for the determination of the filter coefficient

Sampling index of

Can be defined as in Equation 10 below.

를 샘플링 함으로써, 아래 <수학식 11>과 같이 필터 계수가 결정될 수 있다.As shown in Equation 10 above, a polynomial for determining the filter coefficient

By sampling, the filter coefficient can be determined as shown in Equation 11 below.

는 필터 계수곱, i는 주파수 인덱스에 해당한다. 도 6c 및 <수학식 11>은 주파수 인덱스 i가 1에서 16인 경우의 필터 계수를 나타내며, 나머지 주파수 인덱스가 17 내지 32인 경우의 필터 계수는 도 6c의 필터 계수와 y축에 대하여 대칭되는 그래프와 같이 설정될 수 있다. In <Equation 11>,

Is the filter coefficient product, and i is the frequency index. 6C and <Equation 11> show filter coefficients when the frequency index i is 1 to 16, and the filter coefficients when the remaining frequency indexes are 17 to 32 are graphs symmetrical with respect to the filter coefficients and y-axis of FIG. 6C It can be set like this.

7A illustrates an operating method for determining a spectrum shaping vector at a transmitting end of a wireless communication system according to various embodiments of the present disclosure. 7A illustrates a method of operating a transmitter 110 (eg, a spectrum shaping block).

Referring to FIG. 7A, in step 701, the transmitter 110 may check the number of data symbols to be transmitted in the OFDM symbol. According to various embodiments of the present disclosure, a data symbol transmitted by the transmitting terminal 110 may be a symbol having improper characteristics, such as a BPSK data symbol.

In step 703, a sampling index may be determined. For example, the transmitter 110 may determine a polynomial sampling index for determining filter coefficients based on the number of data symbols to be transmitted in the OFDM symbol.

In step 705, the transmitting end may perform sampling. For example, the transmitter may calculate a value obtained by sampling a polynomial for determining a filter coefficient according to a sampling index determined based on the number of data symbols. In various embodiments of the present disclosure, since the polynomial for determining the filter coefficient is symmetric with respect to the vertical axis, a value obtained by sampling the polynomial by a number corresponding to half of the number of data symbols to be transmitted in the OFDM symbol is determined as a coefficient, Values corresponding to the remaining clauses may be determined as symmetrical values.

In step 707, the transmitter 110 may determine a vector based on the sampling values. Here, a vector based on sampling values may be referred to as a'sampling vector'. For example, the transmitter 110 may determine a vector composed of the square root of the sampling values determined in step 705.

In step 709, the transmitter 110 may determine a spectrum shaping vector. For example, the spectral shaping vector may be determined by normalizing the vector determined in step 707.

7B illustrates an example of an operating method for determining a spectrum shaping vector at a transmitting end of a wireless communication system according to various embodiments of the present disclosure. 7B illustrates a method of operating a transmitter 110 (eg, a spectrum shaping block).

Referring to FIG. 7B, in step 711, the transmitter 110 may check the number L of data symbols to be transmitted in the OFDM symbol.

의 샘플링 인덱스

를 결정할 수 있다. 즉, 샘플링 인덱스

는 아래의 <수학식 12>와 같이 결정될 수 있다.In step 713, the transmitting terminal 110 uses a polynomial for determining filter coefficients based on the number L of data symbols to be transmitted in the OFDM symbol.

Sampling index of

Can be determined. I.e. sampling index

Can be determined as in Equation 12 below.

는

의 샘플링 인덱스, i는 주파수의 인덱스, L은 OFDM 심볼 내 송신할 데이터 심볼의 수에 해당한다.In <Equation 12>,

Is

A sampling index of, i is a frequency index, and L corresponds to the number of data symbols to be transmitted in an OFDM symbol.

In step 715, the transmitting end has a coefficient |ci| ² can be determined. The transmitter is the sampling value |ci| ² can be determined. More specifically, the sampling value |ci| ² can be determined as in Equation 13 below.

는 다항식

를

에 대하여 샘플링한 값, i는 주파수의 인덱스, L은 OFDM 심볼 내 송신할 데이터 심볼의 수에 해당한다. 다시 말해, OFDM 심볼 내 송신할 데이터 심볼의 수 L의 절반에 해당하는 L/2만큼 다항식

를 샘플링한 값을 결정하고, 나머지 절반 L/2에 해당하는 계수는 i=L/2 축에 대하여 대칭되는 값으로 결정될 수 있다.In <Equation 13>,

Is a polynomial

To

A value sampled for, i is a frequency index, and L corresponds to the number of data symbols to be transmitted in an OFDM symbol. In other words, a polynomial as much as L/2 corresponding to half of the number L of data symbols to be transmitted in an OFDM symbol

A value sampled of is determined, and a coefficient corresponding to the other half L/2 may be determined as a value symmetrical with respect to the i=L/2 axis.

를 결정할 수 있다. 보다 구체적으로 715 단계에서 결정된 L개의 샘플링 값들의 제곱근들로 구성된 샘플링 벡터

를 결정할 수 있다. 샘플링 벡터

는 아래의 <수학식 14>와 같이 결정될 수 있다.In step 717, the transmitter 110 is a sampling vector based on the filter coefficient.

Can be determined. More specifically, a sampling vector consisting of square roots of the L sampling values determined in step 715

Can be determined. Sampling vector

Can be determined as in Equation 14 below.

는 샘플링 값들의 제곱근들로 구성된 샘플링 벡터,

은 벡터

의 원소들, L은 OFDM 심볼 내 송신할 데이터 심볼의 수에 해당한다.In <Equation 14>,

Is a sampling vector consisting of the square roots of the sampled values,

Silver vector

The elements, L, correspond to the number of data symbols to be transmitted in the OFDM symbol.

를 결정할 수 있다. 보다 구체적으로, 스펙트럼 셰이핑 벡터

는 717 단계에서 결정된 샘플링 벡터

를 정규화함으로써 결정될 수 있다. 즉, 스펙트럼 셰이핑 벡터

는 아래의 <수학식 15>와 같이 결정될 수 있다.In step 719, the transmitting end 110 is a spectrum shaping vector

Can be determined. More specifically, the spectral shaping vector

Is the sampling vector determined in step 717

Can be determined by normalizing In other words, the spectral shaping vector

Can be determined as shown in Equation 15 below.

는 계수 벡터,

는

의 2-놈(norm) 값에 해당한다. 다시 말해, 스펙트럼 셰이핑 벡터

는 다항식

를 OFDM 심볼 내 송신할 데이터 심볼의 수 L만큼 샘플링 한 값에 기초하여 결정될 수 있다.In <Equation 15>,

Is the coefficient vector,

Is

Corresponds to the 2-norm value of. In other words, the spectral shaping vector

Is a polynomial

May be determined based on a value sampled by the number L of data symbols to be transmitted in the OFDM symbol.

8 illustrates an operating method for determining a circular convolution filter in a wireless communication system according to various embodiments of the present disclosure. 8 illustrates an operation method of the transmitting terminal 110.

Referring to FIG. 8, in step 801, the transmitter 110 may determine a spectrum shaping vector. In some embodiments, the spectral shaping vector can be determined by normalizing the coefficient vector of the filter.

In step 803, the transmitter 110 may determine a circular convolution filter based on the spectral shaping vector. For example, the circular convolution filter may be determined as shown in Equation 16 below. That is, the spectral shaping vector may be transformed into a circular convolution filter through <Equation 16>.

행렬의 푸리에 변환인

는 항상 순환 행렬(circulant matrix)로 하기 <수학식 17>과 같은 형태를 가질 수 있다. 즉, 순환 행렬은 데이터의 심볼의 수 L에 기반하여 결정될 수 있다.<Equation 16> can be derived from <Equation 3>. At this time

Fourier transform of matrix

Is always a circulant matrix and may have a form as shown in Equation 17 below. That is, the cyclic matrix may be determined based on the number L of symbols of data.

부분은 데이터 시퀀스 벡터(data sequence vector)

을

를 사용해 원형 컨벌루션(circular convolution)하는 동작(operation)으로 볼 수 있다. Therefore, in Equation 16

Part is a data sequence vector

of

It can be viewed as an operation that performs circular convolution by using.

를 대신 고려할 수도 있으며 필터 탭을 0이 아닌 컨벌루션 필터 계수들(nonzero convolution filter coefficients) 수로 정의하면 각 필터 탭 별로 아래와 같은 값들을 적용해 PAPR을 낮출 수 있으며 PAPR 값은 탭 수에 반비례하여 감소한다. 일부 실시 예들에서, 필터 탭은 수신단 120의 수신 성능 및 전력에 대한 정보에 기반하여 결정될 수 있다. 예를 들어, 전력에 대한 정보는 PAPR 값을 의미할 수 있다. 다른 실시 예들에서, L 탭 컨벌루션 필터(tap convolutional filter)를 사용하는 경우 성능이 가장 좋다. 여기서, L 탭 컨벌루션 필터는 L 개의 0이 아닌 컨벌루션 필터 계수들을 포함하는 원형 컨벌루션 필터를 의미할 수 있다. 이 방식의 경우 앞선 다항식(polynomial) 방식과 다르게 특별히 필터를 함수로부터 구하는 과정 없이 L 값에 따라 원형 컨벌루션 필터 계수들(circular convolution filter coefficients)에 0을 패딩(padding) 함으로써 확장 가능하다. 즉, 상기 <수학식 17>의 순환 행렬이 LxL 행렬인 경우, L보다 작은 K 탭 컨벌루션 필터의 원형 컨벌루션 필터 계수들에 L-K 개의 0을 패딩함으로써 확장 가능할 수 있다.Therefore, circular convolution filters instead of spectral shaping vectors for low PAPR

If the filter tap is defined as the number of nonzero convolution filter coefficients instead, PAPR can be lowered by applying the following values for each filter tap, and the PAPR value decreases in inverse proportion to the number of taps. In some embodiments, the filter tap may be determined based on information on reception performance and power of the receiving terminal 120. For example, information on power may mean a PAPR value. In other embodiments, performance is best when an L tap convolutional filter is used. Here, the L tap convolution filter may mean a circular convolution filter including L non-zero convolution filter coefficients. Unlike the previous polynomial method, this method can be extended by padding circular convolution filter coefficients with zeros according to the L value without a special process of obtaining a filter from a function. That is, when the cyclic matrix in Equation 17 is an LxL matrix, it can be extended by padding LK zeros in circular convolution filter coefficients of a K-tap convolution filter smaller than L.

For example, a 3 tap circular convolution filter may be expressed as Equation 18 below.

For another example, a 4 tap circular convolution filter may be expressed as Equation 19 below.

As another example, a K-tap circular convolution filter may be expressed as Equation 20 below.

When K <L, a circular convolution filter may be generated for any K value as described above. And the following is a table showing the values of the circular convolutional filter for reducing PAPR by taps up to 7 taps. For example, the filter value is not unique and a circular shift of the values below in Table 1 below. It can be determined using the value specified.

3tap -0.6972 0.7167 -0.0161 0 0 0 0 4tap -0.1185 -0.35 0.8699 -0.3267 0 0 0 5tap -0.0608 -0.3413 0.8714 -0.3417 -0.0613 0 0 6tap -0.0688 -0.347 0.8765 -0.3225 -0.0512 -0.0074 0 7tap -0.0651 -0.3409 0.8793 -0.3218 -0.0527 -0.0073 -0.0034

You can also use these values to find an equivalent spectral shaping vector.

According to an embodiment, the information on Table 1 may be stored in the transmitter 110. In some embodiments, in order to reduce PAPR, the transmitter 110 adaptively selects one of 3tap, 4tap, 5tap, 6tap, and 7tap in Table 1 to determine a circular convolution filter.

In other embodiments, one circular convolutional filter for a specific filter tap may be stored in the transmitter 110 and the receiver 120 according to the PAPR value and the reception performance required by the system. That is, a circular convolutional filter for a specific filter tap determined in the system may be shared between the transmitting terminal 110 and the receiving terminal 120 and used.

9 illustrates an example of a configuration of a communication unit of a reception terminal 120 of a wireless communication system according to various embodiments of the present disclosure. The configuration illustrated in FIG. 9 may be understood as a partial configuration of the communication unit 220.

Referring to FIG. 9, the communication unit of the receiving terminal 120 may include a CP removal block 910, an IDFT block 915, and a channel equalizer 920.

는 채널을 통해 수신단 120에서 수신된다. 수신된 신호의 벡터

는 N개의 데이터 심볼에 더하여 N_C개의 CP를 포함할 수 있다.Signal transmitted from transmitter 110

Is received at the receiving end 120 through the channel. Vector of received signal

_{May include N C} CPs in addition to N data symbols.

에서 CP를 제거할 수 있다. 예를 들어, CP 제거 블록 910은 수신된 신호에서 CP가 차지하는 심볼의 데이터를 제거할 수 있다. 다시 말해, CP 제거 블록 910은 (N+N_C)개의 엔트리로 구성된 수신된 신호의 벡터

에서 N_C개의 CP를 제거할 수 있다. 즉, CP 제거 블록 910은 (N+N_C)개의 엔트리를 포함하는 수신된 신호의 벡터

를 입력받고, N개의 엔트리를 포함하는 시간 도메인의 심볼 벡터를 IDFT 블록 915로 출력할 수 있다.CP removal block 910 is a vector of the received signal

CP can be removed from. For example, the CP removal block 910 may remove data of a symbol occupied by a CP from a received signal. In other words, the CP removal block 910 is a vector of a received signal consisting _{of (N+N C) entries.}

It is possible to remove _{N C CPs.} That is, the CP removal block 910 is a vector of a received signal including _{(N+N C) entries}

Is received, and a symbol vector in a time domain including N entries may be output as an IDFT block 915.

The IDFT block 915 may convert data in the time domain provided from the CP removal block 910 into data in the frequency domain. For example, the IDFT block 915 may generate frequency domain data mapped to N subcarriers by applying an IDFT matrix to a vector including N data symbols. The vector transformed by the IDFT block 915 may be output to the channel equalizer 920.

를 출력할 수 있다.The channel equalizer 920 may equalize data converted to the frequency domain. For example, the channel equalizer 920 may compensate for distortion of data generated by the channel. Also, the channel equalizer 920 may extract L pieces of data from among N subcarriers. In other words, the channel equalizer 920 is a vector including data allocated to L subcarriers.

Can be printed.

10 illustrates an example of another configuration of a communication unit of a receiving end of a wireless communication system according to various embodiments of the present disclosure. The configuration illustrated in FIG. 10 may be understood as a partial configuration of the communication unit 220. The configuration of FIG. 10 may be connected to the configuration of FIG. 9. For example, it may be connected to the equalizer 920 of FIG. 9, the conjugate block 1010 of FIG. 10, and the WLMMSE block 1020 of FIG.

Referring to FIG. 10, the receiver 120 of FIG. 10 may include a conjugation (*) block 1010, a WLMMSE block 1020, and a phase reverse rotation block 1025.

의 값들에 대한 켤례 값을 생성하고, 생성된 켤례 값들의 벡터

를 WLMMSE 블록 1020으로 출력할 수 있다.The conjugate block 1010 is a vector input from a channel equalizer (eg, channel equalizer 920 in FIG. 9).

Create conjugate values for the values of, and a vector of generated conjugate values

Can be output to the WLMMSE block 1020.

및 켤례 블록 1010으로부터 입력된 벡터

에 대하여 WLMMSE 행렬을 적용함으로써, 송신단 110에서 수행된 스펙트럼 셰이핑에 의한 데이터의 왜곡을 보상할 수 있다. 상세한 WLMMSE 블록의 동작은 아래 도 11a를 참고하여 설명된다.The WLMMSE block 1020 is a vector input from the channel equalizer 920 of FIG. 9

And vector input from conjugate block 1010

By applying the WLMMSE matrix to, it is possible to compensate for distortion of data due to spectrum shaping performed at the transmitter 110. The detailed operation of the WLMMSE block will be described with reference to FIG. 11A below.

11A illustrates an example of a process of applying a WLMMSE matrix at a receiving end of a wireless communication system according to various embodiments of the present disclosure. 11A shows an example of the operation of the WLMMSE block 1020 of FIG. 10.

및 켤례 블록(예: 도 10의 켤례 블록 1010)에 의해 생성된

를 서브 벡터로서 포함하는

크기의 벡터가 WLMMSE 블록 1020로 입력된다. WLMMSE 블록 1020는 입력된 벡터에 대하여 WLMMSE 행렬을 적용하여

크기의 벡터를 출력할 수 있다. 도 11a의 WLMMSE 블록 1020는 도 10의 WLMMSE 블록 1020과 동일하거나 유사한 구성일 수 있다.Referring to FIG. 11A, a vector provided from a channel equalizer (eg, a channel equalizer 920 in FIG. 9)

And the conjugate block (e.g., the conjugate block 1010 in FIG. 10).

Containing as subvectors

A vector of size is input to the WLMMSE block 1020. The WLMMSE block 1020 applies the WLMMSE matrix to the input vector.

You can output a vector of size. The WLMMSE block 1020 of FIG. 11A may have the same or similar configuration to the WLMMSE block 1020 of FIG. 10.

According to various embodiments of the present disclosure, the WLMMSE matrix may be defined as shown in Equation 21 below.

는 송신단 110에서 이용되는 원형 필터 행렬로서, 아래의 <수학식 22>와 같이 정의될 수 있으며,

는 아래의 <수학식 23>과 같이 정의될 수 있다. 또한, <수학식 21>의

는 아래의 <수학식 24>와 같이 정의될 수 있다. 또한, <수학식 21>에서

은 잡음 공분산 행렬(noise covariance matrix)에 해당하며,

는 잡음 의사 공분산 행렬(noise pseudo covariance matrix)에 해당한다.In <Equation 21>,

Is a circular filter matrix used in the transmitter 110, and can be defined as in Equation 22 below,

Can be defined as in Equation 23 below. Also, in <Equation 21>

Can be defined as in Equation 24 below. Also, in <Equation 21>

Corresponds to the noise covariance matrix,

Corresponds to the noise pseudo covariance matrix.

는 스펙트럼 셰이핑 벡터,

은 L-포인트 DFT 행렬이고, L은 OFDM 심볼 내 송신할 데이터 심볼의 수에 해당한다.In <Equation 22>,

Is the spectral shaping vector,

Is an L-point DFT matrix, and L corresponds to the number of data symbols to be transmitted in the OFDM symbol.

은 BPSK 데이터 심볼 벡터,

은 위상이 회전된 데이터 심볼 벡터, L은 OFDM 심볼 내 송신할 데이터 심볼의 수,

는 위상 회전 값에 해당한다. <수학식 23>에서,

는 데이터 심볼이 부적절한 성질을 가질 때 영행렬이 아닌 행렬(non zero matrix)로 구성될 수 있다.In <Equation 23>,

Is a BPSK data symbol vector,

Is the phase-rotated data symbol vector, L is the number of data symbols to be transmitted in the OFDM symbol,

Corresponds to the phase rotation value. In <Equation 23>,

May be configured as a non-zero matrix when the data symbol has inappropriate properties.

는 송신단 110에서 이용되는 원형 필터 행렬,

은 N-포인트 IDFT 행렬, N은 OFDM IDFT 크기, K는 수신단 120에 할당된 부반송파의 수, L은 OFDM 심볼 내 송신할 데이터 심볼의 수에 해당한다.In <Equation 24>,

Is the circular filter matrix used at the transmitter 110,

Is an N-point IDFT matrix, N is an OFDM IDFT size, K is the number of subcarriers allocated to the receiving end 120, and L is the number of data symbols to be transmitted in the OFDM symbol.

를 샘플링하고, 샘플링에 의해 결정된 필터 계수들을 포함하는 필터 행렬

를 이용하여 WLMMSE 행렬을 결정할 수 있다. 다시 말해, WLMMSE 행렬은, 송신 신호의 PAPR을 감소시키기 위한 조건과 수신단 120의 수신 성능을 만족시키기 위한 조건을 만족하는 송신단의 필터 행렬

에 기초하여 결정될 수 있다.According to various embodiments of the present disclosure, the receiving end 120 is a function according to the number K of subcarriers allocated to the receiving end 120.

And a filter matrix including filter coefficients determined by sampling

Can be used to determine the WLMMSE matrix. In other words, the WLMMSE matrix is a filter matrix of a transmitter that satisfies the condition for reducing the PAPR of the transmission signal and the condition for satisfying the reception performance of the receiver 120

It can be determined based on

11B illustrates an example of a process of applying reverse symbol phase rotation at a receiving end of a wireless communication system according to various embodiments of the present disclosure. 11B shows an example of the operation of the phase reverse rotation block 1025 of FIG. 10.

크기의 벡터가 심볼 위상 역회전 블록 1025으로 입력될 수 있다. 심볼 위상 역회전 블록 1025은 입력된 벡터의 데이터의 위상을 변경할 수 있다. 보다 구체적으로, 심볼 위상 역회전 블록 1025은 WLMMSE 블록으로부터 제공된 벡터에 위상 역회전 벡터를 적용함으로써, OFDM 데이터 심볼

을 복원할 수 있다. 본 개시의 다양한 실시 예들에 따르면, 수신단 120은 수신단 120에 할당된 서브 캐리어의 수 L에 기초하여 결정되는 위상 역회전 값

에 따라 WLMMSE 블록을 통해 출력된 벡터의 위상을 변경할 수 있다.Referring to FIG. 11B, the output from the WLMMSE block (eg, WLMMSE block 1020 of FIG. 11A)

A vector of magnitude may be input to the symbol phase inverse rotation block 1025. The symbol phase inverse rotation block 1025 may change the phase of the data of the input vector. More specifically, the symbol phase inverse rotation block 1025 applies a phase inverse rotation vector to a vector provided from the WLMMSE block, thereby forming an OFDM data symbol.

Can be restored. According to various embodiments of the present disclosure, the reception terminal 120 is a phase reverse rotation value determined based on the number L of subcarriers allocated to the reception terminal 120.

According to this, the phase of the vector output through the WLMMSE block can be changed.

12 illustrates a method of operating a transmitter 110 for PAPR reduction in a wireless communication system according to various embodiments of the present disclosure. 11 illustrates an operation method of a transmitter 110.

Referring to FIG. 12, in step 1201, a transmitting terminal 110 may check resources allocated to transmit data to a receiving terminal 120. For example, in the uplink, the transmitting terminal 110 corresponding to the terminal may check a certain number of frequency resources (eg, the number of subcarriers) allocated to transmit one OFDM symbol to the receiving terminal 120. In another example, in the downlink, the transmitting terminal 110 corresponding to the base station may allocate allocated frequency resources to transmit data to the receiving terminal 120 corresponding to the terminal.

According to various embodiments of the present disclosure, the transmitter 110 may not allocate additional frequency resources, but may allocate the number of frequency resources equal to the number of symbols of transmission data. Even when no additional frequency resources are allocated, the transmission/reception terminal 110 according to various embodiments of the present disclosure applies a widely linear receiver that uses an inappropriate characteristic of the BPSK symbol to gain the reception performance of the reception terminal 120. Can be converted to PAPR gain.

According to other embodiments of the present disclosure, the transmitting terminal 110 may allocate a frequency resource larger than the number of transmission data symbols. That is, the transmitter 110 may allocate additional frequency resources. By allocating additional frequency resources, the transmitter 110 may convert the gain of the frequency resource into an additional gain for reducing PAPR.

In step 1203, the transmitter 110 may check the amount of phase rotation and the filter coefficient. According to various embodiments of the present disclosure, the amount of phase rotation and the filter coefficient may be determined based on the number of frequency resources allocated to transmit one OFDM symbol to a receiver. In addition, the filter coefficient may be set to satisfy a condition for minimizing the PAPR of a transmission signal at the transmitting terminal 110 and a condition for satisfying the reception performance of the receiving terminal 120. For example, the filter coefficient may be determined as a value obtained by sampling a polynomial set to satisfy a condition for reducing PAPR and a condition for satisfying reception performance at the receiving end 120 according to an allocated frequency resource.

In step 1205, the transmitter 110 may perform phase rotation on the data. For example, the transmitter 110 may change the phase of the data according to the amount of phase rotation determined based on the number of frequency resources allocated for the data to be transmitted. By changing the phase of the data, it is possible to reduce the PAPR by reducing the probability of constructive interference of the data in the time domain.

In step 1207, the transmitter 110 may perform DFT on data in the time domain whose phase is rotated. For example, the transmitter 110 may convert the data into the frequency domain by applying the DFT matrix to the data in the time domain whose phase is rotated.

In step 1209, the transmitter 110 may perform filtering on data in the frequency domain. For example, the transmitter 110 may apply a filter matrix including the filter coefficients determined in step 1203 to data in the frequency domain. When the additional frequency resource is not allocated, the size of the filter matrix (the number of rows and the number of columns) may be set equal to the number of transmission data symbols. When additional frequency resources are allocated, the number of rows of the filter matrix may be set larger than the number of columns, and accordingly, the size of the vector to which the filter matrix is applied may be greater than the number of data symbols.

According to various embodiments of the present disclosure, steps 1207 and 1209 may be performed as one step. In other words, the filter matrix in step 1209 may be a matrix composed of a product of a filter coefficient and a value for transformation into a frequency domain-in.

In step 1211, the transmitter 110 may map the filtered data in the frequency domain to the subcarrier.

In step 1213, the transmitter 110 may perform IDFT on data in the frequency domain mapped to the subcarrier. For example, the transmitter 110 may convert data in the frequency domain into data in the time domain by applying the IDFT matrix to the data in the frequency domain mapped to the subcarrier.

In step 1215, the transmitter 110 may transmit a signal including time domain data. For example, the transmitter 110 may transmit a signal obtained by up-converting data in the time domain to the receiver 120 through an antenna. Since the time domain data has been subjected to phase conversion and filtering, the up-converted signal may have a low PAPR.

13 illustrates a method of operating a receiver for reducing PAPR in a wireless communication system according to various embodiments of the present disclosure. 13 illustrates an operation method of the receiving terminal 120.

Referring to FIG. 13, in step 1301, the receiving end 120 may check the allocated resource. For example, when the receiving end 120 is a base station, the receiving end 120 may check the number of frequency resources (eg, subcarriers) in order to receive an uplink signal from the transmitting end 110 corresponding to the terminal. As another example, when the receiving end 120 is a terminal, the receiving end 120 may check the number of frequency resources (eg, subcarriers) allocated to the receiving end 120 based on control information received from the transmitting end 110 through a control channel.

In step 1303, the receiving end 120 may check the amount of phase rotation and the filter coefficient. According to various embodiments of the present disclosure, the amount of phase rotation or the filter coefficient may be determined based on the number of allocated frequency resources and the number of data symbols. The number of allocated frequency resources may be confirmed through a control signal in step 1201. According to various embodiments of the present disclosure, the number of data symbols may be the same as the number of allocated frequency resources. That is, additional frequency resources may not be allocated for transmission of data symbols. According to other embodiments of the present disclosure, the number of data symbols may be smaller than the number of allocated frequency resources. That is, additional frequency resources larger than the number of data symbols may be allocated. In this case, the receiving terminal 110 may check the number of data symbols through the control signal transmitted through the control channel. In addition, the receiving end 110 may check the number of data symbols according to the channel state with the transmitting end 120. For example, the receiver 110 may index the number of data symbols at the modulation and coding scheme (MCS) level. The receiver 120 may check the amount of phase rotation based on the number of allocated frequency resources and the number of data symbols. Also, the receiver 120 may check the filter coefficients based on the number of allocated frequency resources and the number of data symbols. According to various embodiments of the present disclosure, the amount of phase rotation or the filter coefficient may be predefined according to the number of allocated frequency resources and the number of data symbols. For example, the amount of phase rotation or filter coefficient is tabled according to the number of allocated frequency resources and the number of data symbols and stored in advance in the transmitting terminal 110 or the receiving terminal 120, or shared through a control channel between the transmitting terminal 110 and the receiving terminal 120. I can. Also, the amount of phase rotation or the filter coefficient may be indexed and set with the MCS level.

In step 1305, the receiving end 120 may calculate the WLMMSE matrix. The WLMMSE matrix according to various embodiments of the present disclosure may be set to satisfy a condition for minimizing PAPR of a transmission signal and a condition for optimizing reception performance of a received signal (minimizing MSE of a received signal). The WLMMSE matrix may be determined based on the filter coefficients identified in step 1201. More specifically, the WLMMSE matrix may be set based on a filter matrix including filter coefficients determined in consideration of receiver characteristics and reception performance of the receiving end.

In step 1307, the receiving terminal 120 may apply the WLMMSE matrix to the received data. The receiver 120 may apply the WLMMSE matrix to data converted from the received signal data into the frequency domain. Prior to the application of the WLMMSE matrix, data converted to the frequency domain may be compensated for distortion caused by a channel through a channel equalizer, or some data may be extracted according to the number of data symbols. The receiver 120 may apply the WLMMSE matrix to L pieces of data that have passed through the channel equalizer. By applying the WLMMSE matrix, data for which distortion of a signal due to filtering at the transmitter 110 is compensated may be output. In particular, the WLMMSE matrix may use an inappropriate property of the BPSK symbol.

In step 1309, the receiving terminal 120 may apply phase reverse rotation. For example, the receiving end 120 may rotate the phase of the data in the opposite direction by the amount of phase rotation determined in step 1303. By performing the phase reverse rotation, it is possible to compensate for the phase rotation performed by the transmitter 110.

In step 1311, the receiving end 120 may demodulate the signal. For example, the receiving end 120 may decode the received data and transmit an acknowledgment (ACK) or a negative ACK (NACK) signal to the transmitting end 110.

14 illustrates PAPR performance when there is no allocation of additional frequency resources in a wireless communication system according to various embodiments of the present disclosure. In the graph of FIG. 14, the horizontal axis represents the PAPR of the signal, and the vertical axis represents the probability of occurrence of a PAPR greater than a certain value. The graph of FIG. 14 is a simulation result for the case where the number L of BPSK data symbols is 12 (L=12), the number of allocated frequency resources K is 12 (K=12), and the OFDM IDFT size is 128 (N=128). Represents. Referring to FIG. 14, when the probability of occurrence of PAPR is 10 ^-3 than a certain value, PAPR to which the PAPR reduction technique according to various embodiments of the present disclosure is applied is 3.75 dB than when the phase rotation technique is applied to DFT-S-OFDM. It can be seen that the PAPR gain of

15 illustrates PAPR performance when an additional frequency resource is allocated in a wireless communication system according to various embodiments of the present disclosure. In the graph of FIG. 15, the horizontal axis represents the PAPR of the signal, and the vertical axis represents the probability of occurrence of a PAPR greater than a certain value. The graph of FIG. 15 is a simulation result for the case where the number L of BPSK data symbols is 12 (L=12), the number of allocated frequency resources K is 16 (K=16), and the OFDM IDFT size is 128 (N=128). Represents. Referring to FIG. 15, when the probability of occurrence of PAPR is 10 ^-3 than a predetermined value, the PAPR to which the PAPR reduction technique according to various embodiments of the present disclosure is applied is applied to the DFT-S-OFDM with a phase rotation technique (root-raised (RRC)). It can be seen that 1.8dB of PAPR gain occurs than when -cosine) shaping) is applied.

16 illustrates PAPR performance when a circular convolution filter is used in a wireless communication system according to various embodiments of the present disclosure. In the graph of FIG. 16, the horizontal axis represents the PAPR of the signal, and the vertical axis represents the probability of occurrence of a PAPR greater than a certain value. The graph of FIG. 16 shows simulation results for the case of a 3-tap circular convolution filter, a 4-tap circular convolution filter, a 7-tap circular convolution filter, and an L-tap circular convolution filter. Referring to FIG. 16, when the probability of occurrence of PAPR is 10 ^-3 than a predetermined value, it can be seen that a smaller PAPR gain occurs as the number of taps increases.

17 is a diagram illustrating another method of operation of a transmitter 110 for PAPR reduction in a wireless communication system according to various embodiments of the present disclosure.

Referring to FIG. 17, in step 1701, the transmitter 110 may apply a filter to data. In some embodiments, the filter may be determined based on receiver characteristics and reception performance of the receiving terminal 120. In other embodiments, when the number of at least one subcarrier is equal to the number of at least one symbol included in data, the filter may have a value obtained by sampling a predetermined polynomial by the number of at least one symbol as a coefficient. When the number of at least one subcarrier is greater than the number of at least one symbol, the filter may have a value obtained by sampling a preset polynomial by the number of at least one subcarrier as a coefficient. In still other embodiments, the transmitter 110 may change at least one phase according to the amount of phase rotation of at least one symbol included in the data, and filter data including at least one symbol whose phase is changed using a filter. have. In still other embodiments, when the number of at least one subcarrier is equal to the number of at least one symbol, the amount of phase rotation may be determined based on the number of at least one symbol. When the number of at least one subcarrier is greater than the number of at least one symbol, the amount of phase rotation may be determined based on the number of at least one symbol and the number of at least one subcarrier.

In step 1703, the transmitter 110 may map the data to which the filter is applied to at least one subcarrier. In some embodiments, the filter may include a circular convolution filter that is determined based on the number of at least one symbol included in the data.

In step 1705, the transmitting terminal 110 may transmit the mapped data to the receiving terminal 120.

FIG. 18 is a diagram illustrating another method of operating a receiver 120 for PAPR reduction in a wireless communication system according to various embodiments of the present disclosure.

Referring to FIG. 18, in step 1801, the reception terminal 120 may receive data from the transmission terminal 110.

In step 1803, the receiving terminal 120 may apply a transformation matrix to the data. In some embodiments, the transformation matrix may be determined based on a filter shared by the transmitting terminal 110 and the receiving terminal 120. Here, the filter may be determined based on receiver characteristics and reception performance of the receiver 120. In other embodiments, when the number of at least one subcarrier allocated to the transmitting terminal 110 is the same as the number of at least one symbol included in the data, the filter samples a preset polynomial by the number of at least one symbol as a coefficient. I can have it. When the number of at least one subcarrier is greater than the number of at least one symbol, the filter may have a value obtained by sampling a preset polynomial by the number of at least one subcarrier as a coefficient. In still other embodiments, the filter may include a circular convolution filter that is determined based on the number of at least one symbol included in data.

In step 1805, the reception terminal 120 may decode data to which the transformation matrix is applied. In some embodiments, the receiving terminal 120 may change the phase of at least one symbol according to the amount of phase rotation of the at least one symbol and decode at least one symbol whose phase is changed. In other embodiments, when the number of at least one subcarrier allocated to the transmitting terminal 110 is the same as the number of at least one symbol, the amount of phase rotation may be determined based on the number of at least one symbol. When the number of at least one subcarrier is greater than the number of at least one symbol, the amount of phase rotation may be determined based on the number of at least one symbol and the number of at least one subcarrier.

The methods according to the embodiments described in the claims or the specification of the present disclosure may be implemented in the form of hardware, software, or a combination of hardware and software.

When implemented in software, a computer-readable storage medium storing one or more programs (software modules) may be provided. One or more programs stored in a computer-readable storage medium are configured to be executable by one or more processors in an electronic device (device). The one or more programs include instructions that cause the electronic device to execute methods according to embodiments described in the claims or specification of the present disclosure.

These programs (software modules, software) include random access memory, non-volatile memory including flash memory, read only memory (ROM), and electrically erasable programmable ROM. (electrically erasable programmable read only memory, EEPROM), magnetic disc storage device, compact disc-ROM (CD-ROM), digital versatile discs (DVDs) or other forms of It may be stored in an optical storage device or a magnetic cassette. Alternatively, it may be stored in a memory composed of a combination of some or all of them. In addition, a plurality of configuration memories may be included.

In addition, the program is provided through a communication network such as the Internet, Intranet, a local area network (LAN), a wide area network (WAN), or a storage area network (SAN), or a combination thereof. It may be stored in an accessible storage device. Such a storage device may access a device performing an embodiment of the present disclosure through an external port. In addition, a separate storage device on the communication network may access a device performing an embodiment of the present disclosure.

In the above-described specific embodiments of the present disclosure, components included in the disclosure are expressed in the singular or plural according to the presented specific embodiments. However, the singular or plural expression is selected appropriately for the situation presented for convenience of description, and the present disclosure is not limited to the singular or plural constituent elements, and even constituent elements expressed in plural are composed of the singular or Even the expressed constituent elements may be composed of pluralities.

Meanwhile, although specific embodiments have been described in the detailed description of the present disclosure, various modifications may be made without departing from the scope of the present disclosure. Therefore, the scope of the present disclosure is limited to the described embodiments and should not be defined, and should be determined by the scope of the claims and equivalents as well as the scope of the claims to be described later.

Claims (24)

In the method of operating a transmitting end in a wireless communication system,
The process of applying filters to the data,
A process of mapping the filter-applied data to at least one subcarrier,
Including the process of transmitting a signal including the mapped data to a receiving end,
The filter uses a matrix including at least one coefficient,
When the number of the at least one subcarrier is the same as the number of at least one symbol included in the data, the at least one coefficient is at least one obtained by sampling a polynomial the same number of times as the number of the at least one symbol The method is determined based on the value of.
The method according to claim 1,
When the number of the at least one subcarrier is greater than the number of the at least one symbol, the at least one coefficient is based on at least one value obtained by sampling the polynomial the same number of times as the number of the at least one subcarrier. How it is determined.
The method according to claim 1,
The process of applying a filter to the data,
A process of changing the phase of the at least one symbol according to the amount of phase rotation of the at least one symbol,
And filtering the data including the at least one symbol whose phase is changed using the filter.
The method of claim 3,
When the number of the at least one subcarrier is equal to the number of the at least one symbol, the amount of phase rotation is determined based on the number of the at least one symbol,
When the number of the at least one subcarrier is greater than the number of the at least one symbol, the amount of phase rotation is determined based on the number of the at least one symbol and the number of the at least one subcarrier.
The method according to claim 1,
The filter includes a circular convolution filter that is determined based on the number of the at least one symbol.
In the operating method of the receiving end in a wireless communication system,
The process of receiving a signal including data from the transmitting end, and
A process of applying a transformation matrix to the data, and
Including the process of decoding the data to which the transformation matrix is applied,
The transformation matrix is determined based on a filter shared by the transmitting end and the receiving end,
The filter uses a matrix including at least one coefficient,
When the number of at least one subcarrier is the same as the number of at least one symbol included in the data, the at least one coefficient is at least one obtained by sampling a polynomial a number of times equal to the number of the at least one symbol. How it is determined based on the value.
The method of claim 6,
When the number of the at least one subcarrier is greater than the number of the at least one symbol, the at least one coefficient is based on at least one value obtained by sampling the polynomial the same number of times as the number of the at least one subcarrier. How it is determined.
The method of claim 6,
The process of decoding the data,
A process of changing the phase of the at least one symbol according to the amount of phase rotation of the at least one symbol,
And decoding the at least one symbol whose phase has been changed.
The method of claim 8,
When the number of the at least one subcarrier is equal to the number of the at least one symbol, the amount of phase rotation is determined based on the number of the at least one symbol,
When the number of the at least one subcarrier is greater than the number of the at least one symbol, the amount of phase rotation is determined based on the number of the at least one symbol and the number of the at least one subcarrier.
The method of claim 6,
The filter includes a circular convolution filter that is determined based on the number of the at least one symbol.
In the apparatus of the transmitting end in a wireless communication system,
A transmitting and receiving unit,
Including at least one processor coupled to the transmitting and receiving unit,
The at least one processor applies a filter to data, maps the filtered data to at least one subcarrier,
The transmitting and receiving unit transmits a signal including the mapped data to a receiving end,
The filter uses a matrix including at least one coefficient, and when the number of the at least one subcarrier is the same as the number of at least one symbol included in the data, the at least one coefficient is the An apparatus that is determined based on at least one value obtained by sampling a polynomial a number of times equal to the number of at least one symbol.
The method of claim 11,
When the number of the at least one subcarrier is greater than the number of the at least one symbol, the at least one coefficient is based on at least one value obtained by sampling the polynomial the same number of times as the number of the at least one subcarrier. Device determined.
The method of claim 11,
The at least one processor,
Changing the phase of the at least one symbol according to the amount of phase rotation of the at least one symbol,
An apparatus for filtering the data including the at least one symbol whose phase is changed using the filter.
The method of claim 13,
When the number of the at least one subcarrier is equal to the number of the at least one symbol, the amount of phase rotation is determined based on the number of the at least one symbol,
When the number of the at least one subcarrier is greater than the number of the at least one symbol, the amount of phase rotation is determined based on the number of the at least one symbol and the number of the at least one subcarrier.
The method of claim 11,
The filter includes a circular convolution filter that is determined based on the number of the at least one symbol.
In the device of the receiving end in a wireless communication system,
A transmitting and receiving unit,
Including at least one processor coupled to the transmitting and receiving unit,
The transmitting and receiving unit receives a signal including data from a transmitting end,
The at least one processor applies a transformation matrix to the data, decodes the data to which the transformation matrix is applied,
The transformation matrix is determined based on a filter shared by the transmitting end and the receiving end,
The filter uses a matrix including at least one coefficient,
When the number of at least one subcarrier is the same as the number of at least one symbol included in the data, the at least one coefficient is at least one obtained by sampling a polynomial the same number of times as the number of the at least one symbol. Device determined based on value.
The method of claim 16,
When the number of the at least one subcarrier is greater than the number of the at least one symbol, the at least one coefficient is based on at least one value obtained by sampling the polynomial the same number of times as the number of the at least one subcarrier. Device determined.
The method of claim 16,
The at least one processor,
Changing the phase of the at least one symbol according to the amount of phase rotation of the at least one symbol,
An apparatus for decoding the at least one symbol whose phase has been changed.
The method of claim 18,
When the number of the at least one subcarrier is equal to the number of the at least one symbol, the amount of phase rotation is determined based on the number of the at least one symbol,
When the number of the at least one subcarrier is greater than the number of the at least one symbol, the amount of phase rotation is determined based on the number of the at least one symbol and the number of the at least one subcarrier.
The method of claim 16,
The filter includes a circular convolution filter that is determined based on the number of the at least one symbol.
The method according to claim 1,
The polynomial is determined based on at least one of a peak-to-average power ratio (PAPR) of the transmitted signal or a reception performance of the receiving end.
The method of claim 6,
The polynomial is determined based on at least one of a peak-to-average power ratio (PAPR) of the transmitted signal or a reception performance of the receiving end.
The method of claim 11,
The polynomial is determined based on at least one of a peak-to-average power ratio (PAPR) of the transmitted signal or a reception performance of the receiving end.
The method of claim 16,
The polynomial is determined based on at least one of a peak-to-average power ratio (PAPR) of the transmitted signal or a reception performance of the receiving end.

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