KR20000068674A - Quadrature frequency division multiplexing demodulator - Google Patents

Abstract

The differential detection circuit 26 differentially detects the output of the FFT circuit 25 from symbol to symbol, and the correlation calculation circuit 27 calculates a correlation value between the differential detection output and the placement information of the subcarrier for pilot transmission. The frequency error calculating circuit 28 calculates the frequency error of the subcarrier interval unit from the peak position of the correlation value, and the carrier frequency correction circuit 23 corrects the carrier frequency by using this. In addition, the phase averaging circuit 29 averages the phases of the differential detection outputs corresponding to the pilot carrier subcarriers, and using this, the phase variation correction circuit 30 corrects the phase variation common to the subcarriers. As a result, carrier frequency synchronization with a shorter infeed time is realized, and the influence of phase variations common to all subcarriers due to the phase noise of the tuner is eliminated.

Description

Quadrature Frequency Division Multiple Signal Demodulation Device {QUADRATURE FREQUENCY DIVISION MULTIPLEXING DEMODULATOR}

In recent years, orthogonal frequency division multiplex (OFDM) transmission schemes have attracted attention in mobile voice-oriented digital voice broadcasting and terrestrial digital television broadcasting.

This OFDM transmission method is a method of modulating a plurality of subcarriers orthogonal to each other by digital data to be transmitted, and multiplexing the modulated waves to transmit them. This method has a feature that when the number of subcarriers used increases from hundreds to thousands, the symbol period of each modulated wave becomes very long, and thus it is difficult to be affected by multipath interference.

Hereinafter, the principle of the OFDM transmission method will be described with reference to FIG.

1 is a block diagram showing the principle configuration of an OFDM transmission scheme. In addition, in FIG. 1, the thick solid line arrow shows a complex signal, and the thin solid line arrow shows a real signal.

First, on the transmitting side, the signal to be transmitted is a data signal input to the OFDM signal modulation device 11 is mapped by a mapping circuit 111 to a signal point on a complex plane according to the modulation scheme of each subcarrier, Inverse Fast Fourier Transform (IFFT) circuit 112 is supplied. The IFFT circuit 112 IFFT-processes the transmitted signal for one symbol and converts it into the time domain to generate a valid symbol period signal, and for each symbol, the rear part of the valid symbol period signal is used as a guide period signal as a valid symbol. By adding in front of the period signal, it has a function of generating an OFDM signal of a base band. The generated baseband OFDM signal is supplied to the quadrature modulation circuit 113. The orthogonal modulation circuit 113 performs frequency conversion of the face band OFDM signal into a signal of an intermediate frequency (hereinafter, referred to as IF (Intermediate Frequency)) band by orthogonally modulating a carrier with a baseband OFDM signal. The signal is converted into a signal in a radio frequency (hereinafter referred to as RF (Radio Frequency)) band by the up converter 114 and output to the transmission path 12.

On the other hand, the OFDM signal input from the transmission path 12 to the OFDM demodulation device 13 on the receiving side is frequency-converted from the RF band to the IF band by the tuner 131 and then supplied to the orthogonal demodulation circuit 132. . The orthogonal demodulation circuit 132 demodulates the input IF band signal into a baseband OFDM signal by orthogonal demodulation, and the demodulation output is supplied to a Fourier transform (FFT (Fast Fourier Transform)) circuit 133. . The FFT circuit 133 outputs an effective symbol period signal from the baseband OFDM signal, converts it into a frequency domain by performing FFT processing, and the output is supplied to the detection circuit 134. The detection circuit 134 detects each subcarrier according to a modulation method, and then demaps the data signal.

However, with the above-described principle configuration, when there is an error between frequencies of carriers used in transmission and reception, data cannot be demodulated correctly. Therefore, conventionally, a method of obtaining a wide range of frequency synchronization by combining two automatic frequency control (hereinafter referred to as AFC (Auto Frequency Control)) circuits within a subcarrier interval and a subcarrier interval unit has been disclosed (for example, in 1996, electronic information). Prospective Proceedings of the Communications Division of the Korean Institute of Communication Sciences, B-512, page 512).

In the AFC method disclosed in the above document, the frequency error within the subcarrier interval is calculated using the correlation therebetween because the guide period signal in the OFDM signal is a rear copy of the effective symbol period signal. The frequency error in the subcarrier interval unit is calculated by using the reference symbols for frequency synchronization inserted at predetermined intervals by the transmitting side.

Hereinafter, the structure and operation of the conventional OFDM signal demodulation apparatus using the AFC method disclosed in the above document will be described with reference to FIGS. 2 and 3.

2 is a schematic diagram illustrating an example of a configuration of a frequency synchronization reference symbol. In Fig. 2, the horizontal axis represents frequency and the vertical axis represents amplitude, the solid line in the figure indicates that a subcarrier exists at that frequency, and the broken line indicates that no subcarrier exists at that frequency. In this example, the presence or absence of a subcarrier is associated with a predetermined pseudo random (hereinafter referred to as PN (Pseudo Noise) series).

3 is a block diagram showing the structure of a conventional OFDM signal demodulation device. In FIG. 3, the thick solid line arrow represents a complex signal, and the thin solid line arrow represents a real signal. In addition, general control signals, such as a clock required for the operation of each component, are omitted so as not to be complicated.

In Fig. 3, the tuner 21 frequency-converts the OFDM signal input from the transmission path from the RF band to the IF band, and its output is supplied to the orthogonal demodulation circuit 22. The orthogonal demodulation circuit 22 demodulates the IF signal of the IF band into a baseband OFDM signal by using a fixed carrier generated therein, and the demodulation output of the carrier frequency (fc) correction circuit 23 It is supplied to the first input terminal. The carrier frequency correction circuit 23 corrects based on a wideband carrier frequency error signal in units of subcarrier intervals supplied to the second input terminal and a narrowband carrier frequency error signal within subcarrier intervals supplied to the third input terminal. The carrier frequency error is corrected by multiplying the carrier by the baseband OFDM signal supplied to the first input terminal, and its output is supplied to the narrowband carrier frequency error calculating circuit 24 and the FFT circuit 25.

The narrowband carrier frequency error calculating circuit 24 calculates the frequency error within the subcarrier interval by using the correlation between the guide period signal in the baseband OFDM signal and the rear end of the effective symbol period signal, the output of which is the carrier frequency. It is supplied to the third input terminal of the correction circuit 23. The FFT circuit 25 converts the effective symbol period signal in the baseband OFDM signal into a frequency domain by FFT processing, and the output thereof is supplied to the power calculation circuit 41 and the detection circuit 31.

The power calculation circuit 41 calculates the power of a signal corresponding to each subcarrier output from the FFT circuit 25, and the calculation result is supplied to the correlation calculation circuit 42. The correlation calculating circuit 42 calculates a correlation value between the output of the power calculating circuit 41 and the PN sequence corresponding to the presence or absence of a subcarrier of the frequency synchronization reference symbol shown in FIG. The carrier frequency error calculation circuit 28 is supplied. The wideband carrier frequency error calculating circuit 28 calculates a frequency error in units of subcarrier intervals from the peak position of the correlation value, and its output is supplied to the second input terminal of the carrier frequency correction circuit 23. The detection circuit 31 recovers the data signal by detecting and then demapping each subcarrier according to a modulation scheme.

The present invention relates to an orthogonal frequency division multiple signal demodulation device used for digital broadcasting or digital communication using an orthogonal frequency division multiplexing transmission method. A technique for removing the effects of phase shift common to all subcarriers.

1 is a block diagram showing an example of a principle configuration of an OFDM transmission method.

2 is a schematic diagram showing a configuration example of a frequency synchronization reference symbol according to a conventional OFDM signal demodulation device.

3 is a block diagram showing a configuration example of a conventional OFDM signal demodulation device.

4 is a schematic diagram illustrating an example of pilot signal arrangement according to the present invention.

Fig. 5 is a block diagram showing the structure of an OFDM signal demodulation device according to the first embodiment of the present invention.

FIG. 6 is a block diagram illustrating an internal configuration example of a differential detection circuit in FIG. 5.

FIG. 7 is a block diagram illustrating a first internal configuration example of the correlation calculation circuit in FIG. 5.

FIG. 8 is a block diagram showing a second internal configuration example of the correlation calculation circuit in FIG. 5.

FIG. 9 is a block diagram illustrating an internal configuration example of an inter-symbol filter circuit in FIG. 8.

FIG. 10 is a block diagram illustrating a third internal configuration example of the correlation calculation circuit in FIG. 5.

FIG. 11 is a block diagram illustrating a fourth internal configuration example of the correlation calculation circuit in FIG. 5.

FIG. 12 is a block diagram illustrating an internal configuration example of a phase shift correction circuit in FIG. 5.

Fig. 13 is a block diagram showing the construction of an OFDM signal demodulation device according to a second embodiment of the present invention.

Fig. 14 is a block diagram showing the construction of an OFDM signal demodulation device according to a third embodiment of the present invention.

Fig. 15 is a block diagram showing the construction of an OFDM signal demodulation device according to a fourth embodiment of the present invention.

Fig. 16 is a block diagram showing the construction of an OFDM signal demodulation device according to a fifth embodiment of the present invention.

17 is a block diagram illustrating a first internal configuration example of a detection circuit in FIG. 16.

FIG. 18 is a block diagram illustrating a second internal configuration example of the detection circuit in FIG. 16.

Fig. 19 is a block diagram showing the construction of an OFDM signal demodulation device according to a sixth embodiment of the present invention.

FIG. 20 is a block diagram illustrating an internal configuration example of a correlation calculation circuit in FIG. 19.

Fig. 21 is a block diagram showing the construction of an OFDMf signal demodulation device according to a seventh embodiment of the present invention.

Fig. 22 is a block diagram showing the construction of the signal demodulation device according to the eighth embodiment of the present invention.

(Initiation of invention)

However, in the above-described conventional method, since the frequency error in units of subcarrier intervals is calculated using the reference symbols for frequency synchronization inserted at predetermined intervals (for example, frames) at the transmitting side, the frequency synchronization is pulled in. The time becomes relatively long.

In addition, in the conventional method, in order to reduce the error of the carrier frequency in the steady state, for example, the time constant of the loop filter provided inside the narrowband carrier frequency error calculating circuit 24 of FIG. You need to set it. Therefore, fast fluctuations such as the phase noise of the tuner cannot be followed (not limited to this example, and can not follow fast fluctuations of the tuner's phase noise and the like even in a general AFC circuit). For this reason, the residual frequency error causes interference between subcarriers (hereinafter referred to as Inter Carrier Interference) and phase shift common to all subcarriers (hereinafter referred to as CPE (Common Phase Error)), which is a factor of error rate deterioration.

Accordingly, an object of the present invention is to provide an OFDM signal demodulation device capable of solving the above problem and shortening the induction time of frequency synchronization and eliminating the influence of CPE due to the phase noise of a tuner.

In order to solve the above problems, the OFDM transmission method according to the present invention is configured as follows.

(1) an apparatus for demodulating an orthogonal frequency division multiplex signal comprising a first pilot signal arranged at a synchronous frequency of every symbol, said apparatus comprising: Fourier transform means for converting said orthogonal frequency division multiplex signal into a frequency axis signal by Fourier transforming; Differential detection means for calculating the variation between symbols by differentially detecting the output of the Fourier transform means between symbols, correlation calculation means for calculating a correlation between the arrangement information of the first pilot signal and the output of the differential detection means; Wideband carrier frequency error calculating means for estimating a carrier frequency error in units of subcarrier spacing by detecting a peak position of the output of said correlation calculating means, and a wideband for correcting a carrier frequency based on an output of said wideband carrier frequency error calculating means; It comprises a carrier frequency correction means.

(2) an apparatus for demodulating an orthogonal frequency division multiplex signal comprising a first pilot signal arranged at a synchronous frequency of every symbol, said apparatus comprising: Fourier transform means for converting said orthogonal frequency division multiplex signal into a frequency axis signal by Fourier transforming; Differential detection means for calculating the variation between symbols by differentially detecting the output of the Fourier transform means and all subcarriers by averaging the phases of the outputs of the differential detection means corresponding to the first pilot signal within symbols. A phase averaging means for estimating a phase variation common to the phase shift correction means for calculating a correction vector of every symbol from the output of the phase averaging means and correcting a phase variation common to all subcarriers based on the correction vector; It is provided.

(3) an apparatus for demodulating an orthogonal frequency division multiplex signal comprising a first pilot signal arranged at a synchronous frequency of every symbol, said apparatus comprising: Fourier transform means for converting said orthogonal frequency division multiplex signal into a frequency axis signal by Fourier transforming; Differential detection means for calculating variation between symbols by differentially detecting the output of said Fourier transform means, correlation calculation means for calculating a correlation between the arrangement information of said first pilot signal and the output of said differential detection means; A wideband carrier frequency error calculating means for estimating a carrier frequency error in units of subcarrier spacing by detecting a peak position of the output of the correlation calculating means, and a wideband carrier for correcting a carrier frequency based on an output of the wideband carrier frequency error calculating means Frequency correction means and the first pilot signal A phase averaging means for estimating a phase variation common to all subcarriers by averaging the phase of the output of the differential detection means in a symbol, and calculating a correction vector of every symbol from the output of the phase averaging means and And phase shift correction means for correcting the phase shift common to all the subcarriers based on this.

(4) In the configuration of (1) and (3), the correlation calculating means determines the correlation magnitude between the arrangement information (two-value signal) of the first pilot signal and the output (complex vector signal) of the differential detection means. It is a structure to calculate.

(5) In the configurations (1) and (3), the correlation calculating means is a signal (complex signal) obtained by averaging the arrangement information (two value signal) of the first pilot signal and the output of the differential detection means in the symbol direction. Is calculated to calculate the correlation size with

(6) In the configuration of (1) and (3), the correlation calculating means is the magnitude of the signal obtained by averaging the arrangement information (two-value signal) of the first pilot signal and the output of the differential detection means in the symbol direction ( Real signal).

(7) In the configurations (1) and (3), the correlation calculating means measures the magnitude of the signal obtained by averaging the arrangement information (two-value signal) of the first pilot signal and the output of the differential detection means in the symbol direction. It is a structure which calculates the correlation with the signal (two-value signal) which binarized compared with the predetermined threshold value.

(8) In the configuration of (7), the correlation calculating means is configured to control the threshold value by the magnitude of the received signal.

(9) In the configurations (1) and (3), the broadband carrier frequency correction means is configured to control the local oscillation frequency of the tuner based on the output of the broadband carrier frequency error calculating means.

(10) In the configurations (1) and (3), the broadband carrier frequency correction means is configured to control the local oscillation frequency of the orthogonal demodulation means based on the output of the broadband carrier frequency error calculating means.

(11) In the configurations (1) and (3), the broadband carrier frequency correction means generates a correction carrier based on the output of the broadband carrier frequency error calculating means, and inputs the correction carrier to the Fourier transform means. It is a structure which multiplies a signal.

(12) In the configuration of (1), the broadband carrier frequency correction means shifts the output signal of the Fourier transform means in the frequency direction based on the output of the broadband carrier frequency error calculating means, and depends on the guide period length. It is configured to correct phase rotation between symbols generated.

(13) In the configuration (3), the broadband carrier frequency correction means is configured to shift the output signal of the Fourier transform means in the frequency direction based on the output of the broadband carrier frequency error calculating means.

(14) In the configurations of (2) and (3), the phase shift correction means is incorporated into the detection means, and the detection means detects the phase shift common to all subcarriers based on the output of the correction vector calculation means. At the same time, it is configured to detect in accordance with the primary modulation method of each subcarrier.

(15) In the configuration of (14), in addition to the first pilot signal, an apparatus for demodulating an orthogonal frequency division multiplex signal for transmitting a second pilot signal distributed and periodically arranged in a subcarrier-symbol region. The detecting means is configured to correct phase variations common to all subcarriers based on the output of the correction vector calculating means, and to synchronously detect each subcarrier using the second pilot signal.

(16) In the configuration of (14), the detection means in the apparatus for demodulating orthogonal frequency division multiplex signals for differentially modulating and transmitting data signals between symbols is based on the output of the correction vector calculating means. The phase variation common to the circuits is corrected, and each subcarrier is differentially detected between symbols.

(17) In the structures (2) and (3), the phase averaging means averages the output complex vector of the differential detection means corresponding to the first pilot signal in a symbol and calculates the phases of all subcarriers. It is a structure which estimates phase shift common to.

(18) In the configuration of (3), the correlation calculating means includes the phase averaging means, and the arrangement information of the two-value signal of the first pilot signal and the complex vector signal output from the differential detection means. Calculating the correlation and supplying it to the broadband carrier frequency error calculating means, and estimating a phase variation common to all subcarriers from the phase angle of the vector obtained by the correlation operation and supplying the phase variation correction means to the phase variation correcting means. It is.

(19) In the configurations (1) to (18), the first pilot signal is configured to include a signal obtained by modulating a set of subcarriers arranged at the synchronization frequency of every symbol to the synchronization phase of every symbol. .

(20) In the configurations of (1), (3) to (13), and (18), m-phase PSK modulation is performed on a set of subcarriers in which the first pilot signal is arranged at a synchronization frequency of every symbol (m is a natural number). When a signal is included, the power supply means for multiplying the output of the differential detection means by the m power is supplied to the correlation calculating means.

(21) In the configuration of (2), (3), (14) to (18), m-phase PSK modulation is performed on a set of subcarriers in which the first pilot signal is arranged at a synchronization frequency of every symbol (m is a natural number). And a power supply means for multiplying the output of the differential detection means by the m power to be supplied to the phase averaging means, and a counting means for multiplying the output of the phase average means by 1 / m. have.

(22) In the structures (2), (3), (14) to (18), m-phase PSK modulation is performed on a set of subcarriers in which the first pilot signal is arranged at a synchronization frequency of every symbol (m is a natural number). ), It is determined in which region of the complex plane region the output of the differential detection means is divided into m by phases, and the output complex vector of the differential detection means is 2π according to the determination result. After rotating only the integer multiple of / m, the phase after rotation is always contained in the same area, and it is the structure further provided with the vector rotation means to supply to the said phase average means.

Hereinafter, as an OFDM transmission method according to the present invention, a 2k mode (1705 subcarriers used for transmission) of DVB-T (Digital Vldeo Broadcasting-Terrestrial) standard, which is a terrestrial digital television broadcasting system of Europe, is taken as an example. A description will be given with reference to FIGS. 4 to 22 according to an embodiment of the invention.

In the above standard, two pilot signals, referred to as distributed pilots (hereinafter referred to as SPs) and continuous pilots (hereinafter referred to as CPs), are transmitted using predetermined subcarriers.

4 is a schematic diagram showing a pilot signal arrangement of the DVB-T standard. In Fig. 4, k on the horizontal axis represents the index of the subcarrier, and n on the vertical axis represents the index of the symbol. In addition, the black annulus represents a subcarrier transmitting a pilot signal, while the white annulus represents a subcarrier transmitting another data.

The distributed pilot is transmitted by using a subcarrier whose index k = k _p satisfying the following expression (1). In Equation 1, mod denotes a redundant operation, and p is any non-integer.

The continuous pilot is k = {0, 48, 54, 87, 141, 156, 192, 201, 255, 279, 282, 333, 432, 450, 483, 525, 531, 618, 636, 714, 759. , 765, 780, 804, 873, 888, 918, 939, 942, 969, 984, 1050, 1101, 1107, 1110, 1137, 1140, 1146, 1206, 1269, 1323, 1377, 1491, 1683, 1704 It transmits using 45 satisfied subcarriers.

The pilot signal is modulated on the basis of the PN sequence w _k corresponding to the sub-carrier index, k is disposed, respectively, are multiplexed in synchronization with the synchronization amplitude and phase of each symbol, as shown in equation (2). In Equation 2, Re {c _k , n} represents the real part of the complex vector c _k , n corresponding to the subcarrier index k, the subcarrier of the symbol index n, and Im {c _k , n} is an imaginary number Represents wealth.

In the above standard, a transmission parameter signal (hereinafter referred to as TPS (Transmission Parameter Signal)) is transmitted using a predetermined subcarrier.

TPS uses 17 subcarriers that satisfy k = {34, 50, 209, 346, 413, 569, 595, 688, 790, 901, 1073, 1219, 1262, 1286, 1469, 1594, 1687}. The same information bits are transmitted as subcarriers in the same symbol.

At this time, if the information bit transmitted in the symbol at index n is set to Sn, the TPS is modulated with differential two-phase PSK (Phase Shift Keying) between symbols as shown in equation (3).

However, with respect to the head symbols of the frame (symbol index, n = 0) is to be an absolute phase modulation based on the PN sequence of the w _k as shown in equation (4).

(Example 1)

Fig. 5 is a block diagram showing the structure of an OFDM signal demodulation device according to the first embodiment of the present invention. In FIG. 5, the same code | symbol is attached | subjected and shown to the same part as FIG. In addition, also in the same figure, a thick solid line arrow shows a complex signal, and a thin solid line arrow shows a real signal. In addition, general control signals, such as a clock required for the operation of each component, are omitted so as not to be complicated.

In Fig. 5, the tuner 21 frequency-converts the OFDM signal input from the transmission path from the RF band to the IF band, and its output is supplied to the orthogonal demodulation circuit 22. The orthogonal demodulation circuit 22 demodulates the OFDM signal of the IF band into an OFDM signal of the baseband by using a fixed carrier generated therein, and the demodulation output of the carrier frequency (fc) correction circuit 23 It is supplied to the first input terminal.

The carrier frequency correction circuit 23 generates a correction carrier in accordance with a wideband carrier frequency error signal in units of subcarrier intervals supplied to the second input terminal and a narrowband carrier frequency error signal within subcarrier intervals supplied to the third input terminal. The carrier frequency error is corrected by multiplying the corrected carrier by the baseband OFDM signal supplied to the first input terminal, and its output is supplied to the narrowband carrier frequency error calculating circuit 24 and the FFT circuit 25.

The narrowband carrier frequency error calculating circuit 24 calculates the frequency error within the subcarrier interval by using the correlation between the guide period signal in the baseband OFDM signal and the rear of the effective symbol period signal, and the output is a carrier. It is supplied to the third input terminal of the frequency correction circuit 23. In addition, the FFT circuit 25 converts the effective symbol period signal in the baseband OFDM signal into a frequency domain by outputting the first symbol of the differential detection circuit 26 and the phase shift correction circuit 30. It is supplied to the input stage.

The differential detection circuit 26 calculates the phase variation between symbols by differentially detecting the symbols corresponding to the respective subcarriers output from the FFT circuit 25 from symbol to symbol, and the result of the calculation is correlated with the correlation calculation circuit 27. The phase averaging circuit 29 is supplied. The correlation calculation circuit 27 calculates a correlation value between the output of the differential detection circuit 26 and the arrangement information of the subcarrier transmitting the CP, and the correlation value is supplied to the broadband carrier frequency error calculation circuit 28. do. The wideband carrier frequency error calculating circuit 28 calculates a frequency error in units of subcarrier intervals from the peak position of the correlation value, and its output is supplied to the second input terminal of the carrier frequency correction circuit 23.

The phase averaging circuit 29 estimates CPE by averaging the output phase of the differential detection circuit 26 corresponding to CP in a symbol, and the output is supplied to the second input terminal of the phase shift correction circuit 30. . The phase shift correction circuit 30 multiplies the output of the FFT circuit 25 supplied to the first input terminal by the correction vector generated based on the output of the phase average circuit 29 supplied to the second input terminal. By correcting the CPE, its output is supplied to the detection circuit 31. The detection circuit 31 detects each subcarrier according to a modulation scheme, and then demaps the data signal.

Specifically, the differential detection circuit 26 is configured as shown in FIG. 6, so that the output of the FFT circuit 25 is supplied to the one-symbol period delay circuit 261 and the complex multiplier 263. The one symbol period delay circuit 261 delays the output of the FFT circuit 25 by one symbol period, and the delay output is supplied to the conjugate circuit 262. The conjugate circuit 262 calculates the complex conjugate by inverting the sign of the imaginary part of the output of the one-symbol period delay circuit 261, and the calculation result is supplied to the complex multiplier 263. The complex multiplier 263 multiplies the output of the FFT circuit 25 by the output of the conjugate circuit 262. The result of the calculation is the output of the differential detection circuit 26 and the correlation calculation circuit 27 and the phase average circuit. It is supplied to 29.

FIG. 7 shows a first configuration example of the correlation calculation circuit 27 in FIG. 5. In the correlation calculation circuit 27, the differential detection output from the differential detection circuit 26 is supplied to the shift register 2701. This shift register 2701 has a plurality of tap outputs corresponding to the arrangement of subcarriers for transmitting CP, and the tap outputs are supplied to an input terminal of the total circuit 2702. The total circuit 2702 calculates the sum of the tap outputs of the shift register 2701, and the result of the calculation is supplied to the power calculating circuit 2703. This power calculation circuit 2703 calculates the power of the output of the total circuit 2702, and the calculation result is supplied to the wideband carrier frequency error calculation circuit 28 as the output of the correlation calculation circuit 27.

According to the configuration shown in FIG. 7, when the subcarriers for transmitting the CP are output to all the tap outputs of the shift register 2701, the output of the correlation calculating circuit 27 shows a peak value. Therefore, in the wideband carrier frequency error calculating circuit 28, the carrier frequency error in subcarrier interval units can be estimated by detecting the peak value of the output of the correlation calculating circuit 27 and calculating the deviation from a predetermined timing. have.

FIG. 8 shows a second configuration example of the correlation calculation circuit 27 in FIG. 5. In FIG. 8, the same code | symbol is attached | subjected to the same part as FIG. 7, and a different part is demonstrated here.

In the correlation calculation circuit 27, the differential detection output from the differential detection circuit 26 is supplied to the intersymbol filter circuit 2704. The intersymbol filter circuit 2704 averages the output of the differential detection circuit 26 in the symbol direction, and the output is supplied to the shift register 2701. The configuration and operation after this shift register 2701 are the same as in the case of the first configuration example shown in FIG.

The intersymbol filter circuit 2704 in FIG. 8 is specifically configured as shown in FIG. 9, so that the output of the differential detection circuit 26 is supplied to the subtractor 27041. This subtractor 27041 subtracts the output of the one-symbol period delay circuit 27044 from the output of the differential detection circuit 26, and the output is supplied to the counter 27042. The counter 27042 multiplies the output of the subtractor 27041 by the coefficient α (0 ≦ α ≦ 1), and the calculation result is supplied to the adder 27043. The adder 27043 adds the output of the counter 27042 and the output of the one-symbol period delay circuit 27044, and the result of the calculation is supplied to the shift register 2701 as the output of the intersymbol filter circuit 2704. . The one symbol period delay circuit 27044 delays the output of the adder 27043 by one symbol period.

The intersymbol filter circuit 2704 configured as in FIG. 9 operates as a low pass filter of an infinite impulse response (hereinafter referred to as Infinite Impulse Response) type, corresponding to each subcarrier output from the differential detection circuit 26. A complex vector is averaged in the symbol direction. In the differential detection circuit 26, a signal obtained by differentially detecting a symbol between subcarriers carrying CP is considered to be a direct current signal having a synchronous amplitude and a synchronous phase of every symbol, ignoring the CPE component. Passes through the intersymbol filter circuit 2704. The signal obtained by differentially detecting the other subcarriers between symbols is blocked by the intersymbol filter circuit 2704 because the amplitude and phase of each symbol are random signals. In addition, since the noise component is also a random signal of every symbol, it is prevented by the inter-symbol filter circuit 2704.

Therefore, by adding the intersymbol filter circuit 2704 to the correlation calculation circuit 27 shown in FIG. 7, the output flow of the correlation calculation circuit 27 is suppressed, and the error in the wideband carrier frequency error calculation circuit 28 is suppressed. The estimation error of can be reduced.

FIG. 10 is a third configuration example of the correlation calculation circuit 27 in FIG. 5. In FIG. 10, the same code | symbol is attached | subjected to FIG. 7 and FIG. 8, and the other part is demonstrated here.

In this correlation calculation circuit 27, the output of the differential detection circuit 26 is averaged in the symbol direction by the intersymbol filter circuit 2704, and is then supplied directly to the power calculation circuit 2703. In other words, the power calculating circuit 2703 in this case calculates the output power of the intersymbol filter circuit 2704. The calculation result is supplied to the shift register 2705. The shift register 2705 has a plurality of tap outputs corresponding to the arrangement of the subcarriers for transmitting the CP, and the tap outputs are supplied to the input terminal of the total circuit 2706. The total circuit 2706 calculates the sum of the tap outputs of the shift register 2705, and the result of the calculation is supplied to the wideband carrier frequency error calculating circuit 28 as the output of the correlation calculating circuit 27.

In Fig. 10, the shift register 2705 holds a real signal, and since the total circuit 2706 also calculates the sum of the real signals, the shift register 2701 and the total circuit 2702 in Figs. The scale can be reduced compared to).

FIG. 11 shows a fourth configuration example of the correlation calculation circuit 27 in FIG. 5. In FIG. 11, the same code | symbol is attached | subjected to the same part as FIG. 10, and the description is abbreviate | omitted.

The comparison circuit 2707 in FIG. 11 extracts the subcarrier for transmitting the CP by comparing the output of the power calculation circuit 2703 with the threshold value set by the threshold setting circuit 2708, and thus the power calculation circuit 2703. Is outputted when the output side of the circuit) is large, and outputs "0" when the output side of the threshold value setting circuit 2708 is large. The output of this comparison circuit 2707 is supplied to the shift register 2709. The shift register 2709 has a plurality of tap outputs corresponding to the arrangement of the subcarriers for transmitting the CP, and the tap outputs are supplied to an input terminal of the total circuit 2710. The total circuit 2710 calculates the sum of the tap outputs of the shift register 2709, and the result of the calculation is supplied to the wideband carrier frequency error calculating circuit 28 as the output of the correlation calculating circuit 27.

In Fig. 11, since the shift register 2709 holds the two-value signal and the total circuit 2710 also calculates the sum of the two-value signals, the shift register 2701 and the total circuit in Figs. Compared to 2702, the scale can be greatly reduced. In addition, if the threshold value output from the threshold value setting circuit 2708 is controlled by the magnitude of the received signal, it is possible to prevent the misjudgment caused by the variation of the output level of the power calculation circuit 2703.

FIG. 12 shows a configuration example of the phase shift correction circuit 30 in FIG. 5. In this phase shift correction circuit 30, the output of the phase average circuit 29 is supplied to the adder 301. The adder 301 forms a cumulative adder together with a register 302 which holds a signal for one symbol period, and accumulates and adds the output of the phase average circuit 29 for each symbol to accumulate phase variations between symbols from the start of operation. The calculation result (output of the adder 301) is supplied to the correction vector calculation circuit (e ^-jφ ) 303. The correction vector calculation circuit 303 calculates a complex vector having an amplitude of 1 by using -1 times the output of the adder 301 as a phase angle, and the calculation result is supplied to the multiplier 304. do. This multiplier 304 multiplies the output of the correction vector calculation circuit 303 and the output of the FFT circuit 25. By this calculation, the CPE can be corrected.

With the above configuration, according to the configuration of the present embodiment, since the carrier frequency error in subcarrier interval units is calculated from the arrangement information of the subcarriers for transmitting the CPs included in every symbol, the frequency synchronization is introduced as compared with the conventional example. It can save time.

In addition, since the phase variation between symbols is calculated and corrected using the CP for each symbol, the influence of the CPE due to the phase noise of the tuner 21 can be eliminated.

(Example 2)

Fig. 13 is a block diagram showing the construction of an OFDM signal demodulation device according to a second embodiment of the present invention. 13, the same code | symbol is attached | subjected to the same part as FIG. In addition, also in the same figure, a thick solid line arrow represents a complex signal, a thin solid line arrow represents a real signal, and general control signals, such as a clock required for operation | movement of each component, are abbreviate | omitted so that description may become complicated.

The OFDM signal demodulation device shown in FIG. 13 corrects the carrier frequency error in the tuner 32 instead of the carrier frequency correction circuit 23 in FIG. The tuner 32 controls the local oscillation frequency in accordance with a wideband carrier frequency error signal in units of subcarrier intervals supplied to the second input terminal and a narrowband carrier frequency error signal in subcarrier intervals supplied to the third input terminal. Frequency conversion of the OFDM signal supplied to one input terminal from the RF band to the IF band, the output is supplied to the orthogonal demodulation circuit 22. Other configurations and operations are omitted because they are the same as in FIG.

(Example 3)

Fig. 14 is a block diagram showing the construction of an OFDM signal demodulation device according to the third embodiment of the present invention. 14, the same code | symbol is attached | subjected and shown to the same part as FIG. In addition, also in the same figure, a thick solid line arrow represents a complex signal, a thin solid line arrow represents a real signal, and general control signals, such as a clock required for operation | movement of each component, are abbreviate | omitted so that description may become complicated.

The OFDM signal demodulation device shown in FIG. 14 corrects the carrier frequency error in the orthogonal demodulation circuit 33 instead of the carrier frequency correction circuit 23 in FIG. The orthogonal demodulation circuit 33 controls the local oscillation frequency according to the wideband carrier frequency error signal in units of subcarrier intervals supplied to the second input terminal and the narrowband carrier frequency error signal within subcarrier intervals supplied to the third input terminal. The demodulation output is supplied to the narrowband carrier frequency error calculating circuit 24 and the FFT circuit 25 by demodulating the OFDM signal of the IF band supplied to the first input terminal into the OFDM signal of the baseband. Other configurations and operations are omitted because they are the same as in FIG.

(Example 4)

Fig. 15 is a block diagram showing the construction of an OFDM signal demodulation device according to a fourth embodiment of the present invention. 15, the same code | symbol is attached | subjected to the same part as FIG. In addition, also in the same figure, a thick solid line arrow represents a complex signal, a thin solid line arrow represents a real signal, and general control signals, such as a clock required for operation | movement of each component, are abbreviate | omitted so that description may become complicated.

The OFDM signal demodulation device shown in FIG. 15 corrects the narrowband carrier frequency error within the subcarrier interval in the carrier frequency (fc) correction circuit 34, and the wideband carrier in the unit of subcarrier interval in the shift circuit 35. This is to correct the frequency error. The carrier frequency correction circuit 34 generates a correction carrier based on the narrowband carrier frequency error signal within the subcarrier interval supplied to the second input terminal, and multiplies the correction carrier by the baseband OFDM signal supplied to the first input terminal. By correcting the carrier frequency error, the output is supplied to the narrow band carrier frequency error calculating circuit 24 and the FFT circuit 25. The shift circuit 35 shifts the output of the FFT circuit 25 in the frequency direction based on the wideband carrier frequency error signal in units of subcarrier spacing supplied to the second input terminal, the output of which is a differential detection circuit 26. And a first input terminal of the phase shift correction circuit. Other configurations and operations are omitted because they are the same as in FIG.

Here, the carrier frequency error in the subcarrier interval unit is a frequency error that is an integer period in the effective symbol period length. However, since the guide period exists in the OFDM signal, the carrier period error in the frequency domain is changed to the guide period length. Generate a phase rotation for every symbol that depends. Therefore, as in the configuration of Fig. 15, in the case of correcting the wideband carrier frequency error by shift in the frequency domain, a means for correcting this phase rotation is required. However, since this phase rotation is common to all subcarriers, when the circuit for CPE removal is provided as shown in FIG. 15, the phase shift correction circuit 30 automatically corrects it.

(Example 5)

Fig. 16 is a block diagram showing the construction of an OFDM signal demodulation device according to a fifth embodiment of the present invention. 16, the same code | symbol is attached | subjected to the same part as FIG. In addition, also in the same figure, a thick solid line arrow represents a complex signal, a thin solid line arrow represents a real signal, and general control signals, such as a clock required for operation | movement of each component, are abbreviate | omitted so that description may become complicated.

The OFDM signal demodulation device shown in FIG. 16 corrects the CPE in the detection circuit 36 instead of the phase shift correction circuit 30 in FIG. 5. The detection circuit 36 generates a correction vector based on the output of the phase average circuit 29 supplied to the second input terminal, and multiplies the correction vector by a detection vector according to the modulation scheme of each subcarrier. Using the detection vector, the output of the FFT circuit 25 is detected and the CPE is corrected, and then demapping to restore the data signal. Other configurations and operations are omitted because they are the same as in FIG.

FIG. 17 shows a configuration example corresponding to a modulation method on the premise of synchronous detection using the SP signal of the detection circuit 36 in FIG. 16. In the detection circuit 36, the output of the FFT circuit 25 is supplied to the first input terminal of the complex divider 3604 and the first input terminal of the complex divider 3608. The pilot generation circuit 3603 generates an SP in synchronization with the output of the FFT circuit 25, and the output is supplied to the second input terminal of the complex divider 3604. The complex divider 3604 divides the SP included in the output of the FFT circuit 25 supplied to the first input terminal from the regular SP output by the pilot generation circuit 3603 supplied to the second input terminal. It is to calculate the characteristics of the transmission path acting on. The output is supplied by a switch (SW) 3605 to an output of the memory 3606 and optionally to a first input of a complex multiplier 3602.

On the other hand, the output of the phase averaging circuit 29 is supplied to the correction vector calculating circuit e ^-jφ 3601. The correction vector calculation circuit 3601 calculates a complex vector having an amplitude of 1 by using the output of the phase average circuit 29 as a phase angle, and the calculation result is the second value of the complex multiplier 3602. It is supplied to the input stage. The switch 3605 selects the output of the complex divider 3604 when the output of the complex divider 3604 corresponds to the SP (one symbol of four symbols when focusing on one subcarrier), and otherwise. (Similarly, three of the four symbols) select and output the output of the memory 3606.

The complex multiplier 3602 is an output of the complex divider 3604 selectively supplied by the switch 3605 to the first input stage or an output of the memory 3606 and a correction vector calculation circuit 3601 supplied to the second input stage. By multiplying the output of the result of the calculation, the operation result is supplied to the filter circuit 3608 and also to the memory 3606. This memory 3606 maintains the output of the complex multiplier 3602 for four symbol periods (until the next SP is transmitted in the subcarrier of interest). By these operations, the CPE can be corrected for the transmission path characteristic acting on the subcarrier (one subcarrier of the three subcarriers) transmitting the SP.

The filter circuit 3608 interpolates the output of the complex multiplier 3602 in the frequency (subcarrier) direction to obtain transmission path characteristics (compensated for CPE) acting on all subcarriers, and the output is a complex divider. Supplied to a second input terminal of 3608. The complex divider 3608 synchronously detects the output of the FFT circuit 25 by dividing the output of the FFT circuit 25 supplied to the first input terminal by the output of the filter circuit 3607 supplied to the second input terminal. It is. The output is supplied to demapping circuit 3609. This demapping circuit 3609 recovers the data signal by demapping the output of the complex divider 3608 according to a modulation scheme.

FIG. 18 shows an example of the configuration of the detection circuit 36 in FIG. 16 that corresponds to the modulation method on the premise of differential detection. In this detection circuit 36, the output of the FFT circuit 25 is supplied to the first input terminal of the one-symbol period delay circuit 3610 and the complex divider 3611. The one symbol period delay circuit 3610 delays the output of the FFT circuit 25 by only one symbol period, and the output is supplied to the first input terminal of the complex multiplier 3602. On the other hand, the output of the phase averaging circuit 29 is supplied to the correction vector calculating circuit e ^-jφ 3601.

The correction vector calculation circuit 3601 calculates a complex vector having an amplitude of 1 by using the output of the phase average circuit 29 as a phase angle, and the calculation result is the second value of the complex multiplier 3602. It is supplied to the input stage. The complex multiplier 3602 multiplies the output of the one-symbol period delay circuit 3610 supplied to the first input terminal with the output of the correction vector calculating circuit 3601 supplied to the second input terminal, thereby providing a signal preceding the one-symbol period. By correcting the CPE, the result of the calculation is supplied to the second input terminal of the complex divider 3611.

The complex divider 3611 differentially detects the output of the FFT circuit 25 by dividing the output of the FFT circuit 25 supplied to the first input terminal by the output of the complex multiplier 3602 supplied to the second input terminal. The output is supplied to the demapping circuit 3612. The demapping circuit 3612 recovers the data signal by demapping the output of the complex divider 3611 according to a modulation scheme.

According to the present embodiment having the above configuration, since part of the processing of the phase shift correction circuit 30 and the detection circuit 31 in the first embodiment can be shared, the circuit scale can be reduced.

(Example 6)

Fig. 19 is a block diagram showing the construction of an OFDM signal demodulation device according to a sixth embodiment of the present invention. 19, the same code | symbol is attached | subjected to the same part as FIG. In addition, also in the same figure, a thick solid line arrow represents a complex signal, a thin solid line arrow represents a real signal, and general control signals, such as a clock required for operation | movement of each component, are abbreviate | omitted so that description may become complicated.

The OFDM signal demodulation apparatus shown in FIG. 19 is configured to execute the processing of the correlation calculating circuit 27 and the phase average circuit 29 in FIG. 5 together with the correlation circuit 37.

FIG. 20 is a configuration example of the correlation circuit 37 in FIG. 19, and the output of the differential detection circuit 26 is supplied to the shift register 371. The shift register 371 has a plurality of tap outputs corresponding to the arrangement of the subcarriers for transmitting the CP, and the tap outputs are supplied to the input terminal of the total circuit 372. The total circuit 372 calculates the sum of the tap outputs of the shift register 371, and the result of the calculation is supplied to the power calculating circuit 373 and the phase calculating circuit tan ^-1 374.

The power calculation circuit 373 calculates the output power of the total circuit 372, and the calculation result is supplied to the broadband carrier frequency error calculation circuit 28 as the first output of the correlation calculation circuit 37. On the other hand, the phase calculation circuit 374 calculates the phase of the output of the total circuit 372, and the result of the calculation is the second output of the correlation calculation circuit 37 to the second input terminal of the phase shift correction circuit 30. Supplied.

Here, when the carrier frequency is synchronized, the subcarrier for transmitting the CP is output to the tap output of the shift register 371, so that the output of the total circuit 372 averages the variation between symbols in the subcarrier for transmitting the CP within the symbol. It becomes one.

According to the present embodiment having the above configuration, since a part of the processing of the correlation calculating circuit 27 and the phase average circuit 29 in the first embodiment can be shared, the circuit scale can be reduced.

(Example 7)

Fig. 21 is a block diagram showing the construction of an OFDM signal demodulation device according to Embodiment 7 of the present invention. 21, the same code | symbol is attached | subjected to the same part as FIG. In addition, also in the same figure, a thick solid line arrow represents a complex signal, a thin solid line arrow represents a real signal, and general control signals, such as a clock required for operation | movement of each component, are abbreviate | omitted so that description may become complicated.

The OFDM signal demodulation device shown in FIG. 21 performs carrier frequency synchronization and CPE removal using TPS, and the power circuit 38 and the counter 39 are added to the first embodiment.

Here, the power multiplier circuit 38 calculates the square power of the complex vector corresponding to each subcarrier output from the differential detection circuit 26, and the result of the calculation is the correlation calculation circuit 27 and the phase average circuit 29. Is supplied. This squared arithmetic eliminates 180 degrees of uncertainty in phase variation due to TPS being differential two-phase PSK modulation between symbols.

The correlation calculating circuit 27 calculates a correlation value between the output of the power circuit 38 and at least one of the placement information in the subcarrier transmitting the CP and the subcarrier transmitting the TPS. The wideband carrier frequency error calculating circuit 28 is supplied. The phase averaging circuit 29 estimates the CPE by averaging the phase of the output of the power circuit 38 corresponding to at least one of CP and TPS in symbols, and supplies the output to the counter 39. do. The counter 39 corrects by halving the phase shift between symbols doubled by the power circuit 38, and its output is supplied to the second input terminal of the phase shift correction circuit 30.

In general, when the TPS is m-phase PSK modulated (m is a natural number), the power circuit 38 calculates the m-th power of the complex vector corresponding to each subcarrier output from the differential detection circuit 26, The counter 39 multiplies the output of the phase average circuit 29 by 1 / m.

In the present embodiment having the above-described configuration, since the carrier frequency error and the phase variation between symbols in subcarrier interval units are calculated and corrected by using TPS in addition to CP, the error caused by the influence of noise is compared with the first embodiment. Can be reduced.

(Example 8)

Fig. 22 is a block diagram showing the construction of an OFDM signal demodulation device according to Embodiment 8 of the present invention. In addition, in FIG. 22, the same code | symbol is attached | subjected and shown to the same part as FIG. In addition, in the same figure, the solid arrow indicates a complex signal, the thin solid arrow indicates a real signal, and the general control signals such as a clock required for the operation of each component are omitted so as not to be complicated.

The OFDM signal demodulation device shown in Fig. 22 performs carrier frequency synchronization and CPE removal using TPS, and the power circuit 38 and the vector rotation circuit 40 are added to the first embodiment.

Here, the power multiplication circuit 38 calculates the square power of the complex vector corresponding to each subcarrier output from the differential detection circuit 26, and the result of the calculation is supplied to the correlation calculation circuit 27. This squared arithmetic eliminates 180 degrees of uncertainty in phase variation due to TPS being differential two-phase PSK modulation between symbols. The correlation calculating circuit 27 calculates a correlation value between the output of the power circuit 38 and the subcarrier transmitting the CP and the subcarrier transmitting the TPS and at least one arrangement information. The wideband carrier frequency error calculating circuit 28 is supplied.

On the other hand, the vector rotation circuit 40 determines in which region of the complex planar region the output of the differential detection circuit 26 is divided by the virtual axis, and according to the determination result of the differential detection circuit 26 By rotating the output complex vector by π and ensuring that the phase after rotation is always in the same region, 180 degrees of uncertainty in phase variation due to differential two-phase PSK modulation between symbols is eliminated, and the output is phase The average circuit 29 is supplied. The phase averaging circuit 29 estimates the CPE by averaging the phase of the output of the vector rotation circuit 40 corresponding to at least one of the CP and the TPS in a symbol, and outputs the phase shift correction circuit 30. Is supplied to the second input terminal.

In general, when the TPS is m-phase PSK modulated (m is a natural number), the vector rotation circuit 40 includes the output of the differential detection circuit 26 in any region of the complex planar region divided into m by phases. In accordance with the result of the determination, the output complex vector of the differential detection circuit 26 is rotated by an integer multiple of 2π / m so that the phase after rotation is always included in the same region.

In the present embodiment having the above-described configuration, similarly to the seventh embodiment, since the carrier frequency error and the phase variation between symbols in the subcarrier interval unit are calculated and corrected using the TPS in addition to the CP, compared with the first embodiment, The error due to the influence of noise can be reduced.

In the embodiment of the present invention, the power calculation inside the correlation calculation circuits 27 and 37 may be used to calculate the magnitude of the signal such as the amplitude, the sum of the amplitudes of the real part and the imaginary part.

Further, in the embodiment of the present invention, the phase averaging circuit 29 averages the output complex vector of the differential detection circuit 26 or the vector rotation circuit 40 corresponding to at least one of the CP and the TPS within the symbol. The CPE may be approximated by calculating the phase.

Further, in the embodiment of the present invention, the wideband carrier frequency error calculating circuit 28 determines the synchronous state of the carrier frequency on the basis of the output of the correlation calculating circuit 27, and when in the synchronous state, the unit of subcarrier spacing By stopping the output of the carrier frequency error signal, and having the protection function for the front and the rear in the synchronization determination, it is possible to prevent the malfunction due to the influence of noise or fading.

In the above description, the 2B mode of the DVB-T standard has been described as an example, but in Embodiments 1 to 8, a set of subcarriers arranged at the same frequency for each symbol is modulated with the same phase for each symbol. A transmission scheme such as transmitting a signal may be used. In Examples 7 to 8, transmission such as transmitting a signal obtained by m-phase PSK modulation (m is a natural number) of a set of subcarriers arranged at the same frequency for each symbol. Needless to say, that's the way.

As described above, the OFDM signal demodulation apparatus according to the present invention calculates the frequency error in the unit of subcarrier interval by using the pilot signal arranged at the same frequency for each symbol, thereby reducing the frequency synchronization lead-in time. .

In addition, it is possible to eliminate the influence of the CPE due to the phase noise of the tuner by calculating and correcting the phase variation between symbols using pilot signals arranged at the same frequency for each symbol.

As described above, according to the present invention, an OFDM signal demodulation device capable of shortening the frequency synchronization lead-in time more and eliminating the influence of the CPE due to the phase noise of the tuner can be provided.

Claims (22)

An apparatus for demodulating an orthogonal frequency division multiplex signal comprising a first pilot signal arranged at the same frequency every symbol, Fourier transform means for converting the orthogonal frequency division multiplexed signal into a frequency axis signal by Fourier transforming; Differential detection means for calculating the variation between symbols by differentially detecting the output of said Fourier transform means; Correlation calculation means for calculating correlation between the arrangement information of the first pilot signal and the output of the differential detection means; Wide-band carrier frequency error calculating means for estimating a carrier frequency error in units of subcarrier intervals by detecting the peak position of the output of the correlation calculating means; And a wideband carrier frequency correction means for correcting a carrier frequency on the basis of the output of the wideband carrier frequency error calculating means. An apparatus for demodulating an orthogonal frequency division multiplex signal comprising a first pilot signal arranged at the same frequency every symbol, Fourier transform means for converting the orthogonal frequency division multiplexed signal into a frequency axis signal by Fourier transforming; Differential detection means for calculating the variation between symbols by differentially detecting the output of said Fourier transform means; Phase averaging means for estimating a phase variation common to all subcarriers by averaging the phase of the output of the differential detection means corresponding to the first pilot signal in a symbol; Orthogonal frequency division multiplexing signals comprising a phase shift correction means for calculating a correction vector for every symbol from the output of the phase averaging means, and correcting a phase shift common to all subcarriers based on the correction vector. Demodulation device. An apparatus for demodulating an orthogonal frequency division multiplex signal comprising a first pilot signal arranged at the same frequency every symbol, Fourier transform means for converting the orthogonal frequency division multiplexed signal into a frequency axis signal by Fourier transforming; Differential detection means for calculating the variation between symbols by differentially detecting the output of said Fourier transform means; Correlation calculation means for calculating correlation between the arrangement information of the first pilot signal and the output of the differential detection means; Wide-band carrier frequency error calculating means for estimating a carrier frequency error in units of subcarrier intervals by detecting the peak position of the output of the correlation calculating means; Broadband carrier frequency correction means for correcting a carrier frequency based on an output of said broadband carrier frequency error calculating means; Phase averaging means for estimating a phase variation common to all subcarriers by averaging the phase of the output of the differential detection means corresponding to the first pilot signal in a symbol; Orthogonal frequency division multiplexing signals comprising a phase shift correction means for calculating a correction vector for every symbol from the output of the phase averaging means, and correcting a phase shift common to all subcarriers based on the correction vector. Demodulation device. The method according to any one of claims 1 to 3, And said correlation calculating means calculates the magnitude of the correlation between the arrangement information by the binary signal of said first pilot signal and the complex vector signal output from said differential detection means. The method according to any one of claims 1 to 3, The correlation calculating means calculates the magnitude of the correlation between the arrangement information by the binary signal of the first pilot signal and the complex signal obtained by averaging the output of the differential detection means in the symbol direction. Signal demodulation device. The method according to any one of claims 1 to 3, The correlation calculating means calculates a correlation between the arrangement information by the binary signal of the first pilot signal and the magnitude of the real signal obtained by averaging the output of the differential detection means in the symbol direction. Signal demodulation device. The method according to claim 1 or 3, The correlation calculating means compares the arrangement information by the two-value signal of the first pilot signal and the magnitude of the signal obtained by averaging the output of the differential detection means in the symbol direction with a predetermined threshold value and the two-value conversion signal. An orthogonal frequency division multiple signal demodulation device for calculating the correlation. The method of claim 7, wherein And said correlation calculating means controls said threshold value by the magnitude of a received signal. The method according to any one of claims 1 to 3, The broadband carrier frequency correction means controls the local oscillation frequency of the tuner which performs band conversion of the orthogonal frequency division multiplex signal input from the transmission path based on the output of the broadband carrier frequency error calculating means. Multiple signal demodulation device. The method according to claim 1 or 3, The broadband carrier frequency correcting means controls the local oscillation frequency of the orthogonal demodulation means for orthogonal demodulating the baseband orthogonal frequency division multiplexing signal based on the output of the broadband carrier frequency error calculating means. Demodulation device. The method according to any one of claims 1 to 3, The broadband carrier frequency correction means generates a correction carrier based on the output of the broadband carrier frequency error calculating means, and multiplies the correction carrier by an input signal of the Fourier transform means. Device. The method of claim 1, The broadband carrier frequency correcting means shifts the output signal of the Fourier transforming means in the frequency direction based on the output of the broadband carrier frequency error calculating means, and corrects phase rotation between symbols generated depending on the guide period length. Orthogonal frequency division multiple signal demodulation device, characterized in that. The method of claim 3, wherein And said wideband carrier frequency correcting means shifts the output signal of said Fourier transforming means in the frequency direction based on the output of said wideband carrier frequency error calculating means. The method according to claim 2 or 3, The phase shift correction means is built in detection means for detecting the output of the Fourier transform means according to the modulation method of each subcarrier, and is common to all subcarriers based on the output of the correction vector calculating means at the same time as the detection. An orthogonal frequency division multiple signal demodulation device for correcting in-phase variation. The method of claim 14, In an apparatus for demodulating an orthogonal frequency division multiplex signal which transmits, in addition to the first pilot signal, a second pilot signal distributed in a subcarrier-symbol region and distributed periodically, the detecting means is configured to perform the correction vector calculation means. An orthogonal frequency division multiple signal demodulation device, characterized in that synchronous detection of each subcarrier is performed using the second pilot signal while correcting a phase shift common to all subcarriers based on the output. The method of claim 14, In an apparatus for demodulating an orthogonal frequency division multiplex signal for differentially modulating a data signal between symbols and transmitting the signal, the detection means corrects phase variations common to all subcarriers based on the output of the correction vector calculating means, respectively. Orthogonal frequency division multiple signal demodulation device, characterized in that differential detection between the subcarrier of the symbol. The method according to claim 2 or 3, The phase averaging means estimates a phase variation common to all subcarriers by averaging the output complex vector of the differential detection means corresponding to the first pilot signal in a symbol and calculating the phase. Frequency Division Multiple Signal Demodulation Device. The method of claim 3, wherein The correlation calculating means includes the phase averaging means and calculates a correlation between the arrangement information by the two-value signal of the first pilot signal and the complex vector signal output from the differential detection means to calculate the broadband carrier frequency error means. And a phase shift common to all subcarriers from a phase angle of the vector obtained by the correlation operation, and supplied to the phase shift correcting means. The method according to any one of claims 1 to 18, And the first pilot signal includes a signal obtained by modulating a set of subcarriers arranged at the same frequency for each symbol in the same phase for each symbol. The method according to any one of claims 1, 3 to 13, 18, When the first pilot signal includes a signal obtained by m-phase PSK modulation (m is a natural number) of a set of subcarriers arranged at the same frequency every symbol, the output of the differential detection means is multiplied by the correlation calculation means. An orthogonal frequency division multiple signal demodulation device, characterized in that it further comprises a power supply means for supplying to the power supply. The method according to any one of claims 2, 3 and 14 to 18, When the first pilot signal includes a signal obtained by m-phase PSK modulation (m is a natural number) of a set of subcarriers arranged at the same frequency for each symbol, the output of the differential detection means is multiplied by m, and the phase average is obtained. Power means for supplying the means; An orthogonal frequency division multiple signal demodulation device further comprising counting means for multiplying the output of said phase averaging means by 1 / m. The method according to any one of claims 2, 3 and 14 to 18, When the first pilot signal includes a signal obtained by m-phase PSK modulation (m is a natural number) of a set of subcarriers arranged at the same frequency every symbol, By determining in which region of the complex planar region the output of the differential detection means is divided into m by phases, and rotating the output complex vector of the differential detection means by an integer multiple of 2π / m according to the determination result, An orthogonal frequency division multiple signal demodulation device comprising a vector rotation means for supplying the phase averaging means after the phase after rotation is always included in the same region.