KR101364226B1 - Motor drive control apparatus and air-conditioning equipment - Google Patents

Abstract

An object of the present invention is to perform position sensorless control capable of detecting the position of a rotor from a low speed region including a stop state to a high speed region and having a small current distortion and not requiring detection of a neutral point potential.
When the 180-degree conduction means (control means) 6 supplies the pulse signal PWM-controlled by the 180-degree conduction to the inverter 3 as means for solving such a problem, each switch element (Sup) of the inverter 3 ~Swn) are PWM controlled and switched. As a result, the three-phase alternating current controlled by PWM is supplied from the inverter 3 to the alternating-current motor 1. At this time, the adjusting means 8 stops the pulse signal of 180-degree energization output from the stopping means 7 in accordance with the operating state (rotational speed, torque, motor current, motor applied voltage, Change the period. Thus, the stopping means 7 stops the pulse signal supplied from the 180-degree energizing means 6 to the inverter 3 for a period corresponding to the stop period set by the adjusting means 8, And stops transmission.

Description

TECHNICAL FIELD [0001] The present invention relates to a motor drive control device and an air-

TECHNICAL FIELD The present invention relates to a technique for driving and controlling an AC motor by a position sensorless inverter of a 180-degree conduction type by PWM (Pulse Width Modulation) control.

As a technology of a motor drive control device that drives an AC motor inexpensively and robustly, there is a position sensorless control by an inverter that performs PWM control. In this position sensorless control, since the rotor phase is estimated on the basis of the current flowing in the AC motor and the applied voltage, it is not necessary to attach the position sensor. It is desired that such a position sensorless control can realize a wide driving range and a high operation efficiency equal to or more than that in the case of control with a position sensor.

As the position sensorless control, there is a method of estimating the rotor phase based on the induced (induced) voltage of the alternating-current motor. As a method of estimating the rotor phase based on the induced voltage, several control methods have been proposed in accordance with an energizing method (for example, a 120-degree energizing method or a 180-degree energizing method) of an inverter for performing PWM control. Here, it is assumed that the position sensorless 120 degree electric current method using the organic voltage is called the 120 degree electric current method using the organic voltage, and the 180 degree electric current method using the position sensorless using the organic voltage is called the 180 degree electric current method using the organic voltage. The problem with the position sensorless control method using these induced voltages is that it is known that the estimation error of the rotor phase tends to occur because the induced voltage becomes small in the low speed region of the alternating current motor. Particularly, at the time of stopping the AC motor, the induced voltage can not be estimated because the induced voltage is not generated.

Therefore, in order to solve such a problem, a position sensorless control method using magnetic saturation has been proposed. This technique uses magnetic saturation to estimate the rotor phase even at low speeds including stopping. For example, when a 120-degree conduction method is used, a method of detecting a voltage due to magnetic saturation occurring in an open phase (a 120-degree conduction method using a magnetic saturation type) has been proposed (See Patent Document 1). Here, the open phase is a phase in which both the upper and lower arm elements of the inverter circuit are stopped. In the technique of Patent Document 1, since the base voltage generated in the open phase by magnetic saturation changes depending on the rotor phase, the rotor phase is estimated by detecting this base voltage.

However, since the current waveform is distorted in the 120-degree conduction method, iron loss is increased and the efficiency of the AC motor is lowered. Thus, in the technique of Patent Document 1, a method of switching the 120-degree energization type using magnetic saturation in the low speed range and the 180-degree energization type using the induced voltage in the middle / high speed range is disclosed. Since the latter (180-degree energization method using an induced voltage) is sinusoidal wave driving, the current distortion is smaller than that of the former (120-degree energization type using magnetic saturation) and the efficiency deterioration of the AC motor can be suppressed. On the other hand, since electrons (120-degree conduction method using a magnetic saturation type) must be used in a low speed range including stopping, there is a problem that a current distortion becomes large in a low speed range.

That is, although the 180-degree current-carrying type using the induced voltage can reduce the current distortion, the estimation error of the rotor phase becomes large because the induced voltage becomes small at the low speed region of the alternating-current motor. Further, in the magnetic saturation-use type 120-degree energization method, the rotor phase can be accurately estimated even at a low speed region of the alternating-current motor by the electromotive voltage generated by the magnetic saturation generated in the open circuit. However, .

Therefore, in order to estimate the rotor phase at a low speed range including stopping when the 180-degree energization type using the induced voltage with a small current distortion is used, a method of detecting the neutral point potential (Refer to Patent Document 2). This method estimates the rotor phase by detecting the neutral point potential by using the fact that the detected neutral point potential depends on the rotor phase as well as the open-circuit voltage.

Japanese Patent Application Laid-Open No. 2009-189176 Japanese Patent Application Laid-Open No. 2010-74898

However, in the case of the 180-degree conduction method using the neutral point described in Patent Document 2, the phase of the rotor can be estimated from a low speed range to a high speed range with small current distortion. However, since the neutral point potential is detected, do. That is, in order to detect the neutral point potential using the neutral point use type 180-degree energization method, it is necessary to provide the detection wiring in the AC motor, so that the detection system of the AC motor becomes complicated and the structure of the AC motor becomes complicated have. When the three-phase alternating-current motor is a compressor driving motor of the air conditioner, it is necessary to draw four lines from a compressor driving motor provided inside the compressor. As a result, there is a fear that the air conditioning equipment is cost-up or the reliability of the air conditioning equipment is lowered. In addition, since the wiring of the detection system must be changed inside the air conditioner, existing air conditioners can not be used as they are, which is a problem that the versatility is not excellent.

SUMMARY OF THE INVENTION It is therefore an object of the present invention to provide a motor control device capable of detecting the position of a rotor from a low speed region to a high speed region including a stopping state and having a small current distortion, It is an object of the present invention to provide a motor drive control device capable of performing position sensorless control.

In order to solve the above problems, an air conditioning apparatus using the motor drive control apparatus and the motor drive control apparatus of the present invention converts DC power supplied from a DC power supply into desired AC power by PWM control, Voltage detecting means for detecting a motor-applied voltage applied to a terminal of the alternating-current motor, and a pulse signal of 180-degree energization to output a pulse signal A control means for PWM-controlling the inverter; a stop means for stopping a pulse signal of a predetermined phase among the pulse signals by a predetermined stop period; and a control means for stopping the operation state when the AC motor is at a predetermined rotation speed or lower And an adjusting means for variably adjusting the stopping period set in the stopping means in accordance with the control signal.

According to the present invention, it is possible to perform the position sensorless control in which the position of the rotor can be detected from the low speed region including the stop state to the high speed region, the current distortion is small, and the detection of the neutral point potential is unnecessary.

1 is a block diagram showing a circuit configuration of a motor drive control device according to a first embodiment;
2 is a circuit diagram showing a detailed circuit configuration of the voltage detecting means shown in Fig.
Fig. 3 is a waveform diagram of a U-phase voltage, a U-phase current and a pulse signal in the inverter shown in Fig. 1. Fig. 3 (a) Enlarged view.
Fig. 4 is a diagram showing the U-phase terminal voltage Vua in the energizing period Ton and the stop period Toff in the waveform diagram shown in Fig. 3; Fig.
5 is a waveform diagram of a U-phase voltage, a U-phase current and a pulse signal when a real machine is driven by the motor drive control device of the first embodiment, wherein (a) shows the waveform of the U- (B) shows the waveform of the U-phase current, and (c) shows the waveform of the pulse signal.
6 is an image view at the time of detecting each phase terminal voltage of the alternating-current motor shown in Fig. 1; Fig.
Fig. 7 is a characteristic diagram showing the relationship between the rotor phase and the electromotive voltage of the alternating-current motor shown in Fig. 1; Fig.
Fig. 8 is a characteristic diagram showing the relationship between the stop period (Toff) of the pulse signal of the inverter shown in Fig. 1 and the current distortion?; Fig.
Fig. 9 is a switching waveform diagram of a 120-degree energization type inverter using magnetic saturation. Fig.
Fig. 10 is a characteristic diagram showing the relationship between the rotational speed [omega], the current distortion [epsilon], and the stop period (Toff) of the pulse signal of the alternating-current motor shown in Fig.
Fig. 11 is a waveform diagram of an ideal U phase current in the inverter shown in Fig. 1. Fig.
12 is a conceptual diagram of an alternating-current motor in the case where a winding variable means and a magnetic flux amount variable rotor are provided.
13 is a configuration diagram of an air conditioner using the motor drive control device according to the second embodiment.
14 is a waveform diagram of torque ripple (pulse) and motor current rms value of an AC motor for driving a compressor.
Fig. 15 is a vector diagram of the three-phase alternating-current voltage of the alternating-current motor driven by the motor drive control device shown in Fig.
Fig. 16 is a configuration diagram of phase estimation at the time of free-run restart in the 180-degree energizing means shown in Fig. 1. Fig.

Next, a mode for carrying out the invention (hereinafter referred to as " embodiment ") will be described in detail with reference to the drawings as appropriate.

"summary"

The motor drive control device according to the present embodiment is an inverter that performs PWM control by a 180-degree energizing method for supplying power to an AC motor under position sensorless control. The inverter controls the switch elements of the upper and lower arms of the inverter in a predetermined stop period (Open phase) is set at a predetermined value. And the magnet position (rotor phase) is detected using the measured electromotive voltage to perform the position sensorless control from the low speed range including the stopping time to the high speed range Lt; / RTI >

In the 180-degree energization method, the rotor phase is generally measured based on the motor current. However, when the rotation speed of the motor is less than the predetermined value, the rotor phase can not be accurately measured based on the motor current. However, if the switch element is stopped, the base voltage can be accurately measured. At this time, the measurable voltage is the sum of the electromotive voltage and the induced voltage. The electromotive voltage depends on the rotor phase. Further, the induced voltage differs depending on the rotational speed of the motor. Therefore, if the rotational speed of the motor is known, the induced voltage can be obtained, and the induced voltage can be obtained by subtracting the induced voltage from the measured induced voltage. In this manner, the position sensorless control is performed at a low speed range including stopping (the rotational speed of the motor is equal to or less than a predetermined value) by using the stop period of the open phase in the 180 degree energization method. Further, by adjusting the stop period of the open phase according to the rotation speed of the motor, etc., current distortion is reduced.

&Quot; First embodiment "

<Overall Configuration of Motor Drive Control Device>

Fig. 1 shows a circuit configuration of the motor drive control device 100 according to the first embodiment. The motor drive control apparatus 100 includes an inverter 3 for converting the DC power supplied from the DC power supply 2 into a desired AC power by PWM control and supplying the AC power to the AC motor 1, Current detecting means 4 for detecting three-phase alternating currents Iu, Iv, Iw (motor current) flowing through the alternating-current motor 1 driven by the alternating-current motor 3, A voltage detecting means 5 for detecting AC voltages Vu, Vv and Vw (motor applied voltage) and 180-degree energizing means (control means) 6 for PWM-controlling the inverter 3 with a 180- A stop means 7 for stopping a pulse signal of a predetermined phase from a pulse signal of 180 degree energization for a predetermined stop period and a control means for stopping the stop means 7 in accordance with the operation state when the AC motor 1 is at a predetermined speed or lower, (8) for adjusting the stopping period of the 180-degree energized pulse signal output from the control unit (7).

<Outline of Operation of Motor Drive Control Apparatus>

In the motor drive control device 100 shown in Fig. 1, the DC power supply 2 is connected between the positive terminal side terminal 3a of the inverter 3 and the negative terminal side 3b of the inverter 3, VDC). The inverter 3 includes switch elements (Sup, Sun, Svp, Svn, Swp, Swn) composed of three-phase bridges and supplies the DC voltage VDC supplied from the DC power supply 2 to 3 (Vu, Vv, Vw). When the inverter 3 applies the three-phase alternating current voltages (Vu, Vv, Vw), that is, the motor applying voltage to the alternating-current motor 1, , Iw) flows. The alternating-current motor 1 outputs torque corresponding to the three-phase alternating current (motor current) Iu, Iv, Iw output by the PWM control from the three-phase inverter 3.

When the 180-degree conduction means (control means) 6 supplies the pulse signal PWM-controlled by the 180-degree conduction to the inverter 3, the switching elements (Sup, Sun, Svp, Svn, Swp, Swn) are switched at the PWM controlled timing. Thereby, three-phase alternating current (motor current) Iu, Iv, Iw PWM-controlled by the inverter 3 is supplied to the AC motor 1.

At this time, the adjusting means 8 adjusts the driving frequency of the inverter 3 and the operating state of the AC motor 1 (for example, the rotational speed, torque, motor current, motor applied voltage, etc. of the AC motor 1) Therefore, the stop period of the 180-degree energized pulse signal output from the stopping means 7 is changed. The stopping means 7 stops the pulse signal of the predetermined phase phase of 180 degree supplied from the control means 6 to the inverter 3 by a period corresponding to the stop period set by the adjusting means 8, .

Therefore, the inverter 3 stops supplying the motor current to be supplied to the AC motor 1 for the stop period (i.e., the open phase section) of the current waveform that is energized 180 degrees by the PWM control. Thus, the AC motor 1 generates a base voltage due to magnetic saturation in the stop period (open phase section), so that the phase of the AC motor 1 is detected by the base voltage even without providing a position sensor .

Accordingly, it is possible to estimate the rotor phase with high precision by the base voltage due to magnetic saturation occurring in the stop period (open phase section) of the pulse signal. In addition, since the motor current is extremely small in the low speed range where the rotation speed is lower than the predetermined value, the current distortion can be suppressed even if the stop period of the pulse signal is set.

Hereinafter, the operation of each sub-element of the motor drive control apparatus 100 shown in Fig. 1 will be described in detail.

<Current Detection Means>

1, the current detecting means 4 detects the bus line current IDC flowing through the negative terminal 3b of the inverter 3 and detects a three-phase alternating current Motor current) Iu, Iv, Iw are extracted. However, current detecting means may be provided on each of the phase terminals 3c, 3d and 3e of the inverter 3 so as to directly detect the three-phase alternating currents Iu, Iv and Iw on the output side.

<Voltage Detecting Means>

The voltage detecting means 5 includes U-phase voltage detecting means 5a, V-phase voltage detecting means 5b and W-phase voltage detecting means 5c, Vva, and Vwa (see Fig. 2) of the phase terminals 3c, 3d, and 3e of the inverter 3, that is, the motor applied voltage,

Fig. 2 shows a detailed circuit configuration of the voltage detecting means 5 shown in Fig. As shown in Fig. 2, the U-phase voltage detecting means 5a includes a first voltage dividing resistor 5aa, a second voltage dividing resistor 5ab, and a switch means 5ac. On the other hand, the voltage dividing means for dividing the detected voltage is not limited to this. When the internal pressure of the later-described magnetic saturation phase estimating means 6a (see Fig. 16) is high, it is not necessary to provide a voltage dividing resistor to divide the voltage. The V-phase voltage detection means 5b and the W-phase voltage detection means 5c have the same configuration as the U-phase voltage detection means 5a.

2, the U-phase voltage detecting means 5a includes the switch means 5ac for blocking the voltage detecting means 5 and the inverter 3, thereby reducing the circuit loss of the voltage detecting means 5 . That is, when the switch means 5ac is ON, the U-phase terminal voltage Vua is divided and the divided voltage of the U-phase terminal voltage Vua is input to the magnetostrictive phase estimation means 6a. At this time, losses occur in the first and second voltage-dividing resistors 5aa and 5ab. Thus, when voltage detection is not performed, the circuit means of the voltage detection means 5 can be reduced by turning off the switch means 5ac.

That is, in the motor drive control apparatus shown in Fig. 1, the voltage detecting means 5 is provided with switch means (switch circuit) 5ac for separating from the terminals of the AC motor 1, as shown in Fig. 2 The voltage detection is unnecessary, and the power loss when the voltage detecting means 5 is not used can be reduced.

<Generation of Stop Period>

The waveforms of the U-phase voltage, the U-phase current and the pulse signal in the inverter 3 shown in Fig. 1 will now be described with reference to Fig. 3 (a). FIG. 3 (b) is a partially enlarged view of FIG. 3 (a). 3 (a), the vertical axis indicates the neutral point of the alternating-current motor 1 as a reference potential. 3 (a), the U-phase voltage Vu is an ideal voltage waveform and is represented by a cosine function of the voltage phase? V. The upper and lower Un and U of the U-phase switch elements Sup and Sun based on the pulse signal of the PWM control outputted by the 180-degree energizing means 6 1). The ON / OFF waveform is ON when the signal level is "1", and OFF when it is "0". On the other hand, the switching characteristics of the switching elements Sup and Sun are ideal.

Here, it is assumed that both of the switching elements Sup and Sun are turned OFF by the PWM control, or only one of them is turned ON. The stop period Toff represents a period in which both of the switch elements Sup and Sun on U are stopped based on the stop signal of the stopping means 7 (see Fig. 1) to be described later. In this stop period Toff, the switch elements Sup and Sun are both stopped in the OFF state. That is, the stop period Toff becomes an open phase section. The period other than the stop period Toff is the energization period Ton that performs switching by PWM control of normal 180-degree energization.

Next, the waveform characteristics of the U-phase terminal voltage Vua of the inverter 3 will be described with reference to FIG. 3 (b) and FIG. 4 is a table showing the U phase terminal voltage Vua by dividing the case of the energizing period Ton and the stop period Toff. As shown in Fig. 4, in the energization period Ton, the U-phase terminal voltage Vua becomes VDC / 2 when the switch element Sup is ON, and -VDC / 2 when the switch element Sun is ON, / 2.

In the stop period Toff, the U-phase terminal voltage Vua is as follows. Immediately after the stop period Toff, as shown in Fig. 3 (b), a reflux period in which a reflux current flows in diodes connected in reverse parallel to the switch element Sup and the switch element Sun, (Tr). This reflux period Tr is a period during which the U-phase current Iu is refluxed to the diode element connected in reverse parallel to the switching element Sup or the switching element Sun. At this time, the U-phase terminal voltage Vua depends on the polarity of the U-phase current Iu.

4, when the U-phase current Iu is positive (Iu > 0), the diode elements in antiparallel to the switching element Sun are excited by the return current The negative terminal 3b and the U-phase terminal 3c of the inverter 3 become the same potential and the U-phase terminal voltage Vua becomes -VDC / 2. When the U-phase current Iu is negative (Iu < 0), diode elements in antiparallel connection to the switching element Sup are conducted by the return current, and the positive terminal 3a of the inverter 3, And the U-phase terminal 3c become the same potential, and the U-phase terminal voltage Vua becomes VDC / 2.

When the U phase current Iu becomes zero (Iu = 0) and the reflux period Tr ends, the U phase terminal voltage Vua becomes ON / OFF of the switch elements Svp, Svn, Swp, and Swn Depending on the state. That is, when the switch element Svp and the switch element Swp are turned ON, the V-phase terminal 3d and the W-phase terminal 3e are electrically connected to the positive terminal 3b, Becomes VDC / 2.

Similarly, when the switch element Svn and the switch element Swn are turned ON, the V-phase terminal 3d and the W-phase terminal 3e and the negative-electrode side terminal 3b are electrically connected to each other and the U-phase terminal voltage Vua ) Becomes -VDC / 2. When the switch element Svp is ON and the switch element Swn is ON, the U phase terminal voltage Vua becomes the base voltage V0 by the magnetic saturation phenomenon. When the switch element Svn is ON and the switch element Swp is ON, the U phase terminal voltage Vua becomes equal to the base voltage V0a (i.e., when the base voltage V0 is measured) by the magnetic saturation phenomenon And the voltage is shifted by 180 degrees with respect to the voltage phase).

That is, in the stop period Toff during which the switch elements Sup and Sun of the upper and lower arms of the U phase are OFF together, when the V phase and the W phase are two-phase operation, Or the base voltage VOa is generated. Therefore, the rotor phase can be estimated by the base voltage V0 of the U phase or the base voltage VOa.

&Lt; Waveform at Driving by Practical Operation &gt;

Fig. 5 is a diagram showing a case where when the motor drive control device 100 of the first embodiment is driven by the two-phase modulation type PWM control method, the stop period Toff of the U phase pulse signal in the vicinity of the zero- And waveforms of voltage, current, and pulse signal when the real time is driven by setting the stop period Toff of the U-phase pulse signal. Voltage level on the horizontal axis, voltage level on the vertical axis, current level, and pulse level. However, the U phase voltage has the neutral point of the AC motor 1 as the reference potential.

5 (a) shows the U-phase terminal voltage Vua of the inverter 3, Fig. 5 (b) shows the U-phase current Iu flowing in the AC motor 1, The pulse signal Up of the switching element Sup of the switching element 3 and the pulse signal Un of the switching element Sun of the inverter 3 are shown.

As shown in (c) of FIG. 5, it can be confirmed that the pulse signals Up and Un are OFF in the stop period Toff and the intervals in which the pulse signals Up and Un are stopped are set have. Further, since the section in which the pulse signal is stopped is set, it is also confirmed that the U phase current Iu becomes zero in the section of the stop period Toff.

<Voltage due to magnetic saturation>

Here, the magnetic saturation phenomenon will be described. Fig. 6 is an image view when detecting the phase terminal voltages of the alternating-current motor 1 shown in Fig. 1. When the switch element Svp and the switch element Swn are ON (see Fig. 4) Terminal voltages Vua, Vva, and Vwa. In this mode, the V-phase terminal voltage Vva is VDC / 2, the W-phase terminal voltage Vwa is -VDC / 2, and the U-phase terminal voltage Vwa is ideally, from the symmetry of the circuit configuration of the AC motor 1, The voltage (Vua) will be O. However, the inductance of each phase of the alternating-current motor 1 is not uniform because it is affected by the rotor phase? By magnetic saturation. Therefore, as shown in Fig. 6, the U-phase terminal voltage Vua is generated as the base voltage V0. Similarly, when the switch elements Svn and Swp are ON, the U-phase terminal voltage Vua generates the base voltage VOa (see FIG. 4).

Fig. 7 is a characteristic diagram showing the relationship between the rotor phase and the electromotive force of the alternating-current motor 1 shown in Fig. 1, in which the rotor phase? Is shown on the horizontal axis and the electromotive voltage V0 is shown on the vertical axis. It is known that the base voltage V0 of the alternating-current motor 1 shown in Fig. 7 is a periodic function of twice the rotor phase [theta]. Using the phase relation, The rotor phase? Of the alternating-current motor 1 can be estimated from the detected value of the electromotive voltage V0 of the alternator 5a.

&Lt; 180 degree conduction means &gt;

1, the 180-degree energizing means 6 includes a self-saturation type phase estimation means 6a, an organic voltage type phase estimation means 6b, a phase estimation switching means 6c, a voltage command means 6d, PWM control means 6e and a speed estimation means 6f and is provided with PWM control signals (pulse signals) of the switching elements Sup to Swn of each phase of the three phases of the inverter 3, The speed estimation value?

The magnetic saturation type phase estimation means 6a calculates the phase difference between the detected value of the base voltage V0 and the phase difference between the detected value of the base voltage V0 and the base voltage V0 in accordance with the relationship between the rotor phase? And estimates the rotor phase &amp;thetas; Hereinafter, such a phase estimation method is referred to as a magnetic saturation type phase estimation method. The advantage of this magnetic saturation phase estimation method is that the rotor phase? Can be estimated from the detected value of the base voltage V0 in the low speed range including the stopping of the AC motor 1. This is because the U-phase base voltage V0 is generated every time the V-phase and the W-phase are energized regardless of the rotation speed. On the other hand, the disadvantage of the self saturation phase estimation method is that the current waveform is distorted because no current flows in the stop period Toff as shown in Fig.

The induced voltage phase estimation means 6b estimates the rotor phase? On the basis of the three-phase alternating currents Iu, Iv and Iw extracted from the current detection means 4 in the energization period Ton . In the energization period Ton, since the U-phase current Iu flows in a sinusoidal waveform by the ordinary 180-degree energization, the rotor phase? Can be estimated similarly to the 180-degree energization method using the induced voltage. Hereinafter, such a phase estimation method is referred to as an organic voltage phase estimation method. An advantage of this organic voltage phase estimation method is that current distortion is small because the stop period Toff is not required. On the other hand, the disadvantage is that since the induced voltage becomes smaller as the speed becomes lower, the phase estimation precision of the rotor is lowered in the low speed range.

The phase estimation switching means 6c switches between the magnetic saturation type phase estimation method and the organic voltage type phase estimation method on the basis of the energization period Ton or the stop period Toff, (? a). For example, when the rotational speed of the AC motor 1 is higher than the predetermined value (middle / high speed position), the system is switched to the induced voltage type phase estimation system. When the rotational speed of the AC motor 1 is lower than a predetermined value , It switches to the magnetic saturation type phase estimation method.

The voltage command means 6d calculates the command values Vu *, Vv * and Vw * of the three-phase AC voltages Vu, Vv and Vw on the basis of the phase estimated value? A of the rotor phase? do. Then, the command values (Vu *, Vv *, Vw *) are transmitted to the PWM control means 6e.

The PWM control means 6e converts the voltage command values Vu *, Vv * and Vw * obtained from the voltage command means 6d into a PWM control signal for 180-degree conduction based on the PWM control. This PWM control signal is a pulse signal whose ON / OFF duty is controlled, and performs PWM control by switching each switch element (Sup to Swn) of the inverter 3.

The speed estimating means 6f estimates the rotational speed omega of the alternating-current motor 1 by pseudo-differentiating the estimated phase angle? A that is the estimated value of the rotor phase and outputs the estimated value of the rotational speed omega And outputs the speed estimation value? A to the adjustment means 8. [

<Stop Means>

The stopping means 7 shown in Fig. 1 outputs a stop signal for stopping the pulse signal of the PWM control outputted from the PWM control means 6e to each of the switch elements Sup to Swn. The stop signal output from the stop means 7 can stop the switch elements Sup to Swn in preference to the PWM control pulse signal output from the PWM control means 6e. Therefore, even if the pulse signal is output from the PWM control means 6e all the 180 seconds, the pulse signal of the predetermined phase is stopped during the stop period Toff set by the stop means 7.

<Adjusting Means>

Next, a description will be given of methods [1] to [5] in which the adjustment means 8 shown in Fig. 1 adjusts the stop period Toff of the stop signal outputted from the stop means 7. Fig.

Prior to describing each method, the premise common to each method will be described.

First, the relationship between the stop period Toff of the pulse signal and the current distortion? Will be described. 8 is a characteristic diagram showing the relationship between the stop period Toff and the current distortion? Of the pulse signal of the inverter 3 shown in Fig. 1, in which the stop period Toff is shown on the horizontal axis, ε). However, in order to simplify the explanation, it is assumed that the current waveform is distorted only by the stop period Toff.

As shown in Fig. 8, the current distortion epsilon when the stop period Toff is 60 degrees is epsilon 1 and the current distortion epsilon when the stop period Toff is the electrical angle O is O. Here, the electrical angle of 60 degrees is a value representing the stop period (Toff) when the 120-degree energization type using magnetic saturation is used, and the electrical angle O is the stop period when using the induced voltage type 180-degree energization method Toff).

Therefore, the reason why the stop period Toff when the magnetic saturation-use type 120-degree energization method is used will be 60 electrical degrees will be described with reference to Fig. Fig. 9 shows a switching waveform in a 120-degree energization type inverter using magnetic saturation. As shown in Fig. 9, in the 120-degree energization method, any one of the phases is always in the stop period Toff. For example, as shown in Fig. 9, the U phase + side and the W phase-side are energized in the range of 0 to 60 degrees in electric angle, and the upper and lower arms of the V phase are in the stop period Toff. In the electrical angle range of 60 to 120 degrees, the V phase + side and the W phase-side are energized, and the upper and lower arms of the U phase are the stop period Toff.

Therefore, when the stop period Toff is set to 60 degrees or more, electric currents of two or more phases are stopped at the same time, so that no current flows through all three phases of the alternating-current motor 1, Torque can not be output. Therefore, in order to avoid such a problem, the stop period Toff is 60 degrees or less in electrical angle. On the other hand, if the respective stop periods Toff are not made uniform, it is possible to set the stop period Toff of any one phase to 60 degrees or more, but this is not preferable because the symmetry of the voltage and current waveform is lost. That is, the stop period Toff when the magnetic saturation-use type 120-degree energization method is used is 60 electrical degrees.

Returning to Fig. 8, in order to suppress the current distortion?, It is only necessary to use the 180-degree energization method using the induced voltage. However, since the induced voltage is low in the low speed range, the 180-degree energization method using the induced voltage can not realize the position detection of the rotor by the 180-degree energization method using the induced voltage.

8, the relationship between the current distortion? And the stop period (Toff) of the pulse signal is shown by adding the rotational speed? Of the alternating-current motor 1 as a variable as shown in Fig. 10, when the rotational speed [omega] is less than [omega] 0, the magnetic saturation use type 120-degree energization method is executed. When the rotational speed [omega] It can be considered as a comparative example of the present embodiment that the 180-degree energization method is switched. In this comparative example, when the transition from the high-speed range H to the low-speed range L is made, the current is rapidly switched from the 180-degree energization to the 120-degree energization, ε) increases rapidly.

Therefore, in the method [1], in order to reduce the current distortion?, As shown in Fig. 3, the stop period Toff in which both the switch elements Sup and Sun of the U- A new energizing method is set up in which the section of the phase is set. In this new energization method, it is assumed that the stop period Toff is adjusted based on the rotation speed estimation value? A of the alternating-current motor 1. This makes it possible to frequently change the magnitude of the current distortion? In accordance with the rotational speed? Of the alternating-current motor 1, as indicated by the curve Q in FIG. 10, The current distortion? Can be suppressed as compared with the case of the comparative example. 1, the adjustment means 8 adjusts the stop period Toff output from the stop means 7 based on the rotation speed estimation value? A outputted from the speed estimation means 6f do.

In other words, in consideration of the characteristics of each phase estimation method, the adjustment means 8 increases the stop period Toff output from the stop means 7 as the rotational speed? Of the AC motor 1 becomes lower Therefore, the magnetic saturation type phase estimation method is preferentially used. However, the stop period Toff at this time is set to be equal to or less than the stop period of the 120-degree energization type using magnetic saturation, that is, the electrical angle is 60 degrees or less. That is, the stop period Toff of the pulse signal by the 180-degree energization method is increased in the range of the electric angle of 60 degrees or less as the rotational speed, torque, motor current, and motor applied voltage of the AC motor 1 become lower . On the other hand, when the rotational speed [omega] of the alternating-current motor 1 is high, an organic voltage type phase estimation method is used.

Further, in the method [2], the adjustment means 8 is configured such that, as the torque of the alternating-current motor 1, the three-phase alternating current / voltage of the inverter 3 (that is, the motor current, The period Toff may be expanded and the magnetic saturation type phase estimation method may be preferentially used. This is because the energizing period of the bus current I D is short and the three-phase alternating currents Iu, Iv and Iw can not be accurately extracted under the condition that the stop period Toff is set to 60 degrees or less This is because the estimation precision in the organic voltage phase estimation method is lowered.

In addition, in the method [3], the adjusting means 8 is configured so that the rotational speed? Of the alternating-current motor 1 is higher or the torque of the alternating-current motor 1, the three- As the voltage (i. E., Motor current, motor applied voltage) or the like is increased, the stop period Toff is reduced and the organic voltage phase estimation method is preferentially used. The reason is that the three-phase alternating currents Iu, Iv and Iw can be accurately extracted because the energizing period of the bus current I D is long, contrary to the case of the method [2] This is because the estimation precision of the method is not degraded. On the other hand, the stop period Toff may be minimized and reduced to zero.

That is, the stop period Toff of the 180 energized pulse signal is reduced or made zero as the rotational speed? Of the AC motor 1, the torque, the motor current, and the motor application voltage become higher. As a result, the current distortion? Can be reduced and the efficiency of the AC motor 1 can be improved.

On the other hand, in the method [4], the adjustment means 8 may change the stop period Toff in accordance with the drive frequency of the inverter 3 in order to improve the efficiency of the AC motor 1. [ For example, the stop period Toff is shortened as the drive frequency of the inverter 3 becomes higher and the stop period Toff is made longer as the drive frequency of the inverter 3 is lowered.

In the method [5], the adjusting means 8 may set the stop period Toff such that the U-phase current Iu includes a timing at which the U-phase current Iu becomes zero (zero cross point). 11 is a waveform diagram of an ideal U-phase current Iua in the inverter 3 shown in Fig. 1, that is, a waveform diagram of an ideal U-phase current Iua ignoring the effect of the stop period Toff. 11, the U phase current Iua is a cosine function of the current phase θi, and the points P1 and P2 are the timings at which the U phase current Iua becomes zero (zero cross point) Respectively. Therefore, if the stop period Toff is set so as to include the zero-cross points P1 and P2, the U phase current Iu at the start of the stop period Toff in Fig. The absolute value IuO becomes small (close to zero).

At this time, since the reflux period Tr is also shortened in FIG. 3 (b), it is possible to detect the base voltage VOO at the time immediately after the stop period Toff as well as the base voltage V0 So that the detection timing of the rotor phase can be made faster and the phase estimation precision can be improved. Further, after the detection of the base voltage VOO immediately after the stop period Toff, the cancellation of the cancellation period Toff immediately may narrow the cancellation period Toff and suppress the current distortion. On the other hand, the dead time set for short-circuit prevention of the upper and lower switch elements of the same phase is equivalent to the stop period Toff. Therefore, the U phase current Iu may be detected during the dead time, and if it is less than the threshold value, the dead time may be extended and the base voltage V0 may be detected.

That is, the adjusting means 8 preferably sets the stop period Toff of the 180-degree energizing pulse signal so as to include a period in which the motor current of the alternating-current motor 1 becomes zero. 3 can be shortened and the detection timing of the rotor phase can be made faster by the base voltage VOO immediately after the stop period Toff so that the stop period Toff can be narrowed have. As a result, the current distortion can be further reduced.

<Winding variable means of AC motor>

The adjusting means 8 may adjust the stop period Toff on the basis of the magnetic flux amount of the rotor of the alternating-current motor 1 or the number of windings of the stator. 12 shows the concept of the alternating-current motor 1 when the alternating-current motor 1 is provided with the winding variable means and the magnetic flux amount variable rotor. 12, the AC motor 1 includes U-phase winding variable means 1a, V-phase winding variable means 1b and W-phase winding variable means 1C, The magnetic flux amount variable rotor 1d may be connected with the means 1a, 1b, 1C by varying the winding number of each winding of the alternating-current motor 1.

It is known that the 180-degree energizing means 6 shown in Fig. 1 can vary the induced voltage coefficient or inductance of the alternating-current motor 1 in accordance with the operating state. This is to increase the operating range of the AC motor 1 and is generally used in applications where a large torque is required, for example, in an automobile or a washing machine at a low speed. Therefore, by applying this technique, a configuration including the U-phase, V-phase and W-phase winding varying means 1a, 1b, 1c and the magnetic flux control variable rotor 1d as shown in Fig. 12 can be realized .

At this time, if the induced voltage coefficient or the inductance is increased, the magnetic saturation phenomenon becomes strong, so that the phase estimation accuracy of the magnetic saturation type phase estimation method is improved. As a result, the adjustment means 8 can narrow down the stop period Toff, and as a result, the current distortion can be suppressed to be small. Further, when the induced voltage coefficient is increased, the current value required for outputting the same torque becomes smaller, and as a result, the absolute value of the current distortion can be suppressed. Further, when the inductance is increased, the harmonic component of the current distortion is suppressed, so that the iron loss of the alternating-current motor 1 can be reduced.

That is, when the magnetic flux amount of the rotor of the alternating-current motor 1 can be freely changed, the adjusting means 8 adjusts the stopping period of the pulse signal of the 180-degree energizing method in accordance with the magnetic flux amount of the alternating- The alternating-current motor 1 can be stably driven by changing the output voltage Toff.

When the number of turns of the windings of the stator of the alternating-current motor 1 is switched, the adjusting means 8 sets the stop period (Toff) of the 180-degree energizing pulse signal in accordance with the number of windings of the alternating- The AC motor 1 can be stably driven.

As described above, as the control system of the inverter 3 that performs the PWM control, the first (first) control unit including the current detecting unit 5, the 180-degree energizing unit 6, the stopping unit 7, According to the configuration of the motor drive control apparatus 100 of the embodiment, the rotor phase is estimated by using the 180-degree conduction method using the induced voltage when the rotational speed is higher than the predetermined rotational speed (middle / high speed rotation range) At a speed lower than or equal to the rotation speed (low speed range), the stop period Toff corresponding to the operating conditions of the AC motor 1 is set, and the rotor phase is estimated by the electromotive force by magnetic saturation. Thus, it is possible to perform the position sensorless control from the low speed range including the stopping of the AC motor 1 to the high speed range while suppressing the current distortion to the minimum.

&Quot; Second Embodiment &

Next, as a second embodiment, an air-conditioning apparatus 10 using the motor-drive control apparatus 100 of the first embodiment will be described with reference to Figs. 13 and 14. Fig. On the other hand, the description of the same contents as in the first embodiment will be omitted. 13 shows a configuration diagram when the motor drive control apparatus 100 according to the first embodiment is applied to the control of the AC motor 1 for driving the compressor 9 of the air conditioner 10. Fig.

On the other hand, in Fig. 13, the compressor 9 is used as a driving source of a thermal cycle in the air-conditioning equipment 10. [ 13 denote inverters 3, current detecting means 4, voltage detecting means 5, 180-degree energizing means 6, stop means 7, And adjustment means (8).

14 shows the waveforms of the torque ripple and the motor current effective value of the alternating-current motor 1 that drives the compressor 9 and shows the time on the horizontal axis and the load torque? Of the compressor 9 on the vertical axis. And the current effective value I1 of the alternating-current motor 1 are shown. As shown in Fig. 14, the load torque? Of the compressor 9 is a torque ripple having a periodicity because the operation cycle of the compressor 9 repeats compression and expansion at regular intervals. At this time, the method of stabilizing the rotational speed [omega] of the alternating-current motor 1 is, for example, a method of reversing the phase of the torque ripple By using the torque ripple suppression control for canceling the ripple component of the torque by flowing the torque current to the alternating-current motor 1.

14, the current effective value I1 of the alternating-current motor 1 also pulsates in synchronization with the load torque tau. This is for canceling the pulsation of the load torque? By the torque pulsation of the AC motor 1. In Fig. 14, the points P1 to P3 are minimum values of the current effective value I1 of the alternating-current motor 1 in each pulse cycle. However, if there are two or more minimum values (not shown) of the current effective value I1 of the alternating-current motor 1 for every pulse period, an open phase (stop period Toff) may be set for each of the minimum values.

That is, in the second embodiment, the adjusting means 8 may be set so that the plurality of stop periods Toff include the minimum points P1 to P3 of the motor current effective value I1, respectively. With such a setting, the current distortion occurs only when the motor current effective value I1 is small, but it is possible to suppress the change of the torque due to the current distortion and suppress the interference to the torque ripple suppression control. Thus, the compressor 9 can be stably driven at a low speed. As a result, the air conditioner 10 using the motor drive control device 100 performing such control can attain wide output and high efficiency.

Since the compressor 9 used in the air-conditioning equipment 10 has a large torque ripple, the motor current effective value I1 pulsates. In this case, however, the stop period Toff of the 180- It may be set to include a period in which the current effective value of the alternating-current motor becomes extremely small. On the other hand, when there are two or more minimum values of the current effective value of the alternating-current motor, the stop period Toff may be set corresponding to each minimum value.

On the other hand, when one period of the mechanical system (mechanical system) of the air conditioner 10 is different from one period of the electric angle of the alternating-current motor 1, the adjusting means 8 sets the stopping period Toff, It is appropriate to set a period in which the minimum value of the rms value of the motor current exists every one cycle of the motor current.

&Quot; Third Embodiment &

In the third embodiment, the case of restarting the idling AC motor 1 will be described with reference to Figs. 15 and 16. Fig. Fig. 15 shows the relationship of the vector of the three-phase alternating-current voltage of the alternating-current motor driven by the motor drive control device shown in Fig. Fig. 16 shows a configuration example of phase estimation at the time of free-run restart in the 180-degree energizing means 6 shown in Fig. Descriptions overlapping with those of the first embodiment will be omitted. However, if there is no external force, the AC motor 1 is stopped as long as all the switching elements (Sup to Swn) of the inverter 3 are stopped.

When an external force is applied to the alternating-current motor 1, the alternating-current motor 1 starts rotating (revolving) even when the inverter 3 is stopped. Then, as shown in Fig. 15, an induced voltage Vω is generated in the alternating-current motor 1 in accordance with the rotational speed?. The U-phase, V-phase, and W-phase components of the induced voltage Vω at this time are Vωu, Vωv, and Vωw, respectively. Further, the organic voltage phase [omega] is an angle formed by the organic voltage V [omega] and the U-phase direction. The d-axis represents the rotor direction, and the phase difference between the organic voltage phase?? and the rotor phase? is 90 degrees.

Here, the state in which the AC motor 1 is stably started in a state in which the AC motor 1 is rotating (idling) is referred to as a free run restart. In this free-run restart, it is necessary to estimate the rotor phase [theta] before start-up in order to prevent start-up shock.

At this time, a method of estimating the rotor phase? Before the start of the AC motor 1 will be described using the configuration shown in Fig. Until the free run restart operation, there is no energized phase. Therefore, the bus line current IDC does not flow, and the organic voltage phase estimation means 6b of the 180-degree conduction means 6 can not be applied to the phase estimation. Further, since the base voltage obtained from the voltage detection means 5 is only the W-phase induced voltage V ?, the magnetic saturation type phase estimation means 6a can not be applied. Therefore, the free-standing type phase estimation unit 6g is used to estimate the rotor phase? From the W-phase induced voltage V ?.

As described above, the free-run type phase estimation unit 6g can estimate the rotor phase? Because the phase difference between the induced voltage phase?? And the rotor phase? In Fig. 15 is 90 degrees have. That is, as disclosed in Japanese Patent Application Laid-Open No. 2005-137106, by using the technique of detecting the rotor phase from the induced voltage phase by magnetic saturation, the rotor phase? Of the AC motor 1 is estimated can do. After the estimation of the rotor phase [theta], the 180-degree energization type using the induced voltage is started.

That is, the phase estimation switching means 6c switches the phase estimation method from the free-run type phase estimation means 6g to the organic voltage type phase estimation means 6a. However, when the rotational speed is low, since the induced voltage Vω is small, the estimation accuracy is low in the method of estimating the rotor phase from the organic voltage phase, so that a starting shock occurs due to the detection error of the rotor phase, Failure or failure to restart.

Thus, in the third embodiment, after estimating the rotor phase? Using the free-run type phase estimation unit 6g, the phase estimation system is switched to the magnetic saturation type phase estimation unit 6b. At the same time, the magnetic saturation type 180 degree energization method is started with the phase of the absolute value of the voltage detection value being the minimum as the stop period Toff, thereby reliably restarting the free run while suppressing the start shock.

In Fig. 15, for example, when the voltage phase? Is 0???? / 3, | V? U |> | V? V | and | V? W |> | V? V |

That is, the absolute value of the V-phase induced voltage V? V is the smallest among the three phases of the U-phase, V-phase and W-phase.

Thus, the stopping means 7 sets the stop period Toff in the V-phase and starts energization between the U-phase and the W-phase. By stopping the phase in which the absolute value of the voltage value is the smallest, it is possible to suppress the phase difference ?? between the motor voltage V1 and the induced voltage V? .

That is, when the alternating-current motor 1 is revolving, the stopping means 7 outputs a pulse signal of an absolute value of the voltage detection value detected by the voltage detecting means 5 to a minimum value in a predetermined stop period Toff, . Thus, the 180-degree energizing means (control means) 6 can start the AC motor 1 by PWM-controlling the inverter 3 by an image pulse signal not corresponding to the stop period Toff. Thereby, there is no fluctuation at the start of the AC motor 1, and smooth starting can be performed.

As described above, the motor-drive control apparatus 100 of the present embodiment and the air-conditioning equipment 10 using the motor-drive control apparatus 100 can control the DC power supplied from the DC power supply 2 by the PWM control to the desired AC power A current detecting means 4 for detecting a current flowing through the AC motor 1 and a current detecting means for detecting a current flowing through the terminal of the AC motor 1, A control means (8) for PWM-controlling the inverter by outputting a pulse signal of 180-degree conduction, and a control means (8) for outputting a pulse signal of a predetermined phase (7) for stopping the AC motor (1) for a predetermined stop period, an adjusting means (7) for variably adjusting a stop period set in the stop means (7) in accordance with an operating condition when the AC motor (8). The motor drive control apparatus 100 and the air conditioner 10 can detect the position of the rotor from the low speed range to the high speed range including the stopping state and the current distortion is small, , It is possible to perform the position sensorless control in which the detection of the neutral point potential is unnecessary.

While the embodiments of the motor drive control device 100 and the air conditioner 10 according to the present invention have been described in detail, the present invention is not limited to the above-described embodiments, Various modifications are possible without departing from the scope.

That is, the present invention is not limited to the contents of the above-described first to third embodiments, and various modifications are possible. In other words, each of the above-described embodiments is a detailed illustration for explaining the contents of the present invention in an easy-to-understand manner, and is not necessarily limited to those having all the configurations described above. Note that some of the configurations of the embodiments may be replaced with those of the other embodiments, and the configurations of the other embodiments may be added to the configurations of any of the embodiments.

It is also possible to add, delete, or replace the configurations of other embodiments with respect to some of the configurations of the embodiments. The above-described components, functions, processing units, processing means, and the like may be realized by hardware, for example, by designing a part or all of them with, for example, an integrated circuit. Further, the above-described respective structures, functions, and the like may be realized by software by analyzing and executing a program realizing the respective functions of the processor. Information such as a program, a table, and a file for realizing each function may be recorded in a recording device such as a memory, a hard disk, an SSD (solid state drive), or an IC (integrated circuit) card, an SD card, ) Or the like. In addition, the control lines and information lines indicate that they are deemed necessary for explanation, and not all control lines and information lines are necessarily indicated on the product. In practice, almost all configurations may be considered to be interconnected.

According to the present invention, not only the motor drive control device for driving and controlling the AC motor used in the air conditioner but also the motor drive control device for driving the AC motor used in the household appliance such as the refrigerator, the washing machine and the electric vacuum cleaner Can be used.

1: AC motor
la: U-phase winding variable means
lb: V-phase winding variable means
lc: W phase winding variable means
1d: magnetic flux variable rotor
2: DC power source
3: Inverter
3a: Positive electrode terminal
3b Negative electrode terminal
3c: U phase terminal
3d: V phase terminal
3e: W phase terminal
4: current detection means
5: Voltage detection means
5a: U-phase voltage detection means
5aa: first partial pressure resistance
5ab: second partial pressure resistance
5ac: switch means
5b: V-phase voltage detection means
5c: W-phase voltage detection means
6: 180 degree energizing means (control means)
6a: Magnetic saturation type phase estimation means
6b: organic voltage type phase estimation means
6c: phase estimation switching means
6d: voltage command means
6e PWM control means
6f: velocity estimation means
6g: free-run type phase estimation means
7: Stopping means
8: Adjustment means
9: Compressor
10: Air conditioner
100: Motor drive control device
VDC: DC voltage
IDC: Bus current
Sup, Sun, Svp, Svn, Swp, Swn: Switch element
Vu, Vv, Vw: U phase voltage, V phase voltage, W phase voltage
Vua, Vva, Vwa: U phase terminal voltage, V phase terminal voltage, W phase terminal voltage
V1: Motor voltage
Iu, Iv, Iw: U phase current, V phase current, W phase current
I1: Motor current
Vω: induced voltage
Vωu, Vωv, Vωw: U phase component of the induced voltage, V phase component, W phase component
Ton: Energizing period
Toff: Stop period
Tr: reflux period
VO, VOO, VOa: Voltage
θ: rotor phase
? a: phase estimation value
θv: voltage phase
θi: current phase
θω: Organic voltage phase
ω: rotational speed
ωa: velocity estimate
τ: Load torque

Claims (13)

An inverter for converting DC power supplied from a DC power source into desired AC power by PWM control and supplying the AC power to the AC motor,
Current detecting means for detecting a motor current flowing in said alternating-current motor;
Voltage detecting means for detecting a motor-applied voltage applied to a terminal of the alternating-current motor;
Control means for PWM-controlling the inverter by outputting a 180-degree pulse signal,
A stop means for stopping a pulse signal of a predetermined phase among the pulse signals by a predetermined stop period,
And an adjusting means for variably adjusting a stop period set in the stop means in accordance with an operation state when the AC motor is at a predetermined rotation speed or lower. The method according to claim 1,
Wherein the adjusting means variably adjusts the stop period so as to include a period in which the motor current becomes zero in one cycle of electrical angle (electric angle), and sets the stop period to the stop means controller. The method according to claim 1,
Wherein said adjusting means variably adjusts the stop period so as to include a period in which an effective value of the motor current becomes a minimum value in each one electrical cycle and sets the stop period in the stop means, Device. The method of claim 3,
Wherein the adjusting means sets a plurality of stop periods in the stopping means in accordance with each of the minimum values when there are a plurality of minimum values of the rms values of the motor current in one electrical cycle of the motor, Device. The method according to claim 1,
Wherein the adjusting means increases the stop period within a range of 60 degrees or less in electrical angle as at least one of the rotational speed, the torque, the motor current, and the motor applied voltage of the alternating current motor is lowered, And the motor drive control means sets the motor drive control means to the motor drive control means. The method according to claim 1,
The adjusting means sets the stopping period to the stopping means with decreasing or decreasing the stopping period as at least one of the rotational speed of the alternating-current motor, the torque, the motor current and the motor-applied voltage becomes higher And the motor drive control device. The method according to claim 1,
Wherein the adjusting means variably adjusts the stop period in accordance with the drive frequency of the inverter and sets the stop period in the stop means. The method according to claim 1,
Wherein said voltage detecting means comprises switching means for turning on and off the connection state of said terminal of said AC motor and said voltage detecting means. The method according to claim 1,
When the AC motor is idling,
The stopping means stops the pulse signal whose absolute value of the voltage detection value detected by the voltage detecting means becomes the minimum value for a predetermined stop period,
Wherein the control means PWM-controls the inverter by an image pulse signal not corresponding to the stop period and starts the AC motor. The method according to claim 1,
When the alternating-current motor is capable of arbitrarily varying the magnetic flux amount (magnetic flux amount)
Wherein the adjusting means variably adjusts the stop period according to the magnetic flux amount and sets the stop period in the stopping means. The method according to claim 1,
Wherein the adjusting means adjusts the stop period variably according to the number of turns of the winding and sets the stop period in the stopping means. An air-conditioning apparatus driven by the motor drive control apparatus according to any one of claims 1 to 11. 13. The method of claim 12,
Wherein the adjusting means adjusts the stopping period to an effective value of the motor current for every one period of the electric angle when one period of the mechanical system of the air conditioning equipment differs from one period of the electric angle of the alternating current motor , And sets the stopping period to the stopping means.

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