JPS63268488A - Variable speed drive device of induction motor - Google Patents

Abstract

PURPOSE:To set the angular position of axis of flux accurate operated by a current model and to improve the variable speed control performance of a motor, by bringing the rotor resistance set point close to the true value of the rotor resistance of the motor. CONSTITUTION:The first operational means 21 operated the component orthogonal to the primary current vector of the primary voltage vector from the primary voltage vector detected from the terminal of a motor, while the second operational means 22 operates the component orthogonal to the primary current vector of the primary voltage vector of a motor from a torque command value, a flux command value, a rotating speed and a motor constant. An adjustment means 23 revises the set point of the rotor resistance, one of the motor constants used in the operation by the current model and the second operational means 22 or any other intermediate operational value obtained in the course of operation process such as a slip frequency introduced from it so that the operational result by the second operational means 22 may coincide with the operational result of the first operational means 21.

Description

Detailed Description of the Invention [Industrial Application Field] The present invention provides a torque command value for variable speed drive of an induction motor powered by a power converter that can control output voltage magnitude, two frequencies, and a phase. , Magnetic flux command value 9Using a current model that calculates the angular position of the magnetic flux axis from the rotational speed and motor constant, divide the motor's primary current vector into a component parallel to the magnetic flux axis and a component perpendicular to it, and calculate both components. The present invention relates to a variable speed drive device for an induction motor, which includes control devices that control each other independently.

[Conventional technology]

FIG. 1 is a block diagram showing an embodiment of the variable speed drive device for an induction motor according to the present invention, which corresponds to the conventional embodiment except for the parts framed by broken lines. In Figure 1, l represents terminals a, b,
A three-phase induction motor with c, 2 is a power converter with controllable output voltage magnitude 1 frequency and phase. As the power converter 2, for example, a PWM (PWM) range inverter controlled by a pulse width modulation circuit 3 is used.

Two phase currents i□+'CI supplied to the motor 1 are detected by current transformers 41 and 42, and adjusted by the action of the current regulator 5 to match the current command value 1ml"+tct9. , the remaining phase current is IC+=(111
+"+lel"). 6 is a speed sensor, 7 is a speed setter, and 8 is a speed adjuster. The speed regulator 8 receives the output of the speed sensor 6 as an actual speed value, and in order to match this with the speed command value N9 from the speed setting device 7,
For example, Pltli clause calculation is applied to the speed control deviation (N"-N), and this is output as a torque command value. 9 is the command value for the magnitude of magnetic flux, which is equal to 2, of the actual speed value N from the speed sensor 6. This is a magnetic flux command device that outputs as a function.The magnetic flux command value A2 is constant below the base speed, but decreases inversely proportionally as the speed increases above the base speed.

The torque command value τ and the magnetic flux command value v2 are derived from the part surrounded by the broken line. This part contains a current model that is important for vector control of induction motors.

This type of vector control has already been published in many documents and is well known, so its principle will be briefly touched upon here.

Vector control of an induction motor is based on the motor current.

Considering voltage, etc. as space vector quantities, and observing these quantities from above the stator windings, they become alternating current quantities. Therefore, by observing these quantities from above the rotating magnetic field of the motor, we can calculate the values parallel to this magnetic field (magnetic flux axis). By decomposing it into a component and a component orthogonal to this component, it is attempted to control each component independently while treating both components as DC amounts.

FIG. 2 is a space vector diagram of the primary current I of the induction motor. The α-β coordinate system is a coordinate system whose origin is assumed to be on the stator with the center of rotation of the motor, and the α-axis is a fixed axis aligned with the winding axis of one phase, for example, and the β-axis is a fixed axis perpendicular to this. It is the axis. On the other hand, the M-T coordinate system is a rotating coordinate system in which the origin is the center of motor rotation, the M axis is in the direction of the magnetic flux vector, and the T axis is orthogonal to the M axis. The angle that the M axis (direction of the magnetic flux vector) makes with the α axis is indicated by φ, and the primary current vector IOM axis component (magnetizing current) is i, 4. T-axis component (torque current)
is indicated by ia. The primary current vector is at an angular position ε with respect to the α axis and at an angular position θ with respect to the M axis.

The magnitude of the magnetic flux is A2, and the slip angle frequency is ω3. Then, the following current model equations (11 to (3)) express the relationship between these and the above 'MELT using motor constants.

φ=Jω, dt=f(ω2+ωs t ) a t
'------...-(21 However, R1 secondary resistance, M: mutual inductance, 12: secondary leakage inductance, T!: secondary circuit time constant (= (M+xz)/Rz),
ωl: rotational angular velocity of magnetic flux, ω2: rotor angular velocity,
Unless otherwise specified, all secondary quantities are converted to primary values. On the other hand, the torque τ generated by the electric motor is τ=K・K2・iT −・・・−−m−−−
−・−・−・It is expressed as (4) (K is a proportionality constant). An example of realizing the relationships of equations (11 to (4) above) using a control circuit is the partial control circuit shown in FIG. The magnetic flux command value 'P2 from
is input. The arithmetic unit 11 uses the relationship in equation (11) above, and the divider 12 uses the relationship in equation (4) above to calculate i7=τ/A2 −・−・−・−・−−
--・-・river-(6) magnetizing current command value IM+) torque current command value i.

Calculate. Divider 13. Multiplier 14. The coefficient unit 15 and the secondary resistance setter 16 utilize the relationship of the above equation (3) to calculate the slip angular velocity ω3. Calculate (Rz=Rz.)
.

Slip angular velocity ω8. and the rotor angular velocity ω2 from the speed sensor 6 are added at the addition point 17, ω1=ω,, ω2 −−−・・−・−・−
・−・−+8) is obtained, and the integrator 18 further calculates φ=JωIdt 〜・−・−−−
---1------------- (9) Perform the integral operation to calculate the angular position φ of the magnetic flux vector.

The angular position φ of the magnetic flux axis in the α-β fixed coordinate system thus calculated in the partial phase is given to the coordinate converter (vector rotator) 20.

Assuming that the α-axis of the α-β fixed coordinate system in FIG. 2 is on the a-phase winding axis, the coordinate converter 20 converts the current command value 114+1? in the M-T coordinate system according to the following equation.
is the current command value ial”+ lc as the stator coordinate quantity
Convert to lm.

i+ =V/(iM”)”(it”)2−−−−−−
---QO) These current command values i□+Fl'' are guided to the current regulator 5 described above.

In the vector control of an induction motor using the current model according to the conventional technology described above, since the actual speed value N (rotor angular speed ω2L iM+iy and motor constant is used to calculate the magnetic flux, calculations can be performed even when the speed is zero, and the calculation is simple. A high-performance variable speed drive device can be realized with this configuration.

[Problem that the invention seeks to solve]

However, the motor constants, especially the secondary resistance value R2, vary significantly due to changes in rotor temperature (for example, in the case of copper, 1
(changes by 40% at 00°C), so if the setting value of R2 used as a parameter for magnetic flux calculation in the current model is kept at a constant value R2° regardless of the true value in the motor,
There is a problem in that a deviation occurs between the command value and the actual value of the magnetic flux, resulting in overvoltage and overcurrent during operation, making it impossible to continue normal operation (see 1981 National Institute of Electrical Engineers of Japan Conference No. 540).

Therefore, we use a voltage model that calculates magnetic flux from the voltage, current, and motor constants of the motor, so that the magnitude or angle of magnetic flux determined by the current model matches the magnitude or angle of magnetic flux determined by the voltage model. A method has been proposed to bring the set value of the secondary resistance closer to the true value of the motor by modifying the set value of the secondary resistance or the slip frequency in the current model (Japanese Patent Laid-Open No. 57-83184).
(See Japanese Patent Application Laid-Open No. 139692/1984, etc.).

However, this method also has problems. In order to explain this problem, a voltage model will be briefly explained.

The voltage model calculates the secondary flux linkage vector arc according to the following equation.

Sumire = J (beggar-R effect;) dt −−−−−・・−
------a However, l, secondary leakage inductance 12: secondary leakage inductance M: mutual inductance Lσ-1, +12・M/(M + l): motor voltage vector I: motor primary current vector :Electric motor-order magnetic flux linkage vector Figure 3 shows the model 2〇〇,@ in a block diagram.Among the motor constants, each inductance value is constant regardless of the temperature change of the conductor, but - Like the secondary resistance value R2, the second-order resistance value R varies greatly depending on the temperature.Therefore, the second-order resistance value R.

The percentage of voltage drop due to the rated speed is relatively small, about 10%
% speed or more, the magnetic flux can be calculated accurately, but at speeds below about 10% of the rated speed, the voltage model expressed by 0υ and (12) cannot calculate the correct magnetic flux. Therefore, at speeds below approximately 10% of the rated speed, even if the detected voltage value is large enough to perform integral calculations,
Since the corrective action using the voltage model cannot be effectively used, only the current model has to be used.

Therefore, an object of the present invention is to provide a means for accurately determining the magnetic flux vector even in the above-mentioned low speed range. - [Means for Solving the Problems] According to the present invention, the above object is achieved by the first calculation means 21 and the second calculation means 22, as shown in a broken line frame in FIG. ．． This is achieved by providing the In node means 23. That is, while the first calculation means 21 calculates the component v1δ orthogonal to the primary current vector I of the motor's primary voltage vector from the name of the primary voltage vector detected from the terminal of the motor, the second calculation means 22 is a torque command value like the current model.

Calculate from magnetic flux command value 1 rotation speed and motor constant.

The adjustment means 23 calculates the calculation result v by the second calculation means 22.
16m matches the calculation result v1δ of the first calculation means 21, the current model and the second calculation means use the set value of the secondary resistance, which is the motor constant used for calculation, or the slip frequency derived from this. The intermediate calculation values obtained during the calculation process are corrected.

[Effect]

From the voltage model Ni〇no and αmi shown earlier, is it the primary of the electric motor? ! The relational expression for pressure hector boar is found as follows.

Here, as shown in Fig. 2, the primary current vector is expressed as Let us consider the voltage vector characteristics. This T-δ coordinate system rotates at an angular velocity ωγ (=dε/dt'').

First, the α-jun expression is the component v1□ in the α-β coordinate system.

v1β, tlcrlttβ and v, (expressed using x and A2β, (p is the differential operator d/d t).

Further, if the magnitude of the primary current vector is i and the magnitude of the magnetic flux vector is v2, then from the vector diagram in FIG. 2, the following is obtained.

To convert the components v1° and V1β of the α-β coordinate system into the components V1rlVlδ of the γ-δ coordinate system, the following relationship may be used. From formulas 09 to 09, the δ-axis component of the primary voltage vector curvature V! Determining δ is as follows.

In FIG. 1, the second calculation means 22 calculates the motor constant, magnetic flux command value, torque command value, and rotor angular velocity according to the formula 01, as in the current model in part 1.
− Calculate the δ-axis component V1δ of the next voltage vector. Therefore, with the previous two (5) to (8) and Q, y θ= tan □ −−−１〜−−・−・・−
--------・r2ΦM By using the relationship, the value of the αm formula can be found. In this way, the calculated value obtained by the second calculation means 22 is V! Dotsuru. Similar to the current model, this calculated value V1d'' causes an error because the set value R of the secondary resistance deviates from the true value of the motor.

On the other hand, the first calculating means 21 calculates a component orthogonal to the primary current vector (δ-axis component) from the name of the primary voltage vector actually detected from the motor terminal. - Regarding the next voltage vector name, the component expressed in the α-β coordinate system ■tc
The component VITJVI of r+vtρ expressed in the γ-δ coordinate system
To convert to δ, the above 0η formula can be used. Here, if the α-axis is taken on the a-phase winding axis, the value of the detected primary voltage vector curve in the three-phase system (phase voltage v1..vlc)
can be converted into components of the α-β fixed coordinate system by . Therefore, the calculation means 21 calculates the detected primary voltage vector 2
Convert the value in the three-phase system to the component of the α-β fixed coordinate system using (2tJ2), then use the α7 formula to calculate the δ-axis component
Find lδ. In order to perform the coordinate transformation using Equation 09, it is necessary to know the angular position ε of the primary current vector. This is done by converting, for example, the value in the three-phase system of the primary current vector detected by the current transformers 41 and 42 in Fig. 1 into components of the α-β fixed coordinate system using a formula similar to formula a21. and then transform these components into polar coordinates. The value Vlδ calculated by the calculating means 21 in this manner is accurate because it is obtained from the actual primary voltage of the motor.

Therefore, the calculated value of the calculation means 21 is regarded as the true value Vlδ,
If the secondary resistance setting value R2 in the current model is corrected by the adjusting means 23 so that the calculated value ■IC of the calculating means 22 matches the true value Vlδ from the calculating means 21, the secondary resistance setting value becomes true. value, the current model will be able to output an accurate flux axis angular position. This corrective action is
As shown in FIG. 1, this is done by adding the correction value ΔR2 from the adjusting means 23 to the setting value R2° of the secondary resistance setting device 16, and the secondary resistance setting value used in the calculation is: Rz = Rz. +ΔR6.

By doing so, the current model can accurately calculate the flux axis angular position, thereby ensuring high performance variable speed control of the induction motor.

〔Example〕

4 and 5 show the main parts of the variable speed drive device for an induction motor according to the present invention, that is, the specific implementation of the first calculation means 21, the sand control means 23, and the second calculation means 22 in FIG. Give an example.

According to FIG. 4, in the calculating means 21, the voltage detector 211
The primary voltage vector 2 detected from the motor terminals as a three-phase line voltage is the phase voltage v1. .

After being corrected to Vlc, the three-phase/two-phase converter 212 converts the component V1 of the α-β fixed coordinate system according to equation
z, converted to V1β. Of course, it is also possible to directly convert the line voltage to the component v1(2, V1β).

The primary voltage vector 2 converted into the component V1(2, Vlβ) of the α-β fixed coordinate system is further coordinate-transformed in the vector rotator 213 according to the above 0η formula. As a result, −
The δ-axis component ■1δ of the next voltage vector name is output. In this case, the angular position ε of the primary current vector to be guided to the vector rotator 213 is the current transformer 41.4 in FIG.
However, here, an adder 214 that adds the angular position φ of the magnetic flux axis calculated by the current model and θ additionally obtained by the calculation means 22 described later is used. The output is given by (ε=φ+θ).

The calculated value v1δ of the first calculating means 21 is input to the adjusting means 23 together with the calculated value via'' from the second calculating means 22, which will be described later.
1, the recalculated values are compared. The control deviation from this comparison point 231 is changed to PI? A clause 2
33. The regulator 233 adjusts the control deviation to zero, that is, the calculated value vlC of the second calculating means 22 becomes the first value.
The first
In order to correct the secondary resistance setting value R2 used for calculation by the current model within the broken line frame in the figure, a correction value ΔRz is outputted to be added to the setting value R2° of the secondary resistance setting device 16.

FIG. 5 shows a specific embodiment of the second calculation means 22. According to this, in the calculation means 22, the K/P converter 221 calculates the current command value il4. as a component of the M-T coordinate system from the current model (FIG. 1). By converting iA into polar coordinates, the magnitude il and angle θ of the command current vector are output. The angle θ is determined by the adder 21 in the first calculation means 21.
4 (Figure 4).

Furthermore, the current command value i from the current model. , ia are divided by the above-mentioned i in dividers 2221 and 2222, respectively. The output of each divider is co
sθ and sinθ. On the other hand, the slip frequency ω1 calculated by the current model and the rotor angular velocity ω2 from the speed sensor 6 (FIG. 1) are added by an adder 2231 to obtain the magnetic flux angular velocity ω1. The magnetic flux command value A2 from the magnetic flux command unit 9 (FIG. 1) is differentiated by a differentiator 2241. Multiplier 2251
.about.2253, the adder 2232, and the coefficient unit 2271 correspond to the portion that calculates the value of the second term on the right side in the above-mentioned 0-en expression.

Also, another differentiator 2 that differentiates the angle θ from the coordinate converter
242, and the adder 2233 that adds the differential output of this and the output ω of the adder 2231 is a part that calculates the rotational angular velocity ωT of the primary current vector (γ axis) according to the above equation (2υ). Then, this ωT and converter 2
The multiplier 2243 that multiplies il from 21 and the coefficient 2272 that multiplies its output by the coefficient
This corresponds to the part that calculates the value of the first term on the right side of the equation.

Therefore, by adding the outputs of the coefficient multipliers 2271 and 2272 in the adder 2234, a desired calculated value v16* can be obtained. This calculated value is led to the adjustment means 23 in FIG. 4 mentioned above. Further, the magnetic flux axis rotational angular velocity ω obtained from the adder 2231 is used as a signal for controlling the polarity switch 232 in the adjustment means 23 in FIG. This polarity switching will be explained with reference to FIGS. 6 and 7.

To qualitatively explain the control principle, consider a steady state. dθ/dt=o (i.e., ωT=ω,), p('P2)
When =O, the αφ formula becomes. In this equation, the second term corresponds to the δ-axis component E and cos θ of the induced voltage vector π2. This relationship is shown in the vector diagram of FIG. As you can see from now on, ■! δ is a value unrelated to the voltage drop due to R.

, Now, we will explain the operation when R2 set in the current model of the control circuit deviates from the true value, and the magnetic flux axis (M axis) calculated by the current model and the magnetic flux axis (M' axis) in the motor deviate. do. In FIGS. 6 and 7, * means a command value within the control circuit. FIG. 6 shows the case where the direction of rotation of the magnetic flux is counterclockwise, and FIG. 7 shows the case where the direction of rotation of the magnetic flux is clockwise. Both figures show a state in which the M° axis is angularly advanced compared to the M axis.

First, FIG. 6 will be explained. Assuming that the current flowing through the motor is constant, the M° axis component of this current is 19, iH>i
, 4". This shows that the magnetic flux of the motor is larger, and the magnitude of the induced voltage vector of the motor is also larger (1π21>102'1). If the values of R, , L change Assuming that the voltage vector is not
curvature 1 > l V; "1, and the δ-axis component of the voltage vector name is V! δ> v 1,5".

When the magnetic flux axis shifts in this way, the change affects the δ-axis component of the voltage vector. In this case, in order to make the M axis and the M° axis coincide, the M axis may be rotated counterclockwise. This means increasing the absolute value of the slip angle, which can be achieved by increasing the secondary resistance setting in the control circuit.

In Figure 7, the magnetic flux vector is rotating clockwise, so
The polarity of the induced voltage generated is opposite to that shown in FIG. Focusing on the δ-axis component of the voltage vector, V! δ> v 1
B'', which is the same as in Figure 6, but in order to make the M-axis and M'-axis coincide, the M-axis must be rotated counterclockwise, and this is determined by the absolute value of the slip angle. This means decreasing the secondary resistance setting value in the control circuit.

As can be seen from the explanation of FIGS. 6 and 7, the polarity of the correction value ΔR1 of the secondary resistance differs depending on the rotation direction of the magnetic flux vector, that is, the polarity of the angular velocity ω deprivation of the magnetic flux. The polarity switch 232 in FIG. 6 takes this into consideration.

8 and 9 show different modifications of the calculation means 22 according to FIG. 5. In FIG. FIG. 8 shows the configuration when dθ/dt=0. This is advantageous because the configuration is simple when the torque change is small. The modification shown in FIG. 9 is obtained by further omitting the term p('Pz) in FIG. 8, and is advantageous when the change in magnetic flux is small because the configuration is simple.

Generally, the temperature rise of the secondary resistance follows the thermal time constant of the rotor, and the embodiment of FIG. 9 is sufficient to correct for the average change in the secondary resistance.

The various circuits in the embodiments described above can be easily constructed using any of known analog technology, digital technology, and software technology. In addition, in explaining the embodiments of this principle,
Although a transistor inverter is used, it goes without saying that other conversion devices 1 such as a current source inverter, a GTO inverter, a cycloconverter, etc. can also be used.

Furthermore, it is sufficient to modify the secondary resistance setting value directly (but not by modifying the slip frequency ω□, etc.), so there are various ways to do this. Deformation is possible.

〔effect〕

As described above, according to the present invention, the component orthogonal to the primary current vector of the sum vector of the induced voltage vector of the motor and the primary resistance voltage drop vector The primary resistance is determined by a first calculation means that does not use it as a calculation parameter, and the second calculation means uses the secondary resistance, which fluctuates significantly during operation, like the current model in the control circuit, as a calculation parameter. By correcting the secondary resistance setting value so that the value obtained by the second calculation means matches the value obtained by the first calculation means, the secondary resistance setting value is adjusted to the true value of the secondary resistance of the motor. Since the angular position of the magnetic flux axis calculated by the current model is accurate, the correction operation can be made effective even in a low speed range because it does not include the primary resistance as a calculation parameter like in the past. Therefore, the variable speed control performance of the electric motor can be significantly improved.

[Brief explanation of the drawing]

FIG. 1 is a block diagram showing an embodiment of the device of the present invention, FIG. 2 is a vector diagram for explaining the vector control principle, and FIG.
The figure is a block diagram showing a voltage model for magnetic flux calculation, FIG. 4 is a block diagram showing a specific example of the first calculation means and adjustment means in the embodiment of FIG. 1, and FIG.
6 and 7 are vector diagrams for explaining the role of the polarity switch in FIG. 4, and FIG. 8 and FIG. FIG. 9 shows different modifications of the specific embodiment of the second calculation means shown in FIG. 1--Induction motor, 2--Power converter, 5--Current regulator, 8--Speed controller, 9--Magnetic flux command, B--Current model for calculating the angular position of the magnetic flux axis 16--Secondary resistance setter, 19--Coordinate converter, 21--First calculating means, 22--Second calculating means, 23--Adjusting means. Ko 2 figure 3 concave 5 z 80

Claims (1)

[Claims] 1) Torque command value, magnetic flux command value, rotation speed, and motor constant for variable speed drive of an induction motor powered by a power converter that can control the magnitude, frequency, and phase of the output voltage. Using a current model that calculates the angular position of the magnetic flux axis from In a variable speed drive device for an induction motor, a first calculation means calculates a component of a primary voltage vector of the electric motor orthogonal to a primary current vector from a primary voltage vector detected from a terminal of the electric motor; and a primary voltage of the electric motor. a second calculation means for calculating a component of the vector orthogonal to the primary current vector from a torque command value, a magnetic flux command value, a rotational speed, and a motor constant in the same manner as the current model; The current model uses an intermediate calculation value obtained during the calculation process, such as a set value of a secondary resistance, which is one of the motor constants used for calculation, or a slip frequency derived from this, so that the calculation results of both calculation means match. A variable speed drive device for an induction motor, comprising: adjustment means for modifying; and a variable speed drive device for an induction motor. 2) The variable speed drive device for an induction motor according to claim 1, wherein the second calculation means shares at least a part of the calculation of the current model.