JPH1198898A - Method for controlling vector of ac motor - Google Patents

Abstract

PROBLEM TO BE SOLVED: To stably generate the torque with accuracy without receiving any influence from the iron loss of a stator and, in addition, to achieve a high energy transfer efficiency under the presence of the iron loss of the stator. SOLUTION: An equivalent load current iL of a stator from which the influence of the core loss of the stator is eliminated is generated by using a load current generator 9 constructed based on a relation iL=i1 +R1 .i1 -ν1 )/Rc established among the voltage ν1 , current i1 , a copper loss resistor R1 , and an iron loss resistor Rc of the stator. The current i1 of the stator is indirectly or directly controlled by utilizing the load current iL of the stator.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an induction motor, a synchronous motor, and a hybrid motor thereof, which can be used for a servo among AC motors that generate AC torque at a stator terminal and generate torque and mechanical energy. The present invention relates to a vector control method that can be commonly used for electric motors. In particular, the present invention relates to a vector control method capable of generating high-quality torque and taking high-efficiency energy conversion of electrical energy into mechanical energy in consideration of the influence of stator iron loss.

[0002]

2. Description of the Related Art When an AC motor such as an induction motor or a synchronous motor is used for a servo, control of a stator current is indispensable, and a vector control method has conventionally been known as a control method for this purpose. The vector control method includes a current control step of dividing and controlling a stator current into a d-axis component and a q-axis component on a rotating dq coordinate system including a d-axis and a q-axis orthogonal to each other. As the dq coordinate system at this time, a rotating coordinate system such as a rotor magnetic flux or a stator magnetic flux synchronized with a specific magnetic flux inside the electric motor with zero phase difference is adopted. As a matter of course, for this magnetic flux orientation, the vector control method has a step of internally determining the phase of the rotating coordinate system or taking in the phase information from the outside.

FIG. 11 is a block diagram schematically showing a typical example in which a conventional vector control method is implemented in a case where an induction motor is used as an AC motor. 1 is an induction motor, 2 is a rotor position detector, 3 is a power converter, 4 is a current detector, 5a and 5b.
Denote three-phase two-phase converters and two-phase three-phase converters 6a and 6b, respectively.
Denotes a vector rotator, 7 denotes a phase determiner, and 8 denotes a current controller.

In particular, the phase determiner 7 embodies the step of determining the phase of the dq coordinate system based on the estimation of the magnetic flux inside the motor, and the current controller 8 controls the fixed phase divided into d and q axis components. This embodies means for controlling a step of controlling each component of the daughter current. In FIG. 1, one vector signal composed of two scalar signals is represented by one thick signal line in order to avoid congestion of the signal line and ensure simplicity.

The three-phase stator current detected by the current detector 4 is converted to a two-phase current (ie, vector current) il on a fixed coordinate system by a three-phase to two-phase converter 5a, and then converted to a vector rotator 6a. Is converted to a vector current il in the dq coordinate system by
It is sent to the phase determiner 7 and the current controller 8. The current controller 8 generates a command voltage vl * on the dq coordinate system so that the converted stator current il follows the command current value il *, and sends it to the vector rotator 6b. At this time, the current controller obtains information on the power supply angular frequency ωl from the phase determiner, and generates the command voltage vl * in a form in which the d and q components of the current are also made non-interfering. 6b, the vector signal is converted into a vector command voltage vl * of a fixed coordinate system, and the two-phase / three-phase converter 5b
Send to In step 5b, the two-phase vector signal is converted into a three-phase command voltage and output as a command value to the power converter 3. The power converter 3 generates electric power according to the command, applies it to the induction motor 1, and drives it. When the rotor position information is input from the rotor position detector 2, the phase determiner 7 uses the rotor speed information obtained through the approximate differentiation processing of the position information and the like and the converted current il to determine the orientation. The position of the magnetic flux to be loaded is estimated, that is, the phase of the rotating dq coordinate system is determined as a vector signal, and the vector signal is output to the two vector rotators 6a and 6b. At this time, the power supply angular frequency ω1 is also generated.

[0006]

In the conventional vector control method, the parameters of the phase determiner 7 and the current controller 8 are traditional electric parameters of an induction motor, such as a stator copper loss resistance and a stator. It is determined based on inductance, mutual inductance, rotor copper loss resistance, and rotor inductance. In other words, in the conventional vector control method, the phase determiner and the current controller are designed and configured without considering the effect of the iron loss of the stator at all.

The electromagnetic steel sheet constituting the stator core of the induction motor generates iron loss represented by eddy current loss and hysteresis loss. The eddy current loss increases in proportion to the product of the square of the linkage flux and the square of the frequency, and the hysteresis loss increases in proportion to the product of the square of the linkage flux and the frequency. For this reason, when the phase determiner and the current controller are configured according to the conventional vector control method ignoring the effect of iron loss, the rotation speed of the motor is reduced even when the command current is applied to generate the same torque. The generated torque fluctuated dependently. In addition, the torque ripple became unnecessarily large. In motor control, high-efficiency control that minimizes the loss associated with the conversion of electrical energy to mechanical energy while generating the required torque is often required. With the conventional vector control method that is neglected, it has been difficult to accurately achieve and maintain the loss minimization. The conventional vector control method ignoring stator iron loss which cannot be ignored in practice actually has a problem in the accuracy and stability of torque generation and the efficiency of energy conversion. This problem is described, for example, in the literature (E. Levi: Impactofof).
Iron Loss on Behavior of
VectorControlled Inducti
on Machines, IEEE Trans. on
Industry Applications, Vo
l. 31, No. 6, pp. 1287-1296, 19
95).

As described above, an example of a block configuration in a case where an induction motor is controlled as an AC motor by a conventional vector control method ignoring the stator iron loss, and an example of a block configuration thereof will be described. Was explained. As is well known to those skilled in the art, the schematic block configuration in the case of controlling a synchronous motor as an AC motor by using a vector control method as a control target is basically the same as that of the induction motor. In the case of a synchronous motor,
Since the magnetic flux level of the rotor is known, and the position of the magnetic flux can be obtained directly from the position information of the rotor, the configuration of the phase determiner 7 that outputs vector-like phase information to the vector rotator is extremely simple. However, the schematic block configuration of the synchronous motor required in connection with the present invention is not essentially different from that of the induction motor. Also, as is well known to those skilled in the art, the stators of induction motors, synchronous motors, and their hybrid motors are basically the same as AC motors, and the conventional vector ignoring the stator iron loss described above. The problem of the control method is a problem common to AC motors including synchronous motors.

The present invention has been made in view of the above background, and an object of the present invention is to provide a stator iron loss capable of accurately and stably generating torque without being affected by stator iron loss. Of the present invention is to provide a vector control method for an AC motor that can achieve high energy transmission efficiency in the presence of the motor.

[0010]

In order to achieve the above object, the present invention is directed to a rotating dq coordinate system including a d-axis and a q-axis which are orthogonal to each other. A vector control method for an AC motor having a current control step of dividing and controlling a q-axis component and a stator voltage vl, a stator current il, a stator copper loss resistance Rl, and a stator iron loss resistance R
Relationship built from c

(Equation 1) Alternatively, the stator current il is controlled using a stator load current iL obtained by using at least a mathematically equivalent relation to this.

A second aspect of the present invention is the vector control method for an AC motor according to the first aspect, wherein the stator load current i
The method is characterized in that the stator current il is indirectly controlled by dividing L into a d-axis component and a q-axis component and directly controlling the same.

[0012] The invention of claim 3 is the first and second aspects of the present invention.
The vector control method for an AC motor according to the present invention, characterized in that at least one of the stator voltage and the stator current uses an estimated value instead of an actually measured value.

According to a fourth aspect of the present invention, there is provided the vector control method for an AC motor according to the second aspect, wherein the stator load current i
L is divided into a d-axis component and a q-axis component, and the d-axis component command value and the q-axis component command value of the stator load current for direct control are both command values for torque generation or torque equivalent thereto. It is characterized in that it is changed and generated according to a target value.

Next, the operation of the present invention will be described in detail. Physical quantities such as voltage, current and magnetic flux related to AC motors are
As is well known, it can be represented as a 2 × 1 vector signal. In describing a vector signal, a vector basis must first be determined. In the following description, the unit vector on the orthogonal axis rotating at the arbitrary instantaneous angular velocity ω shown in FIG.

If the stator magnetic flux linked to the stator is expressed as φ1, the relationship between the physical quantities as viewed from the stator side of the AC motor is the equivalent circuit using vector signals shown in FIG. (Abbreviated as a circuit). In the equivalent circuit diagram, reference numeral 13 denotes a stator loss circuit expressing a loss on the stator side, and reference numeral 14 denotes a stator load circuit expressing a lossless load viewed from the stator side. The relation between this vector equivalent circuit composed of the stator loss circuit and the stator load circuit can be described by the following equations (2) to (4) if it is strictly expressed using mathematical expressions.

(Equation 2)

(Equation 3)

(Equation 4) 2 and iR in equations (3) and (4) are iron loss resistance R
c is an equivalent current flowing through c, and vL is a load voltage applied to the load. Ω is the instantaneous angular velocity of the vector base, s is a differential operator, I is a unit matrix, and J is an alternation matrix defined by the following equation.

(Equation 5)

The stator loss can be evaluated as the sum of the loss due to the copper loss resistance Rl and the loss due to the iron loss resistance Rc as shown in the following equation (6).

(Equation 6)

The loss shown in the second term on the right side of the above equation is a loss directly related to the present invention, and does not represent iron loss consisting of eddy current loss and hysteresis loss generated in the magnetic steel sheet core. Not be. First, this fact will be rigorously clarified using mathematical formulas. Therefore, the iron loss resistance Rc is expressed in a more detailed form having a frequency characteristic as described below.

(Equation 7) Here, ω1 is the angular frequency of the stator power. The loss shown in the second term on the right side of the equation (6) can be evaluated as follows by using the equations (3) and (7).

(Equation 8)

The first term on the right side of the equation (8) indicates the stator eddy current loss, and the second term indicates the stator hysteresis loss.
In other words, regarding the eddy current loss and the hysteresis loss, which are the main elements of the iron loss of the magnetic steel sheet core, the eddy current loss is 2% of the linkage flux.
It has long been known that the hysteresis loss increases in proportion to the product of the square of the power and the square of the frequency, and that the hysteresis loss increases in proportion to the product of the square of the interlinkage magnetic flux and the frequency. It is clear that the lost loss appropriately expresses this iron loss characteristic of the magnetic steel sheet. In other words, FIG. 2 used in the description of the present invention relating to the vector control method of the AC motor, and equations (2) to (4) mathematically expressing the same appropriately represent stator iron loss. It is clear that

In an AC motor, a stator and a rotor are electromagnetically coupled, and electric energy applied to the stator is transmitted to the rotor via the electromagnetic coupling and is emitted from the rotor as mechanical energy. Have been. The magnetic flux is responsible for this electromagnetic coupling, and in FIG. 2, the stator magnetic flux φ1 in the stator load circuit expresses this electromagnetic coupling. The stator magnetic flux, which is responsible for the electromagnetic coupling, is, of course,
It is composed of a stator chain intersection of a magnetic flux generated by the stator inductance and a magnetic flux generated by the rotor.

In order to accurately and stably control the torque generated by the rotor or to efficiently control the energy emitted by the rotor, as shown in FIG. The source must control the stator current il after accurately grasping the load current iL flowing into the stator load circuit. According to the invention of claim 1, this becomes practically possible because this load current iL, which is a direct source of torque and mechanical energy in the rotor, becomes available. The signals used in the relationship represented by the equation (1) according to the present invention are only the stator voltage and the stator current that can be directly measured from the stator terminal of the AC motor, and the practicability is clear. In order for the obtained or grasped load current to show an appropriate value, the relationship shown in the equation (1) constructed in the present invention or a relationship mathematically equivalent thereto expresses the stator iron loss appropriately (2). ) ~
The system of equations (4) must be satisfied. The relationship according to the present invention naturally satisfies this, which is (1)
It is easily confirmed by substituting the relation of the equations into the simultaneous equations of equations (2) to (4). As is apparent from the above description, according to the first aspect of the present invention, an effect is obtained that the stator current of the AC motor can be controlled in an appropriate form in consideration of the effect of the stator iron loss. In other words, for an AC motor with non-negligible stator iron loss,
The effect is obtained that the stator current control necessary for achieving high-precision and stable torque generation and high-efficiency energy generation becomes possible.

According to the second aspect of the present invention, in the vector control method according to the first aspect, the stator load current which is a direct source of the rotor torque and the mechanical energy generation is set to d.
The control can be directly performed by dividing the axis component and the q-axis component. As a result, an effect is obtained that the torque and energy generated by the rotor can be directly controlled.

According to the third aspect of the present invention, in the vector control method according to the first and second aspects, an estimated value is used instead of an actually measured value for at least one of the stator voltage and the stator current. Will be able to As a result, for example, when the estimated value is used for the stator current, an effect is obtained that it is less susceptible to noise or harmonic components of the signal included in the measured current value. When the estimated value is used for the stator voltage, an operation similar to that of the stator current is obtained. In addition to this, there is an effect that the voltage detector can be removed.
Also, as will be described later in detail using an embodiment example,
An effect that various uses of the present invention can be obtained, such as taking in an observer or the like useful for improving performance.

According to a fourth aspect of the present invention, in the vector control method for an AC motor according to the second aspect, the stator load current iL is divided into a d-axis component and a q-axis component for direct control. Both the d-axis component command value and the q-axis component command value of the stator load current can be changed and generated according to a torque generation command value or a torque target value equivalent thereto (hereinafter, for simplicity, A command value for torque generation or a target torque value equivalent thereto is called a command torque value or the like). Due to the simultaneousness and freedom in the generation of the d-axis command value and the q-axis command value, various rules that can achieve high efficiency of energy conversion with different evaluation criteria while achieving required torque generation can be obtained. An effect is obtained that the d-axis command value and the q-axis command value can be generated in a mutually associated manner.

The above operation can be theoretically explained as follows. In the AC motor to which the present invention of claims 1 and 2 is applied, the torque τ is generated according to the following equation.

(Equation 9) Np in the above equation means the number of pole pairs of the AC motor. As is clear from equation (9), the torque is calculated by the inner product of two vector signals iL and Jφl. This means that there are countless combinations of load current components that generate the same torque. According to the present invention, one of the innumerable combinations having improved energy conversion efficiency such as the minimum loss or the maximum power factor can be selected and used as a command value for load current control. become.

[0025]

Embodiments of the present invention will be described below in detail with reference to the drawings. FIG. 3 shows a basic structure of an embodiment of a vector control device and an AC motor to which the vector control method of the present invention is applied. 1 such as an electric motor in FIG.
1 to 6 to which the conventional vector control method is applied.
This is the same as 1 and the devices 7 and 8 are the same as the conventional device. In the present embodiment, a major difference from the related art is the presence of the load current generator 9, and the devices 7 and 8 have only small differences in details. Therefore, in the description of the embodiment, the load current generator 9 will be mainly described in detail, and only the differences between the devices 7 and 8 will be described.

The three-phase stator current detected by the current detector 4 is converted into a vector current of a rotating dq coordinate system via a three-phase two-phase converter 5a and a vector rotator 6a, and then is converted to a load current generator 9. Is entered. At the same time, the load current generator 9 receives as input the command voltage vl * of the stator on the same coordinate system and the power supply angular frequency ωl from the phase determiner and receives the load current i
L is output to the phase determiner 7 and the current controller 8.
The phase determiner 7 determines the magnetic flux position as a vector signal while using the rotor position information from the rotor position detector, and outputs this to the vector rotator. When an induction motor is used as the AC motor, current information is also required to determine the magnetic flux position, as will be apparent to those skilled in the art. The current controller 8 also receives a command value iL * of the load current, and generates a command voltage vl * on the rotating dq coordinate system so that the load current iL directly follows the command value iL *. To the vector rotator 6b and the load current generator 9. In the present embodiment, as apparent from the fact that the command voltage vl *, which is the source of the stator current il generation, is determined by directly controlling the stator load current iL, the stator load current iL is determined. , The stator current il is indirectly controlled through direct control. The command value is used as an estimated value of the stator voltage.

FIG. 4 shows the internal structure of the load current generator 9. This load current generator 9 directly realizes the relationship of the formula (1) according to the present invention. However, at the time of realization, the command value is applied as a signal of the stator voltage, and further, depending on the power supply angular frequency in consideration of the frequency-dependent characteristics of a part of the iron loss resistance as shown in the equation (7). Thus, the resistance value corresponding to the hysteresis loss is automatically adjusted. A penetrating arrow in the figure indicates this automatic adjustment.

Next, changes of the phase determiner 7 and the current controller 8 in the conventional vector control method according to the present embodiment will be described. The phase determiner 7 and the current controller 8 of the present embodiment are configured such that the stator current il and the stator voltage and the stator copper loss resistance of the conventional phase determiner and the current controller are converted into a coefficient by a coefficient. (Rc / (Rl +
Rc)). In other words, if the phase determiner and the current controller are designed and realized by replacing the stator current, the stator voltage, and the stator copper loss resistance in the conventional vector control method according to the relationship shown in the following equation (10), Good.

(Equation 10)

The relevance of this permutation, which facilitates the design and implementation of the phase determiner and current controller in the present invention, is illustrated in FIG.
It can be easily understood by those skilled in the art if it is considered that the relationship of the following equation (11) is established in common with the AC motor in the vector equivalent circuit and the equations (2) to (4).

[Equation 11]

By using the output signal of the phase determiner 7 for the vector rotators 6a and 6b, the required magnetic flux orientation can be accurately achieved. Also,
By using the above-described current controller 8, it is possible to generate a stator voltage command value vl * for generating a load current iL that can follow the load current command value iL *.

Next, a second embodiment will be described. FIG. 5 is a block diagram showing the second embodiment.
The difference from the first embodiment is that the arrangement position of the load current generator 9 is changed. That is, in this embodiment, immediately after the three-phase stator current detected by the current detector 4 is converted into the two-phase vector current on the fixed coordinate system by the three-phase to two-phase converter 5a, the load current is generated. The device 9 is arranged to obtain the load current, which is directly controlled. Of course,
Corresponding to this, a stator voltage on a fixed coordinate system is used. Also in the present embodiment, as shown in the figure, the command value is used as the estimated value of the stator voltage. The block configuration of the load current generator 9 in this case is as follows.
The same thing as FIG. 4 may be used. Unlike the first embodiment, the input / output vector signal of the load current generator is a signal on a fixed coordinate system, which is basically an AC signal. The same phase determiner 7 and current controller 8 as in the first embodiment can be used as the phase determiner 7 and the current controller 8 according to the first embodiment, whereby the required magnetic flux orientation and the load current to the load current command value can be obtained. Current control to ensure the following is achieved.

FIG. 6 shows a third embodiment in which an induction motor is used as an AC motor. The major difference between this embodiment and the second embodiment is that
The point is that not only generation of a load current but also determination of a phase for magnetic flux orientation is performed directly on a fixed coordinate system. Therefore, the observer 10 capable of performing the simultaneous execution is configured. The observer 10 uses the current, voltage, and rotor position information obtained by the current detector 4, the voltage detector 11, and the rotor position detector 2 to determine a load current and a phase for magnetic flux orientation. Is output to the vector rotator 6a and the vector rotators 6a and 6b. As the voltage detector 11 in the present embodiment, a voltage detector that detects two line voltages of three phases, converts the voltage to a two-phase vector voltage vl on a fixed coordinate system, and outputs the vector voltage vl is used. The load controller converted into a signal on the rotation dq coordinate system by the vector rotator 6a and the same command value are input to the current controller 8. As the current controller 8,
For example, the same components as those in the first and second embodiments described above may be used. However, in this embodiment, the configuration is simplified and a component that does not use the power supply angular frequency ω1 is used.
As a result, required magnetic flux orientation and current control for ensuring that the load current iL follows the load current command value iL * are achieved.

In the observer according to this embodiment, the relation of the equation (1) constructed by the present invention is changed to the relation of the following equation (12) which is mathematically equivalent, and the estimated value as the stator current is used. I use it.

(Equation 12) FIG. 7 is a block diagram schematically showing the internal structure of the observer 10 using the relationship of the expression (12). The broken line block 10a in the figure is a part utilizing the present invention. That is, the observer uses the estimated value as the stator current in the relationship of the equation (12), uses this to obtain the load current iL, and outputs the load current iL from the observer. This observer can be paraphrased to use the relationship of equation (12) temporarily for estimating the stator current and use this to finally obtain the load current. The observer of the present embodiment is a same-dimensional observer when the load current and the normalized rotor magnetic flux φ2n are used as state variables. In FIG. 7, L1t indicates the total stator leakage inductance, W2 indicates the reciprocal of the rotor time constant, and R2n indicates the normalized copper loss resistance of the rotor. From this it will be clear to a person skilled in the art that 10b
Is an integrator, 10c is a 4x4 system matrix for state variables, 10d is an approximate differentiator for obtaining rotor speed information, 10e is an observer gain, and 10f is phase information of a magnetic flux vector. Means a unitizer for performing In FIG. 10, automatic correction of the stator iron loss resistance value in consideration of the frequency dependence of the stator iron loss is performed as follows.
Not shown to avoid congestion in the figure. As above, one configuration example of the observer using the present invention has been described.

The present invention according to claims 1, 2 and 3 has been described using the first and second examples as an embodiment common to AC motors including induction motors and synchronous motors. In addition, the third embodiment has been described as an embodiment targeting only the induction motor. It will be clear to a person skilled in the art that the invention is not limited to the embodiments described above.

For example, when the AC motor to be controlled is an induction motor, the present invention can be applied to a case where the magnetic flux inside the motor to be orientated is selected from among a stator, a rotor, and a gap. It is. In addition, as is apparent from the above-described embodiment, the method of orienting the magnetic flux, that is, the method of determining the phase of the rotating dq coordinate system is either direct, indirect, or a hybrid of these. However, the present invention is applicable. As described with reference to the embodiment, the present invention can be applied to any of a fixed coordinate system and a rotating coordinate system. Further, application in other devices such as an observer is also possible. Although not shown as an example of an embodiment, even for a sensorless induction motor having no rotor position detector or rotor speed detector, regardless of the combined orientation method, rotor speed estimation method, etc. The present invention is applicable.

When the AC motor to be controlled is a synchronous motor, the magnetic flux inside the motor to be orientated is
The present invention is applicable to any of the stator and the rotor. As described in the embodiment, the present invention can be applied on any of a fixed coordinate system and a rotating coordinate system. Further, application in other devices such as an observer is also possible. The detailed configuration of the observer of the synchronous motor is different from the detailed configuration of the observer of the induction motor because the structure of the rotor is different from that of the induction motor.
It can be applied without difficulty if the equivalent relation of the equation (2) is changed. Although not shown as an example of an embodiment, the present invention is applicable to a sensorless synchronous motor without a rotor position detector regardless of the rotor position estimation method.

In the description of the present invention, in order to ensure the simplicity of the description, the term "rotary AC motor" has been used consistently. However, the present invention is not limited to the rotary AC motor, and the present invention is not limited to the linear AC motor. It is also applicable to AC motors of the form. As is well known to those skilled in the art, a linear AC motor can, in principle, be treated as a straight open version of a rotary AC motor. The present invention can be applied to a linear AC motor by paying attention to which of the stator and the rotor of the rotary motor used in the description of the present invention corresponds to the movable portion of the linear motor. This correspondence depends on the structure of the target linear AC motor and is determined structurally as those skilled in the art are familiar with.

As is clear from the first to third embodiments,
The load current iL, which is a direct source of torque and mechanical energy in the rotor, can be controlled to follow the load current command value iL * according to the present invention. Next, a method for generating the load current command value iL * will be described based on the present invention.

In the present invention, it is assumed that the torque of the AC motor is generated by the relationship between the load current iL and the magnetic flux as described above with reference to the equation (9). On the other hand, in the conventional vector control method ignoring the existence of the stator iron loss, it is assumed that the torque is generated due to the relationship between the stator current il and the magnetic flux. As can be easily understood by paying attention to this difference, as the change generation of the load current command value corresponding to the command torque value or the like, the command value of the d-axis component of the load current iL related to the magnetic flux is held at a constant value. Then, the q-axis component of the load current iL may be automatically changed and corrected in proportion to the command torque value or the like. The invention according to claims 1 to 3 makes it possible in this case to generate torque with high accuracy and high stability.
In addition, to achieve high energy conversion efficiency,
As described in connection with the operation of the present invention, the command values for both the d-axis and q-axis components of the load current may be changed and generated according to the command torque value and the like.

FIG. 8 is a block diagram relating to one embodiment of the present invention. The command converter 12 shown in FIG. 3 controls the respective command values iLd * for controlling the dq-axis components of the load current according to the command torque value τ *, etc.
An apparatus for updating and generating both iLq * is shown. 12a
Denotes an intermediate command generator, 12b denotes a d-axis command generator, and 12c denotes a q-axis command generator. The intermediate command generator 12a performs common and complicated arithmetic processing in the process of generating two command values of the load current from the command torque value etc. τ *, and outputs the processed signal as the intermediate command value λ. doing. The d-axis and q-axis command generators perform simple arithmetic processing on the intermediate command value λ, and obtain the desired command values iLd *, i
Lq * is generated.

The optimum value of the load current value for minimizing the loss caused by the copper loss resistance and the iron loss resistance while generating the required torque is, for example, an optimum control problem for minimizing the loss by restricting the required torque value. And can be determined by this solution. Similarly, for the optimum value of the load current that maximizes the power factor of the stator side power while generating the required torque, for example, an optimal control problem that maximizes the power factor with the required torque value as a constraint condition is constructed. It can be determined by this solution. Generation of such an optimal command value for improving the conversion efficiency of electrical energy to mechanical energy generally requires a solution of a higher-order equation. According to the configuration shown in the present embodiment, two command value generations are required. Since the amount of real-time computation required for the above can be substantially reduced by half, the practical value is great.

When solving a higher-order equation approximately,
The calculation required to generate the command value from the command torque value or the like can be considerably simplified, and the necessity of generating an intermediate command value common to the load current d-axis command value and the q-axis command value is relatively reduced. Therefore, when the approximation processing is employed, the d-axis command value and the q-axis command value may be directly generated from the command torque value or the like. FIG. 9 is a block diagram showing an example of the internal structure of the command converter 12 in such a state. 12 in FIG.
b indicates a d-axis command generator, and 12c indicates a q-axis command generator. However, unlike the command converter of FIG. 8, the d-axis command value and the q-axis command value are directly generated from the command torque value and the like.

It should be pointed out that a processing structure other than that shown in FIGS. 8 and 9 can be employed as the command converter 12. For example, a structure may be adopted in which the d-axis command value itself is adopted as an intermediate command value, a d-axis command value is first generated from a command torque value or the like, and a q-axis command value is generated from the generated d-axis command value. Alternatively, this dual structure may be used. As in the case where the d-axis command value is held constant,
If one of the d and q components of the load current command value for the required torque is determined, the other can be easily determined.

The embodiment of the present invention has been described above, focusing on the structure of the command converter when the d-axis and q-axis command values of the load current are obtained through a rational real-time solution of the equations according to the command torque value and the like. . It should be pointed out that this command converter structure can also be used in the case of using a function table instead of the real-time solution of the equation. For example, a unique relationship between the command torque value and the like and the d and q axis command values is obtained in advance, and a command value function table is prepared in which each is a reference input and a reference output. The same structure as that shown in FIG. 9 can be adopted as the structure for reference. If the relation of generation of the intermediate command value λ from the command torque value τ * is tabulated, the same structure as in FIG. 8 can be adopted. These two
As will be understood from the examples, the same applies to other structures. In particular, if a function table is used for complicated calculations, the calculation load required for real-time processing can be significantly reduced.

In the implementation of the command value function table, the resolution of the reference input becomes a practical problem, and improving the resolution generally leads to an increase in the capacity of the storage element. In order to suppress the increase in the capacity of the storage element while substantially improving the resolution, the input / output relations other than the input / output relations stored in the command value function table should be generated by interpolating from the stored input / output relations. It should be pointed out that it is good. FIG. 10 shows an embodiment in which an interpolation function is used together. Here, an outline of interpolation when a linear function is used as the interpolation function is schematically shown. The horizontal axis in the figure indicates the reference input, and the vertical axis indicates the reference output. The breakpoint portion is the input / output relationship stored in the command value function table, and the straight line portion is the interpolated input / output relationship.

It should be pointed out that the interpolation function used for interpolation is not limited to a linear function as in the above embodiment. In general, any function that can be easily calculated and has a good approximation can be used as an interpolation function. It is also pointed out that different interpolation functions may be used depending on the reference section.

In the above embodiment shown in FIG. 10, the number of reference input / output data and the number of interpolation functions stored in the command value function table are substantially the same, and the interpolation range of the interpolation function is It is a partial supplement to the data before and after the saved reference input / output data. On the contrary, it should be pointed out that the entire dynamic range covered by the command value function table can be totally interpolated by one to several approximation functions. Various methods can be considered as a method of determining the overall interpolation function. For example, simply, the reference input / output may be approximated by a polynomial, and the coefficient at this time may be determined using the least square method or the like. In this case, the entire dynamic range is interpolated by one to several approximation functions. In this case, the command value function table stored in the storage element disappears. Is realized by an approximation function.

The command torque value or the like according to the present invention is not limited to the torque command value at the time of torque control, but may be a torque target value according to a torque command value such as an output signal of a speed controller at the time of speed control. I should point out that

[0049]

As is clear from the above description, the present invention has the following effects. In particular, according to the first aspect of the present invention, an effect is obtained that the stator current of the AC motor can be controlled in an appropriate form in consideration of the effect of the stator iron loss. As a result of this action, for an AC motor having a stator iron loss that cannot be ignored, it is possible to obtain an effect that high-precision and stable torque and energy can be generated in the rotor through control of the stator current. Can be

In particular, according to the second aspect of the present invention, an effect is obtained that the torque and energy generated by the rotor can be directly controlled, and as a result of this operation, the torque and the energy of the rotor can be increased. An effect is obtained that accurate and stable control can be easily performed.

In particular, according to the third aspect of the present invention, it is possible to obtain an effect that noise and signal harmonic components contained in the stator voltage or the stator current are not affected. As a result of this operation, the control such as the torque generation may be based on the fundamental wave component of the signal, and the effect of further improving the control accuracy of the torque generation or the like is obtained. In addition, an effect that the voltage detector can be removed can also be obtained, so that an effect that the cost for realizing the present invention can be reduced can be obtained. In addition, since an effect that various uses of the present invention can be obtained is obtained, it is also possible to obtain an effect that control performance can be further improved through taking into an observer or the like.

In particular, according to the present invention, an effect is obtained that the d-axis component command value and the q-axis component command value of the load current can be simultaneously and freely generated according to the command torque value and the like. . As a result of this operation, the effect that the efficiency of conversion of electrical energy into mechanical energy can be enhanced while ensuring high accuracy and stable generation of required torque and energy achieved by the present invention of claims 1 to 3 is obtained. can get.

[0053]

[Brief description of the drawings]

FIG. 1 shows a vector basis for a vector signal according to the invention.

FIG. 2 is a vector equivalent circuit having a stator core loss.

FIG. 3 is a block diagram showing a basic configuration of a vector control device according to the embodiment;

FIG. 4 is a block diagram showing a schematic configuration of a load current generator according to the embodiment;

FIG. 5 is a block diagram illustrating a schematic configuration of a vector control device according to the embodiment;

FIG. 6 is a block diagram illustrating a schematic configuration of a vector control device according to the embodiment;

FIG. 7 is a block diagram showing a schematic configuration of an observer according to the embodiment;

FIG. 8 is a block diagram showing a schematic configuration of a command converter according to the embodiment;

FIG. 9 is a block diagram showing a schematic configuration of a command converter according to the embodiment;

FIG. 10 is a diagram showing a schematic relationship between data stored in a command value function table and an interpolation function in the embodiment;

FIG. 11 is a block diagram showing a schematic configuration of a conventional vector control device.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 AC motor (induction motor, synchronous motor) 2 Position detector 3 Power converter 4 Current detector 5a Three-phase two-phase converter 5b Two-phase three-phase converter 6a Vector rotator 6b Vector rotator 7 Phase determiner 8 Current Controller 9 Load current generator 10 Observer 11 Voltage detector 12 Command converter 13 Stator loss circuit 14 Stator load circuit

Claims (4)

[Claims] 1. A vector control of an AC motor having a current control step of dividing and controlling a stator current into a d-axis component and a q-axis component on a rotating dq coordinate system composed of d-axis and q-axis orthogonal to each other. A stator voltage vl, a stator current il and a stator copper loss resistance Rl.
IL = il + (Rl.il-vl) / Rc or a stator load current iL obtained using at least a mathematically equivalent relationship with the stator load current iL. A vector control method for an AC motor, comprising controlling a sub-current il. 2. The stator current il is indirectly controlled by dividing said stator load current iL into a d-axis component and a q-axis component and directly controlling said stator current il. Vector control method of AC motor. 3. An AC motor according to claim 1, wherein at least one of said stator voltage and said stator current uses an estimated value instead of an actually measured value. Vector control method. 4. A d-axis component command value and a q-axis component command value of the stator load current for dividing and directly controlling the stator load current iL into a d-axis component and a q-axis component, According to the torque generation command value or the torque target value equivalent to this,
3. The vector control method for an AC motor according to claim 2, wherein the change is generated.