JPS5948592B2 - Flyback EHT and sawtooth current generator - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention is a circuitry having a switching device that periodically becomes non-conductive during a flyback period τ and conductive during a scan period T-τ, and an input terminal connected to the switching device, comprising: a circuitry comprising at least one primary winding and a transformer having a secondary winding, one or more coils connected thereto through which a sawtooth current flows during the scanning period; A flyback EHT and sawtooth current generator having a rectifier connected to obtain a flyback EHT from voltage pulses generated in the secondary winding during a flyback period, the circuit network comprising a transformer and a sawtooth wave current generator. During the flyback period, due to the leakage inductance present between the windings of a first resonant frequency fα that is approximately equal to the value when K of the relative decrease in the slope of this current at the end of the scanning period with respect to the slope at the end of the scan period, and when K in this equation is an odd integer greater than 1 The present invention relates to a flyback EHT and sawtooth current generator, particularly for use in television receivers, having a second resonant frequency f approximately equal to the value of y.

From Dutch Patent No. 88020 and "Television I" Bi-1, written by Kerkhoff and Davrie Werner, 3rd edition, Chapter With a transformer having a primary winding and a secondary winding, the impedance of existing circuit elements such as the windings of this transformer, the leakage inductance between the primary and secondary windings, generally connected to the primary winding It is known to design the inductance and parasitic and non-parasitic capacitance of the coil to appropriate values so that little or no free oscillations occur in the secondary voltage during the scan period.

Such free oscillations result in a loss of useful energy, which is mainly dissipated in the transformer, resulting in overheating of the transformer, and also have the disadvantage that, if the free oscillations are large, the switching device, which should be conducting during the scanning period, becomes non-conducting quickly. has. The above document states that the circuit constitutes a quartic network having two resonant frequencies during the flyback period and that the free oscillation causes the circuit elements to move between these two resonant frequencies Fct.
It has been shown that Fy and Fy can be kept small by designing them to have defined values depending on the duration of the flyback period .tau. and the duration of the scanning period T-.tau..

As is clear from the above formula, the optimal values of fα and Fy are:
It also depends slightly on the extent to which the slope of the sawtooth current flowing through the deflection coil changes when using a so-called S-shaped correction. In theory, a vibration-free scanning period can be obtained by accurately designing fα and f in the flyback EHT and sawtooth current generator.

However, this cannot be achieved in practice due to unavoidable interference effects.

These disturbing effects are as follows. (1) Parasitic reactance exists within the circuit network. This reactance is irresistibly small, but large enough to make vibration-free scanning impossible. (2) There is loss in the network during the flyback period.

Most of it is useful energy that must be extracted through the rectifier. (3) Variations in circuit element tolerances and product characteristics.

These are frequencies fα and Fy. cannot be made precisely equal to the value required for vibration-free scanning. SUMMARY OF THE INVENTION The object of the present invention is to provide a design method in which the generated scanning vibrations can be kept extremely small despite the above-mentioned disturbing influences.
For this purpose, the present invention utilizes the above-mentioned flyback EHT.
In a dual sawtooth current generator, the network has a frequency F
a and Fy at which the impedance at the input terminal of the network is minimum, and the frequency is approximately equal to an integer multiple of the reciprocal of the duration of the scanning period. shall be. The network has a predetermined resonant frequency fβ between the two parallel resonant frequencies fα and Fy at which the impedance at the input terminal of the network is minimum (zero in the case of a lossless network).

This is in fact the resonant frequency of the generator during the scanning period and thus when the switching device is conducting. Flyback E
It is known that only the parallel resonant frequencies Fct and Fy are important and the series resonant frequency fβ is irrelevant for accurately tuning the vibration-free scan of the HT generator.
The value of fβ only affects the waveform of the flyback pulses generated across the input terminals of the network. The invention provides that the value of the frequency fβ has a very large influence on the amplitude of the scanning oscillations remaining as a result of said disturbance influence, and that this amplitude makes the value of fβ approximately equal to an integer multiple of the reciprocal of the duration of the scanning period. This was done based on that knowledge.

This type of generator is preferably designed such that the Q of the generator during the scanning period is greater than or equal to 25 at the frequency fβ.

The applicant's Japanese Patent Application No. 47-39236 describes a flyback EHT and sawtooth current generator in which the flyback network has at least three resonant frequencies fα, F, and Fy or higher. ing.

Almost vibration-free scanning can be achieved by making each of these resonant frequencies at least approximately equal to (where K is 1, 3, 5 or 1 5, 7).
It is a positive integer such that

In such a case, the network has a first series resonant frequency fβ1 between frequencies fα and F, at which the input impedance of the network is minimized, and a first series resonant frequency fβ1 between frequencies fε and F, at which the input impedance of the network is at a minimum. It has two series resonance frequencies FI2. fβ1 and fβ2 are the frequencies at which the network resonates during the scan period. Even in flyback EHT generators of this type, the scanning is not completely vibration-free due to the disturbance effects mentioned above.

The vibrations generated in this case generally consist of two components: a component with frequency fβ, and a component with frequency f/32.
Independently of each other, the frequency fβ1 needs to be optimal, i.e. approximately equal to an integer multiple of the reciprocal of the duration of the scanning period, to bring the fβ1 component to a minimum amplitude, and to bring the fβ2 component to a minimum amplitude, It has been found that the frequency fβ2 needs to be approximately equal to an integer multiple of the reciprocal of the duration of the scanning period.

However, on the other hand, it has been found that the lowest frequency components generally have significantly larger amplitudes than the higher frequency components.

For a flyback EHT and sawtooth current generator, the network has a third resonant frequency fε approximately equal to the above equation with K being an odd integer and y during the flyback period Fct and f. In this case, a significant reduction in scanning oscillations can be achieved in advance if the frequency fβ1 at which the impedance of the input terminal of the network between fα and F is minimized is approximately equal to an integer multiple of the reciprocal of the duration of the scanning period.

As mentioned above, an optimal design is achieved when the frequency fβ between F6 and F, at which the impedance at the input terminal of the network is minimum, is also approximately equal to an integer multiple of the reciprocal of the duration of the scanning period. The invention will be explained with reference to the drawings. The example of FIG. 1 shows a transformer 1 having a primary winding 2, one or more auxiliary windings 3 and a secondary winding 4 coupled to the primary winding.

The lead tap 6 of the primary winding 2 is connected to the positive terminal of the voltage supply source 7, and its negative terminal is grounded. multiple deflection coils 9
, a series circuit of a linearity corrector 10 and an S-shaped correction capacitor 11 is arranged between the 20th output tap 8 of the primary winding and the lower terminal. The lead tap 8 and the lower terminal of the primary winding are positioned symmetrically with respect to the lead tap 6 to energize this series circuit symmetrically with respect to ground. A transistor 12 acting as a switch is provided between the upper terminal of the primary winding and ground, and a capacitor 13 is connected in parallel with this transistor. One end of the secondary winding 4 is grounded, and the other end is connected to a rectifier circuit comprising a rectifier 14 and a smoothing capacitor 15. The flyback EHT obtained from the rectifier is connected to an accelerating anode (not shown) of the television picture tube. A switching pulse is applied between the base and emitter of transistor 12 via isolation transformer 18, series inductance 19 and parallel diode 20, which periodically shuts off transistor 12 at the end of each scan period.

Transistor 12 is a so-called slow switching transistor, and elements 19 and 20 are provided to accelerate the switching off of this transistor at the end of the scanning period. FIG. 2 is a simplified equivalent circuit diagram of the circuit of FIG.

In this equivalent circuit diagram, E indicates a voltage source 7, and SW indicates a switch consisting of a transistor 12 and a diode 20. C, is the collector-emitter capacitance of the transistor 12, the primary winding 2, the auxiliary winding 3, and the deflection coil 9.
and the capacitance of the capacitor 13 including the parasitic capacitance of the linearity corrector 10. L1 represents the inductance of the primary winding 2, the deflection coil 9 and the linearity corrector 10 connected thereto, converted into the terminal to which the switch is connected. L2 is the leakage inductance between the secondary winding 4 and the primary winding 2, and C2 is the parasitic capacitance of the secondary winding and the input capacitance of the rectifier circuit (these are also converted to the terminals to which the switch is connected). 5. During the scan period the switch SW is closed. This causes a voltage E from the voltage supply 7 to be developed across the capacitor C1 and also across the inductor L1. As a result, the voltage E from the voltage supply 7 is developed across the capacitor C1 and also across the inductor L1. (sawtooth wave) current flows through the inductor L1. The switch SW is made non-conducting by the pulse applied to the base electrode of the transistor 12, and due to the magnetic energy present in the inductor L1, the switch SW is made non-conducting. Free oscillations occur at .These oscillations generate pulse-like voltages V1 and 2, or flyback pulses, across capacitors C and C2. At the same time as the collector potential of transistor 12 becomes negative with respect to ground, the collector base of transistor 12 P
The n-junction becomes forward biased and the next scan period begins. Therefore, the switch SW in the equivalent circuit diagram in Figure 2
automatically closes as soon as the flyback voltage present across the switch becomes zero. The sawtooth current in the circuit of FIG. 1 flows through the diode 20 during the first part of the scan period.
, base-collector junction of transistor 12, transformer 1
The energy flows through the deflection coil 9 to the voltage supply source T, and returns the energy to the voltage supply source T.

Some time after the start of the scan period, the base-emitter junction of transistor 12 is made conductive by a pulse applied to the base electrode of transistor 12, so that during the second part of the scan period, a sawtooth current of opposite polarity is applied to the voltage. It flows from the source T through the transformer 1 and the deflection coil 9 and then through the collector and emitter electrodes of the transistor to ground, the voltage supply T supplying energy to the network. During the scan period, the sawtooth current must flow only through the inductor L, so that free oscillations are not generated due to the electrical or magnetic energy present in the inductor L2 and capacitor C.

Such a scan period without oscillations makes the current through the inductor L, equal to zero during the entire scan period and thus also equal to zero at the beginning and end of the flyback period, and the voltage across the capacitor C, It can be obtained by making it equal to the battery voltage -E. In order to satisfy this condition, the following two relationships must be satisfied for each resonant angular frequency α of the flyback period circuit network, that is, the circuit network when the switch SW is open.

In these relations, τ is the flyback period, K is an odd integer, IO is the sawtooth current value at the beginning of the flyback period, i'o is the time derivative of this current at the beginning of the flyback period, and φα is the phase. It is a corner.

φB can be eliminated from both equations. In this case, ατ is expanded in a series sequence of τi'o/IO. If we limit this salvage number sequence to the first two terms, then 1. ' becomes.

When a perfectly linear sawtooth current flows through the inductor L, and thus the deflection coil, the derivative (slope) i' of this current during the scan period is constant, and the sawtooth current d-
Since the slope at the beginning of the line period is always equal to the slope at the end of the scan period and i'o=i', IO is obtained, and when both sides are multiplied by 1, τ holds.

Here, T-τ is the scanning period. However, if the deflection current has the slightly S-shaped characteristic customary in television receivers as a result of the S-shaped correction capacitor in FIG. 1 (which is ignored in the equivalent circuit in FIG. 2), approximately To. 9, holds true for ′τ1.

where S is the relative attenuation of the slope of the deflection current at the end of the scan period with respect to the slope of the deflection current at the center of the scan period. In this case the above condition for ατ changes to α.

fα=・The element becomes.

In order to obtain vibration-free scanning, formula (1) is used as described above.
and (2), and therefore equation (3) must be applied to each resonant frequency of the flyback network. The network of FIG. 2 has two resonant angular frequencies α=2πFa and y=2πFy during flyback, ie.

This network is constructed so that Fa is K=
1, and F is designed to satisfy equation (3) when K=3, K=5, etc. In addition to the frequencies fα and F, the network has a third characteristic frequency fβ, namely the frequency between Fa and F, at which the impedance of the network is at a minimum.

For the network of FIG. 2, this is the series resonant frequency of L2 and C2, ie β2=(2πfβ)2=]÷G.

The values of fα and F at the predetermined period T and the previously selected flybutter period τ are determined by the conditions given by equation (3). However, fβ can be freely selected between F6 and F. It is known that for the values fα and F, determined according to equation (3), the value of fβ determines the shape of the flyback pulse generated across the input terminals of the network.

This is shown in FIG. 3 by curve 1 for various values of fβ.
As shown in this figure, for low values of fβ, voltage V1 decreases below the battery voltage value before the end of the flyback period.
To prevent this, fβ is generally chosen large enough,
In particular, fβ is located within the range of Vteiwagoβ(F,).
Fβ to fα and fβ! f (:Tf,), it is necessary to use switching devices in a flyback EHT and sawtooth current generator so that these switching devices do not conduct before the end of the predetermined flyback period τ. It is clear that the position of the frequency fβ also significantly influences the amplitude of the vibrations occurring during the scanning period.

This oscillation occurs because a vibration-free scanning period cannot be achieved due to the disturbance effects mentioned above. The influence of this fβ on the vibration amplitude is shown in FIG. 3 by a curve ER. This curve shows the ratio of the energy producing the undesired oscillations during the scanning period to the total electromagnetic energy of the network during the flyback period as a function of fβ. As is clear from this, this energy ratio has a maximum value at a predetermined value of fβ, and a minimum value at a value located between them. The maximum value of the ER curve occurs at a value of fβ that satisfies .

However, since πfβ(T−τ) is greater than 1, the maximum value occurs approximately at .

This is the case when fβ(T−τ)=n+% (n=integer).

The minimum value located between the maximum values occurs at a value of fβ that fits .

The curve in Figure 3 has a flyback ratio of τ / 0.15 and fα
is determined for a flyback EHT/sawtooth current generator that satisfies equation (3) when K=1 and f satisfies equation (3) when YK=3 (third high frequency tuning).

FIG. 3 shows the values of n associated with various minimum values. It can be seen that in order to obtain both a primary flyback pulse of satisfactory waveform and a scanning oscillation of minimum amplitude, n must be chosen equal to 7 in this case. It is clear that designing fβ as described above provides two important advantages.

(1) For the given nature and magnitude of said disturbing influences, which make precise tuning of the sawtooth generator impossible, the amplitude of the scanning oscillations caused thereby is kept as small as possible.

In other words, for a given permissible amplitude of scanning vibration, fairly large deviations of the cage and F from their exact tuning values can be tolerated. (2) Variations in fβ caused by various manufacturing tolerances and variations in product characteristics on the same production line have little effect on the amplitude of the scanning vibration.

On the other hand, if FB is n-7 between the maximum value and minimum value, for example
4, for example, when fβ= ]7, fβ
Manufacturing variations greatly affect the amplitude of scanning vibration.

The curve marked ER in Figure 3 is determined for a network with a certain amount of resistive loss (Q approximately 20) in the oscillating circuit consisting of E, SW, L, , C2 when the switch is closed. .

It was confirmed that when these losses are reduced, the maximum value of the ER curve increases and the minimum value decreases. By reducing the resistive losses, especially the copper and iron losses of the transformer 1, the scanning vibration can be further reduced by designing fβ to the above-mentioned optimum value. If fβ is not optimally designed, such measures will result in increased scanning oscillations. It is preferred to select the Q of the scanning circuit to be 25 or higher to obtain optimal results using the correct design of fβ. It is confirmed that if fβ is designed correctly in conjunction with Q of 25, the vibrational energy ratio (ER) will be only about H% for a relatively large deviation (5%) of the ratio Fy/fα from its optimum value. This Q can be easily determined from the exponential decrease in scanning oscillation during the scanning period.

The maximum values in the ER curve located on both sides of the minimum value≠← are located at approximately fβ=蓓; and 卆ι.

What is the interval between the minimum and maximum values for this? It is. Furthermore, FIG. 3 shows that it is necessary not to deviate more than about ±i=F from the appropriate value of the fβ force field 6, so that it is sufficiently located within the minimum value of the ER curve. This makes T-τ about 55
For a television receiver set in μSec, the permissible deviation of fβ is approximately ±2.3 KHz. As described above, absolute precision (KHz) is required for fβ, and relative precision is not required. Corresponding elements in the example of FIG. 4 are designated by the same reference numerals as in FIG.

Compared to the example of FIG. 1, the example of FIG.
It comprises an additional transformer winding 5 connected via 17 to the collector electrode of the transistor 12. The equivalent circuit diagram is shown in FIG. In this equivalent circuit diagram, L3 and C3 indicate the inductance of the coil 16 and the capacitance of the capacitor IT, and L4 is the leakage inductance between the windings 4 and 5. As explained in the specification of Japanese Patent Application No. 47-39236 filed by the applicant, the circuit network of FIG. By selecting the wave numbers so as to satisfy equation (3) where K is an odd integer, for example, K=1.5 and 7, an almost vibration-free scanning period can be obtained.

In this case, the network has two frequencies ψ and fβ, at which the impedance of the network is minimum zero, and the first
The frequency fβ, is located between fα and F6, and the second frequency fβ, is located between F, and Fy.

Due to the disturbance effects mentioned above, some vibrations also occur in this case during the scanning period. These vibrations consist of a first component of frequency fβ, and a second component of frequency fβ. Within predetermined limits (±i d), fβ is an integer multiple of the reciprocal of the duration of the scanning period. ; Nli is an integer), the amplitude of the first component can be adjusted to a minimum value, and similarly the amplitude of the first component can be adjusted to a minimum value by choosing fβ, which is the inverse of the duration of the scanning period, within The amplitude of the second component can also be adjusted to a minimum by making it equal to an integral multiple of the numerical value (m is an integer).

In the equivalent circuit diagram of FIG. 5, fβ can be increased by decreasing L3 and L, increasing C and C3, and slightly adjusting other elements, leaving the other frequency positions unchanged. realizable.

Increasing Flj, can be achieved by decreasing C3, increasing C, and adjusting other components slightly.

[Brief explanation of the drawing]

FIG. 1 is a circuit diagram of a first example of a flyback EHT and sawtooth current generator according to the present invention, FIG. 2 is an equivalent circuit diagram of the circuit in FIG. 1, and FIG. 3 is a diagram explaining the principle of the present invention. FIG. 4 is a circuit diagram of a second example of a flyback wave/sawtooth current generator according to the present invention, and FIG. 5 is an equivalent circuit diagram of the circuit of FIG. 4. 1...Transformer, 2...Primary winding, 3...Auxiliary winding, 4...Secondary winding, 5...Additional transformer winding, T
... Voltage supply source, 9... Deflection coil, 10... Linearity corrector, 11... S-correction capacitor, 12...
・Transistor (switching device), 13... Capacitor, 14... Rectifier, 15... Smoothing capacitor, 16, IT... Parallel LC circuit, 18... Separation transformer, 19... Series inductance , 20...parallel diodes.

Claims (1)

[Scope of Claims] 1. A circuitry having at least one switching device that periodically becomes non-conductive during a flyback period τ and conductive during a scanning period T-τ, and an input terminal connected to the switching device. a primary winding, one or more coils connected thereto through which a sawtooth current flows during a scan period, and a circuitry including a transformer having a secondary winding, and a rectifier circuit in the secondary winding. A flyback EHT and sawtooth current generator connected to obtain a flyback EHT from voltage pulses generated in the secondary winding during a flyback period, the circuitry comprising: a transformer winding; Due to the leakage inductance present during the flypack period, the formula: (K)/(2π){1+[4]/[π^2K^2][τ
]/[T-τ](1-[2]/[3]S)} where K is an odd integer S is the slope of this current at the end of the scan period relative to the slope of the sawtooth current at the center of the scan period. The first resonance frequency f_α is approximately equal to the value when the correction coefficient K, which is equal to the relative decrease amount of the slope, is 1.
and has a second resonant frequency f_y that is approximately equal to the value when K in this equation is an odd integer greater than 1, and the circuit network is located between frequencies f_α and f_y, A flyback EHT/sawtooth current generator having a frequency f_β at which the impedance at the input terminal is minimum, and the frequency f_β being approximately equal to an integer multiple of the reciprocal of the duration of the scanning period.