DE2258132C2 - Flyback high voltage and sawtooth generator - Google Patents

Description

where K = 1 and 5 is a correction factor that corresponds to the relative decrease in the slope of the sawtooth current at the end of the trace with respect to this inclination in uer center of the trace period, and a second resonant frequency fy, the said expression mi · K is equal to an odd integer greater than 1 at least almost corresponds, wherein in said network at a frequency fß lying between fx and fy the impedance of the network at the input terminals is minimal, characterized in that the network is tuned so that the frequency fß is approximately a whole multiple of the reciprocal of the duration of the trace time (T-τ)

2. Flyback high voltage and sawtooth current generator according to claim 1, characterized in that the quality factor of the generator is more than 25 during the trace time at said frequency fb.

3. flyback high-voltage and sawtooth current generator according to claim 1 or 2, wherein said network has a third, between Sx and fy resonance frequency k during the flyback time, the said expression for K almost corresponds to an odd integer, characterized in that the first frequency fß \ lying between fm and k , the impedance of the network at the input terminals being minimal, almost corresponds to a whole multiple of the reciprocal value of the duration of the trace time

4. Return voltage and sawtooth current generator according to claim 3, characterized in that the second frequency lying between A and fy fβi, the impedance of the network is minimal at the input terminals, too almost corresponds to a whole multiple of the reciprocal value of the duration of the travel time

The invention relates to a flyback high voltage and sawtooth current generator, in particular for a television display arrangement with switching means which are periodically non-conductive during a flyback time ν and conductive during a trace time T-v and with a network with input terminals connected to the switching means, the network having a Transformer contains at least one primary winding, with possibly one or more coils connected to it, through which said sawtooth current flows during the ramp-down time, as well as with a secondary winding to which a rectifier circuit for generating the aforementioned high voltage from the voltage pulses occurring in the secondary winding during the ramp-down time is connected, which network, also as a result of the leakage inductance existing between the transformer windings, is tuned so that it has a first resonance frequency foe during the flyback time, which is at least almost the Exdr uck corresponds to:

- ί

1 +

K ² Τ-ι

where K = X and 5 is a correction factor which corresponds to the relative reduction in the inclination of the sawtooth current at the end of the run-out time compared to this inclination in the middle of the run-out time, as well as a second resonance frequency fy, which is greater than the mentioned expression with K being an odd whole number than 1 at least almost corresponds, with the impedance of the network at the input terminals being minimal in the said network at a frequency fß lying between fix and fy

From DE-PS 9 77 115 it is known in one Flyback high voltage and sawtooth current generator for a television display device to use a transformer with only one or more Primary windings and with a secondary winding and the impedances of these circuit elements present, such as the transformer windings, the leakage inductance between the primary windings and the Secondary winding, the inductance of the coils mostly connected to the primary winding and the parasitic as well as non-parasitic capacitances to be dimensioned in such a way that no or there are few oscillations in the secondary voltage. Such oscillation phenomena / have the disadvantage that useful energy is lost because this useless energy is mainly absorbed in the transformer is, whereby overheating of the transformer can occur, and that in the case of large oscillation phenomena, the circuit means that are conductive in the delay time can become non-conductive at an early stage.

In the cited patent it is stated that the cited network forms a fourth order network value with two resonance frequencies during the flyback time and that the cited oscillation phenomena can be kept small by dimensioning the circuit elements so that these two parallel resonance frequencies fet and fy have specific values , which are dependent on the duration of the return period τ and the duration of the trace period TT. As can be seen from the above expression, the optimal value of / 5x and fy is also to some extent of the extent to which the inclination of the sawtooth current through the deflection coils

15th

when using so-called S-correction of this current varies, dependent.

Theoretically, a completely vibration-free trace can be obtained in a high-voltage and sawtooth current generator by precisely dimensioning fx and fy. In practice, however, this is not completely achievable due to unavoidable disruptive influences. These disruptive influences include:

1. The presence of scatter actaazes in the network, which on the one hand are so small that they can be not in the hand, but on the other hand large enough to make a vibration-free run impossible.

2. The presence of losses in the network during the retrace time. An essential part of this is formed by useful energy that is taken from the rectifier.

3. Tolerances in individual parts and production variations which make it impossible to make the frequencies fx and fy exactly the same as the values necessary for a vibration-free trace.

The invention is based on the object in one. Generator of the type mentioned to improve the suppression of spurious vibrations; for this purpose it has the characteristic that the network is tuned in such a way that the frequency fb corresponds approximately to a whole multiple of the reciprocal value of the duration of the trace time

Said network always has a series resonance frequency fß lying between the two parallel resonance frequencies fx and fy , the impedance of the network at the input terminals being minimal (in the case of a completely lossless network, zero). This is essentially the oscillation frequency of the generator during the run-out time, that is to say with conductive switching means. It is known that for precise adjustment of the return high voltage generator fx on a vibration-free trace only the parallel resonance frequencies and fy are important and the series resonant frequency / is not essential It has the location of fiB only an influence on the shape of the flyback pulse occurring at the input terminals of the network .

The invention is based on the finding that, however, the value dsr frequency also fiB a huge infiuence on the amplitude of the result dei mentioned disturbances still remaining Hinlaufschwingungen has and that this amplitude is minimal when the value of fp? Whole INEM multiples of the reciprocal almost corresponds to the duration of the run-out time

Such a generator is preferably designed in such a way that the quality factor of the generator during the trace time at said frequency fb is more than 25

If with a high voltage and sawtooth generator the return network is at least has three parallel resonance frequencies or more, In this case, too, an almost vibration-free approach can be achieved in that each of these resonance frequencies is at least almost the following Expression is liked the same:

{■ ♦ •.-VrM'-i ')}

65

where K is an odd positive integer, for example 1,3,5 or 1,5,7. In such a case the network has a first series resonance frequency fβ \ between fx and k , the input impedance of the network being minimal and between k and fy a second series resonance frequency fßi, where the input impedance of the network is minimal. fß1 and fß2 are also the frequencies to which the network resonates during the trace time

As a result of the disruptive influences mentioned, the outward travel will not be completely free of vibrations even in the case of such a high-voltage return generator. The vibrations that then occur usually contain two parts, one with the frequency fß \ and one with the frequency Ιβϊ.

It now turns out that, independently of one another, for a minimum amplitude of the / Pi component, the frequency fß \ optimal, i.e. approximately corresponding to a whole multiple of the reciprocal value of the duration of the trace time, must be selected, and for a minimum amplitude of the / ^ - Part of the frequency f $ i must be approximately equal to a multiple of the reciprocal value of the duration of the outward air time.

On the other hand, it turns out that the component with the lower frequency generally has a significantly greater amplitude than the component with the higher frequency. In the case of a flyback high-voltage and sawtooth current generator, with the aforementioned network having a third resonance frequency k between fx and fy during the flyback time, which corresponds at least to the given expression for K corresponding to an odd integer, a significant reduction in the trace oscillations is therefore already achieved if the one between fx and fe lying frequency fß \ where the impedance of the network at the input terminals is minimal and corresponds approximately to a whole multiple of the reciprocal value of the duration of the trace time

As already mentioned above, on the other hand, an optimal dimensioning is achieved by the fact that the second frequency fßr lying between ft and fy , the impedance of the network at the input terminals being minimal, also corresponds to a whole multiple of the reciprocal value of the duration of the trace time

Embodiments of the invention are shown in the drawings and are described below described in more detail. It shows

F i g. 1 shows a first embodiment of a flyback high voltage and sawtooth current generator, wherein the invention can be used,

F i g. 2 the salt circuit diagram of the exemplary embodiment according to Fig. 1,

F i g. 3 shows a diagram to explain the present invention;

4 shows a second embodiment of a return high voltage and sawtooth current generator, wherein the invention can be applied,

F i g. 5 the equivalent circuit diagram of the exemplary embodiment according to FIG. 4th

The example according to FIG. 1 shows a Transformer 1 with a primary winding 2, one or several auxiliary windings 3 firmly coupled to the primary winding, and with a secondary winding 4. A tap 6 of the primary winding is connected to the positive pole of a voltage source 7, whose negative pole is connected to ground. Between the second tap 6 and the underside of the primary winding, the series connection reads from some deflection

coils 9 of a linear correction arrangement IO and an S correction capacitor II. The tap 8 and the underside of the primary winding are symmetrical with respect to the tap 6, so that said series circuit is fed symmetrically with respect to ground.

A transistor 12, which acts as a switch, is also located between the upper end of the primary winding and ground a capacitor 13 connected in parallel to it. Said secondary winding 4 is grounded on the one hand and on the other hand connected to a rectifier circuit consisting of the rectifier 14 and a Smoothing capacitor 15 is made; the high voltage generated by the rectifier becomes the end anode of a not shown in detail supplied television picture tube. ι ϊ

Switching pulses that periodically block the transistor 12 at the end of each trace time are via a Isolation transformer 18. a series inductor 19 and a Paraüeldiods 20 between the base and the Emitter of transistor 12 is supplied. The transistor 12> o is a so-called lazy switching transistor and elements 19 and 20 are incorporated around the Shutting down the transistor at the end of the trace time to accelerate.

FIG. 2 shows the simplified equivalent circuit diagram of the circuit arrangement according to FIG. 1. Therein is £ the voltage source 7, SW represents the switch formed by the transistor 12 and the diode 20. Q is the capacitance of the capacitor 13 increased by the collector-emitter capacitance of the transistor and the transformed stray capacitances of the primary winding, the Auxiliary windings, the deflection coils and the linearity correction circuit. L \ is the inductance of the primary winding and the deflection coils and linearity correction circuit connected to it, all π transformed according to the terminals to which the switch is connected. L2 is the leakage inductance between the secondary and primary winding and C2 is the leakage capacitance of the secondary winding and the input capacitance of the rectifier circuit, also transformed according to the terminals to which the switch is connected.

The switch SW is closed during the run-down time. The voltage £ of the supply voltage source 7 is therefore on the capacitor C- and at the same time on the j> inductance L 1. As a result, a (sawtooth) current that changes linearly over time will flow through the inductance L 1. If the switch SW is rendered non-conductive as a result of a pulse applied to the base electrode of the transistor 12, free oscillations will occur in the network as a result of the magnetic energy present in L 1. These oscillations cause pulse-shaped voltages Vi and V2 on the capacitors Cl and C2. the so-called return impulses. As soon as the return pulse at C1 drops to the value of the supply voltage, ie as soon as the KoHektorpotential of the transistor 12 becomes negative with respect to ground, the collector-base-pn-transition of the transistor gets in the forward direction and a next down time begins. The switch SW in the equivalent circuit diagram according to FIG. 1c therefore closes automatically as soon as the return voltage present at this switch equals zero.

It should be noted that in the circuit diagram of FIG. 1 the sawtooth current during the first part of the trace time via the diode 20. the base-KoHector junction of the transistor and then via the transformer and the deflection coils to the supply voltage source and in this way the supply voltage source returns energy. Some time after the beginning of the The base-emitter junction is determined by the pulses fed to the base electrode of the transistor of the transistor is made conductive, so that during the second part of the flyback time of the now in its polarity reversed sawtooth current from the supply voltage source via the transformer and the Deflection coils and then via the collector electrode and the emitter electrode of the transistor to ground can flow, wherein the supply voltage source supplies the network with energy.

It must now be ensured that only sawtooth current flows through L 1 during the run-out time and that no oscillation phenomena occur at the same time as a result of electrical or magnetic energy which is present in inductance L 2 and in capacitor C2. Such a vibration-free follow-up time

entire run-down time and therefore also at the beginning and at the end of the run-down time, the current through L 2 is equal to zero and the voltage at C2 corresponds to the battery voltage - E.

In order to meet this requirement, the following relationships must apply to each resonance frequency λ of the return network, i.e. the network with the switch SW open:

ff I =

a Ig Φ α =

(D

(M)

Here τ is the duration of the return time. K is an odd integer, io is the size of the sawtooth current at the start of the ramp-down time, / 0 is the time differential quotient at the start of the ramp-down time and Φ _{χ is} a phase angle. From the two equations, Φ, can be eliminated. Then for otr there is a power series in r / Vfo. If this series is restricted to the first two terms, one finds:

ar = '- +

K-

If a purely linear sawtooth current flows through L 1 and consequently through the deflection coils, the following applies approximately:

r / "

2 τ TT

where Τ-τ is the duration of the trace period If, however, as a result of the S-correction capacitor according to FIG. 1. the one in the equivalent circuit according to FIG. 2 neglected, the deflection current has a somewhat S-shaped character, which is common with television display systems, so the following applies approximately:

Τ-τ V 3

where S is the relative decrease in the slope of the

Deflection current at the end of the trace time compared to this tendency in the middle of the trace time. the The above condition for «r then changes into

ατ = Κη

"-T7

2 r I 'A'V Γ-, V' 3

/ "Λ = " II

fa _2f .1

(III)

As noted above, equations (I) and (II) and consequently also equation (III) for> n every resonance frequency of the return network apply.

ar

L, C ₁

L ₁ C ₂

L, C ₁

The network according to Fig. 2 has two resonance frequencies η = 2π f <x and γ = 2π / γ during the flyback time, for which filt:

, C ₁

L ₁ C, J

L ₁ L ₁ C ₁ C ₂

and in order to maintain a vibration-free run-out, the network is dimensioned in such a way that fix corresponds to equation (Hl) for K - 1 and at the same time fy corresponds to equation (III) for, for example, K = Z or K = 5 and so on. In addition to the frequencies fa and fy , the network also has a third characteristic frequency fß . which is the frequency between fix and fy. where the input impedance of the network is minimal (0). For the network according to FIG. 2, this is the series resonance frequency of L 2 and C2

ß- = (2 .τ // ir

Ll C2

For the given period T and the previously selected ramp-down time r, the values of fix and fy lie uürCi'l Uli. Ulli UtI G! Ci <_llUllg (!!!) gCgCUCIIC DCuillgUIlg fixed. However, the value fß can still be freely selected between fa and k. 4ϊ

It is known that when values of fa. and fy the value of / ß determines the shape of the return pulse occurring at the input terminals of the network. This is in FIG. 3 shown by the direction indicated by curves Vl closer v), which were found at different values of the FSS. As can be seen from this figure, for low values of fβ, the voltage Vl drops below the value of the battery voltage before the end of the retrace time. In order to avoid this, Iß is generally chosen to be large enough, in particular fß is in the area

/ 1öify ~ <fß <fy.

60

It should be evident that. if you are in the area fa. <fß <] / fxfy

selects such switching means must be used in the ramp-down high-voltage and sawtooth current generator that they do not become conductive before the end of the predetermined ramp-down time τ.

It has now been found that the position of the frequency fß at the same time has a significant influence on the amplitude of the oscillations occurring during the trace time, which result from the fact that a completely vibration-free trace time cannot be achieved as a result of the disturbing influences described above. This is in FIG. 3 indicated in more detail by the curve ER. As a function of Iß, this curve gives the ratio between the energy which caused the undesired oscillation in the travel time to the total electromagnetic energy present in the network during the return time. It can clearly be seen that this energy ratio has maxima at certain values of f 3 and minima at the values in between.

It has been found that the maxima in the £ 7? Curve occur at feet of feet , and good for those:

πίβ (T- r) - tangent {πίβ (T- τ)).

However, since πίβ (T- τ) is large compared to 1, it can be said that the maxima mentioned occur approximately at

tangent (π ■ fß ■ (T-τ)) = ^ Ό

This is the case when fß Γ-τ) = π + '/ 2 (η is an integer). The minima lying between the maxima occur at those values of fß for which fß (T- r) = η therefore applies

T-T

The curves according to FIG. 3 are intended for high-voltage and sawtooth current generators where foe corresponds to equation (III) for Ar = I and fy for K = 3 (third harmonic tuning) and for a flyback ratio τ1 7 "= 0.15. In FIG The values τοπ.

pulses and a minimum amplitude of the trailing oscillations in this case, preferably n = 7 .

It should be evident that the above-mentioned dimensioning of φ yields two essential advantages.

1. With the type and size of the disturbing influences mentioned, which make precise tuning of the generator impossible, the amplitude of the trailing oscillations caused thereby is kept as small as possible. Or, for a certain permitted amplitude of the trailing oscillations, significantly larger deviations from fix and fy compared to their exact tuning value can be permitted.

2. The scatter of φ in the different copies of one and the same production series caused by the tolerances and production scatter has practically no influence on the amplitude of the trailing oscillations, in contrast to when φ is, for example, in the middle between a maximum and a minimum, for example

fli =

T-T

fß =

IO

—— lying maxima in the ER curve are on

So the distance between a minimum and an adjacent vuxium is

may differ. In the case of television display

Then the production spreads in φ will have a great influence on the amplitude of the trailing oscillations.

It should be noted that the curve indicated by ER ü in Fig. 3 is intended for a network which, when the switch is closed, contains a certain amount of ohmic losses in the oscillation circuit formed by E, SW, L2, C2 (quality factor Q approximately equal to 20) . It now turns out that when these losses are reduced, the maxima of the EE curve become higher and the minim = »lower. By excluding the ohmic losses, in particular by reducing the copper and iron losses in the transformer 1, with the above-mentioned optimal dimensioning of fß , a further reduction in the trace oscillations can be obtained. If the vessel is not optimally dimensioned, such a measure will result in an increase in the inlet vibrations.

To obtain optimal benefit of the correct dimensioning of the FSS, the quality factor Q of the Hinlaufkreises is therefore preferably greater than 25 selected With proper dimensioning of FSS in combination with a P = 25 is at a relatively large (5%) deviation of the ratio h / fix compared to the optimal value of this ratio, a vibration energy ratio (ER) of only about-% was found

The quality factor mentioned can simply be derived from the exponential decrease in the trailing oscillations determine during the follow-up time. The two on either side of a minimum T-T

From FIG. 3 it is further evident that in order to lie sufficiently in a minimum of the ER curve

f no more than about ± the optimal value

voltages, where T-t can be approximately 55 μ $ , the permissible deviation from fβ is thus approximately ± 2.3 kHz. From the above it follows that an absolute (in kHz) rather than a relative (in%) accuracy is required for fß.

In the exemplary embodiment according to FIG. 4, corresponding parts are given the same reference numerals as in FIG. 1 specified. Compared to the embodiment of FIG. 1, FIG. 4 has an additional transformer winding 5, which is connected to the collector electrode of the transistor 12 via a parallel LC circuit 16, 17 with one side of the + pole of the supply voltage source 7 and with the other side is. The equivalent circuit diagram is shown in FIG. 5 shown. L 3 and Ci therein essentially represent the inductance of the coil 16 and the capacitance of the capacitor 17, while L 4 represents the leakage inductance between the windings 4 and 5. As has already been explained in the current patent application (PHN. 5578), the network according to FIG. 5 three resonance frequencies fixed, k and fy , and an almost vibration-free follow-up time can be obtained, namely that these frequencies are chosen such that they all three correspond to equation (III), where K is an odd integer, for example K equals 1.5 and 7 corresponds.

In this case the network has two frequencies φ \ and Φ2, the input impedance of the network being minimal (0), and of which the first (φ \) lies between fix and k and the second (φι) between k and fy . As a result of the disturbing influences mentioned at the beginning, oscillations will occur again to some extent during the trailing time. These oscillations are made up of a first component with the frequency φ \ and a second component with a frequency / j? ₂ exist.

If now φ \ is within the given limits

is chosen to be equal to a whole multiple of the reciprocal value of the duration of the trace time

TT

with η as an integer

the amplitude of said first component can be set to a minimum and also can be set within the given limits by φ ₂

65

T-τ

and

T-T

equal to a whole multiple of the reciprocal value

Il

the duration of the follow-up time can be selected

• - with m as an integer

the amplitude of said second component can be set to a minimum.

It should be noted that in the equivalent circuit according to FIG

an increase in / jS can be achieved with the remaining frequencies remaining the same, by reducing L 3 and LA and by increasing Cl and C3 with a slight adjustment of the other elements. An increase in / p ₂ can be carried out by reducing C3 and increasing CI with a small adjustment to the remaining elements.

For this purpose 3 sheets of paper

Claims (1)

Patent claims: 1. Return high voltage and sawtooth current generator, especially for a television playback arrangement with switching means that are periodically non-conductive during a return time τ and conductive during a delay time Τ-τ and with a network with input terminals connected to the switching means, the network containing a transformer with at least one primary winding, with possibly one or more coils connected to it, through which said sawtooth current flows during the ramp-down time, and with a secondary winding to which a rectifier circuit is connected to generate the aforementioned high voltage from the voltage pulses occurring in the secondary winding during the ramp-down time , which network, also as a result of the leakage inductance between the transformer windings, is coordinated in such a way that it fixes a first resonance frequency during the flyback time. which at least almost corresponds to the expression: