DE2258132A1 - RETURN HIGH VOLTAGE AND SAW GEAR GENERATOR - Google Patents

Description

Dr.-Inj. Hans-DietriA ZeIIw 2258132

PatcntciiKiIt M. V. Philips' GloeHampenfabrfelcea

au · «·. Ml- 6059 27 · 'lor. ^ 972

"Return high voltage · and sawtooth current donor".

The. The invention relates to a flyback high-voltage and single-tooth current generator, in particular for a television display device, with switching means which are periodically non-conductive during a flyback time T and conductive during a trace time TT and which are connected to the switchgear with an input network, the network eileiinWaMormator contains bits of at least one primary winding, or possibly one or more coils connected to it, through which the mentioned tooth current flows during the delay time, as well as bits of a secondary winding to which a Gleiohriohterschaltung zvsm generating the mentioned high voltage from the occurring in the flyback time on the secondary winding Voltage pulses are connected, which · network also as a result of the • leakage inductance between the transformer windings during the flyback time has a first resonance frequency f oC, which at least almost corresponds to the following expression!

309826/0761

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PHB.6059

where K - 1 and S is a correction factor iet which corresponds to the relative reduction in the slope of the sawtooth current as the end of the trace time compared to this slope in the middle of the trace time, as well as a second resonance frequency ty , which corresponds to the mentioned expression pressure for K corresponding to an odd integer greater ale at least almost corresponds to 1.

It is known from Dutch patent specification 88020 » to use one transformer with only one in flyback high voltage and sawtooth current generators for television display devices or more primary windings and with a secondary winding and the Impedances of these existing circuit elements, such as the transformer * windings, the leakage inductance between the primary windings and the secondary winding, the inductance of the coils mostly connected to the primary winding, as well as the parasitic and non-parasitic capacitances are to be dimensioned in such a way that little or no oscillations occur in the secondary voltage during the run-out time. Such oscillation phenomena have the

Naohteil on that useful energy is lost, that dl · ·· useless ·

Energy is mainly absorbed in the transformer, creating a Overheating of the transformer can occur and therewith large Excessive oscillation can be diverted at an early stage by the maintenance means that are conducting in the run-out time.

In the literature above, it is shown that the network referred to as billing a 4th order network during the return time with swel resonance frequencies and that the mentioned oscillations can be kept small by the fact that the circuit element · are dimensioned in such a way that these two resonance frequencies

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f <K and ty have very specific values which depend on the duration of the return period T * and the duration of the trace period T - £ *. As can be seen from the expression given at the beginning, the optimal value of f is d. and ty also depends to some extent on the extent to which the inclination of the sawtooth current through the deflection coils varies when a so-called S correction of this current is used.

Theoretically, a completely vibration-free trace can be obtained in a high-voltage and sawtooth current generator by precisely dimensioning f OC and f y. In practice, however, this is unattainable due to unavoidable disruptive influences. These disturbing influences are among others

1. The presence of stray reactances in the network, which on the one hand are so small that you cannot control them, on the other hand but large enough to make a vibration-free outward run impossible.

2. Las presence of losses in the baiting during the Return time. A significant part of this is made up of useful energy formed, which is taken from the rectifier.

3. Tolerances in individual parts and production spreads "which make it impossible to make the frequencies fet and Sy exactly enough to the values necessary for a vibration-free trace].

The aim of the invention is to create a dimensioning measure with which, in spite of the disturbing influences mentioned, the outgoing vibrations that occur can still be kept small and the high-voltage and sawtooth current generator according to the invention has the characteristic that the network mentioned has an intermediate f ot and ty has a frequency f fl , the impedance of the network at the input terminals being minimal, which is about a whole

309825/0761 bad original

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times the reciprocal value corresponds to the duration of the trace time.

Said network always has a series resonance frequency T / B lying between the two parallel resonance frequencies f ot and fy, the impedance of the network at the input terminals being minimal (zero in the case of a completely lossless network). This is essentially the oscillation frequency of the generator during the trace time, that is to say when switching means are conductive. It is known that only the parallel resonance frequencies f oc and f y are important and the series resonance frequency f is not essential for the exact tuning of the return high-voltage generator to a vibration-free trace. The position of f / 3 only has an influence on the shape of the return pulse occurring at the input terminals of the network.

The invention is based on the knowledge that the value of the frequency f / 3 also has a very large influence on the amplitude of the trailing oscillations still remaining as a result of the disturbing influences mentioned and that this amplitude is minimal if the value of f / 3 is a whole Multiples of the reciprocal value almost corresponds to the duration of the trace time.

Such a generator is preferably designed in such a way that the quality factor of the generator during the trace time in the mentioned frequency f / 3 is more than 25.

In a current patent application (ΓΗΝ.5578) de * "Applicant, high-voltage and sawtooth current generators have been described, the return network having at least three resonance frequencies f Λ, f £ and f γ or more in that each of these resonance frequencies is at least almost made equal to the following expression "

BATH ORIGINAL

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2 VI 7 ^ 2 _T , 2 T-

(1-fs)}

where E is an odd positive integer, for example 1, *> ·, 5 or 1, 5> 7. In such a case, da3 Netzw§rk between f and f oc £ f a first series resonance frequency / S _Λ, wherein the input impedance of the network is minimal and between f e and f> a second series resonance frequency f / _3? , whereby the input impedance of the network is minimal. f / 3. and f / S _? are also the frequencies to which the network resonates during the trace time.

As a result of the disturbing influences mentioned, is also with With such a reverse voltage generator, the trace is not completely be vibration-free. The vibrations that then occur usually contain two parts, one with the frequency f / S. and one with the frequency f / 3 ".

It now turns out that, independently of one another, for a minimum amplitude of the ΐ / 3rd component, the frequency f / 3. optimal, that is to say it must be chosen according to a whole multiple of the reciprocal value of the duration of the trace time and for a minimum amplitude of the f / f "component the frequency f / 3 η must be approximately a whole multiple of the reciprocal value of the duration of the trace time . . - '

On the other hand, it turns out that the component with the lower frequency generally has a significantly greater amplitude than the component with the higher frequency. In a flyback high voltage and sawtooth current generator , the said network having a third resonance frequency f £ between f oL and f ν during the flyback time, which corresponds at least to the given expression for K corresponding to an uneven whole number, was therefore already a significant reduction of the trailing oscillations reached,

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when the ot between f and f £ ^ A frequency lying wherein said impedance network is dea miniirel to the Eingnngskleninen, approximately an integral multiple of the reciprocal of the duration of the trace corresponds.

As already mentioned above, on the other hand, an optimal dimensioning is achieved by the fact that the second frequency between ff and ty is f / 3 _? , where the impedance de. ^c ; Network at the input terminals is minimal, also corresponds to a whole multiple of the reciprocal value of the duration of the trace time.

Embodiments of the invention are in the drawings and are described in more detail below. 3a show

Fig. 1 shows a first embodiment of a return high voltage and saw tooth current generator, wherein the invention is applied can be,

Fig. 2 shows the Eraatz circuit diagram of the embodiment Fig. 1,

Fig. 3 is a diagram: to explain the present invention,

Fig. 4 shows a second embodiment of a return high voltage and sawtooth atomic generator to which the invention is applied can be,

5 shows the equivalent circuit diagram of the embodiment example according to Fig. 4.

The embodiment of Fig. 1 shows a transformer 1 with a primary winding 2, one or more fixed to the Primary winding coupled auxiliary winding 3, and with a secondary winding 4 · A tap β of the primary winding is rit the positive pole of an S; ice voltage source 7 connected, the Kinuapol connected to! lasse i3t. Between the second tap 8 and the bottom of the

3 0 9 8 2 S / 0 7 6 1 ■ ^{; BA} ® original

: Γ -Γ 2259132

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The series connection consists of some deflection coils 9 a linearity correction arrangement 10 and an S correction capacitor 11. The tap 8 and the underside of the irimary winding are opposite the tap 6 symmetrically, so that said series connection • is fed symmetrically with respect to ground.

Between the upper end of the primary winding and ground lies a transistor 12 acting as a switch with one connected in parallel with it Capacitor 13. Said secondary winding 4 is on the one hand connected to ground and, on the other hand, to a rectifier circuit consisting of the rectifier 14 and a smoothing capacitor 15; the high voltage generated by the rectifier is fed to the end anode of a television picture tube (not shown).

Switching pulses that periodically end the transistor 12

lock each trace time, are via an isolating transformer 18, a Series inductance 19 and a parallel diode 20 between the base ″ and fed to the emitter of transistor 12. The transistor 12 is on so-called lazy switching transistor and the elements 19 and 20 are included to switch off the transistor at the end of the trace time accelerate.

2 shows the simplified equivalent circuit diagram of the circuit arrangement according to FIG. 1. Here, E is the voltage source 7 »SW is the switch formed by the transistor 12 and the diode 20. C ₁ is the capacitance of the capacitor 13, increased by the collector - Emitter capacitance of the transistor and the transformed stray capacitances of the primary winding, the auxiliary windings, the deflection coils and the linearity correction circuit. L ₁ is the inductance of the primary winding and the deflection coils and linearity correction circuit connected to it, all transformed according to the terminals to which the>

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Switch is connected. L "is the leakage inductance between the Secondary and primary winding and C "is the stray capacitance of the secondary winding and the input capacitance of the rectifier circuit, also transformed according to the terminals to which the switch is connected is.

The switch SW is closed during the follow-up time. The voltage E of the supply voltage source 7 is therefore across the capacitor C ₁ and at the same time across the inductance L1. As a result, a (sawtooth) current that changes linearly over time will flow through the inductance L1. Will one of the base electrode of transistor 1? supplied pulse of the switch SW not made conductive, free oscillations will occur in the network as a result of the magnetic energy present in L1. These oscillations cause pulse-shaped voltages V1 and V2, the so-called return pulses, on the capacitors C1 and C2. As soon as the return pulse at C1 drops to the value of the supply voltage - ^, ie as soon as the collector potential of the transistor 12 becomes negative with respect to ground, the collector-base-pn-junction "of the transistor" turns into the forward direction and a next delay time begins. The switch SW in the equivalent circuit diagram of FIG. 1c, therefore closes automatically as soon as the presence of this switch flyback voltage becomes zero. _#

It should be noted that in the circuit diagram according to FIG. 1, the sawtooth current during the first part of the trace time via the diode 20, the base-collector junction of the transit gate and then via the transformer and the deflection coils to the supply voltage source flows and opens this way the supply voltage source supplies energy. Some time after the start of the trace time, the base electrode of the transistor supplied pulses of the base-emitter junction of the

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Transistor made conductive so that during the second part of the
Ramp-down time of the sawtooth current, now reversed in polarity, from the supply voltage source via the transformer and the deflection coil and then via the collector electrode and the emitter electrode of the
The transistor can flow to ground, the supply voltage source supplying the network with energy.

Fun must be ensured that only sawtooth current flows through L1 during the run-out time and that no exchange phenomena occur at the same time as a result of electrical or magnetic energy that is present in the inductance L2 and in the capacitor C2. Such a vibration-free follow-up time is achieved if it is ensured that during the entire run-in time and therefore also on
At the beginning and at the end of the flyback time the current through L2 is zero and the voltage at C2 corresponds to the battery voltage -E.

In order to meet this requirement, each resonance " frequency £ X of the return network, i.e. the network when the network is open Switch SW, the following relationships apply

ar - kV + 2JZf ₀₀ '"' (ι) '

■■ ".: ■ - i '

Ätgfi oc = -2. - ■■ - · - ■ .. ■ -.- ■■ (Π)

O,

Here, t is the duration of the return time, K is an odd whole number, i is the size of the sawtooth current at the beginning of the return time, i ^{1 is} the time differential quotient at the beginning of the return time and φ ^ a
Phase angle. From the two equations, φ ^ can be eliminated.
Then for et V a power series arises in Tl ¹ / i, If this series is restricted to the first two terms, one finds:

• 2- i · «- '■'■" ■■■■ ^:: ' ^: ' ■ ■ "■ '■ - '·" ■■■

K TT ₊ ßj

Y. fr - · i. ο

If through L1 And consequently through ?: the deflection coils a purely "linear"

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SSg # tooth current flows, the following applies approximately:

^Γι Ό 2 t I ₀ - τ - r

where T-Z * is the duration of the trace period. However, if as a result of the S-correction capacitor according to FIG. 1, which is neglected in the equivalent circuit diagram according to FIG. 2, the deflection current is somewhat S-shaped Has character, which is generally customary in television display devices is, then approximately applies:

T ν

⁰ * i- ~ (1 -Is)

i _o τ -r - 3

where S is the relative decrease in the inclination of the deflector 3 at the end represents the trace time versus this slope in the middle of the trace time. Pie above condition for going off then over in

^X1r f

withTO.- follows

As noted above, must be used to maintain a vibration-free The equations (i) and (il) and consequently also the Equation (Hl) apply to each resonance frequency of the return network.

The network according to FIG. 2 has two resonance frequencies 0C- 2Tf fau and y · 2 Tf ty during the flyback time, for which purpose slipped

2Ö / T f Tf * TfI if A ITf f pt τ η * T T f f

yt- / UjtKt * lirtO— AJrt ^^ IV ^ X. - "- ¹ I ^ i OO JLjn ^ 4 / ^ ³ Λ O 1

and in order to maintain a vibration-free path a, d, « ⁸ network is dimensioned in such a way that f (X corresponds to equation (ill) for K ■ 1 and at the same time fy to equation (ill) for, for example, K = 3 or K * 5 us * ,, .

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Except for the frequencies f ex. and fy, the network also has a third characteristic frequency f / β, which is the frequency between f CX and f /, the input impedance of the network being minimal (O). For the network according to FIG. 2, this is the series resonance frequency of L2 and C2, ie A ^z «(2 Tf t * ) ² =. Given the Feriode T and the previously selected ramp time T, the values are f (X and iy determined by the given with the equation (ill) condition. F The value / $, however, may be between foe. And f £ still freely selected will.

It is known that for values of f «- and fy determined according to equation (Hl), the vert of f / S determines the shape of the return pulse occurring at the input terminals of the network. This is illustrated in more detail in FIG. 3 by the curves indicated by VT, which were found at different values of f / 3. As can be seen from this figure, for low values VQn f / 3, the voltage V1 drops below the V.'ert of the battery voltage before the end of the ramp-down time. To avoid this, f / 3 is generally chosen to be large enough, in particular f / 3 lies in the region yf »of y <C ϊλ / ^ iy .

it should be evident that if one f /?. in the region f «. 4. f / 3 <. V f «- selects such switching means must be used in the flyback high voltage and sawtooth current generator that they do not become conductive ^{before the end of the predetermined flyback time 2" 1.}

It has now been found that the location of the frequency f / 3 at the same time has a significant influence on the amplitude of the during the follow-up time has occurring vibrations, which arise as a result, that as a result of the initially described disturbing influences a completely vibration-free follow-up time is unattainable. This is shown in Fig. 3 by the curve ER indicated in more detail. This curve gives as a function of f / 3

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the ratio between the energy that is said in the trace has caused undesired oscillation compared to the total electromagnetic energy present in the network during the flyback time. It is clear that this relation of energy should be kept with certain Values of f / 3 maxima and at the intermediate values Has minima.

It was found that the maxima in the ER curve occur at values of f / 3, for which applies

7rf /?> (T - T ) - tangent (Vf / 1 (T- Z) J.

However, since TT f / 3 (T - V ) is large compared to 1, it can be said that the mentioned kaxima occur approximately at

tangent ^ iT . f / 3. (Ί -?) \ & C ^ f

This is the case if Ϊλ (Τ - £ *) ■ η + 3/2 (n is an integer). The minima lying between the maxima occur at those values of f ^ 3 for which fyg (T - 2 "*)« η, ie f / 3 *, applies.

The curves according to FIG. 5 Bind are determined for reverse high-voltage and SSgezahnstromgenerator, where f pt corresponds to equation (Hl) for k 1 and f y for K - 3 (third harmonic tuning) and for a flyback ratio Γ / Τ - 0.15 . The values of η associated with the different minima have been entered in FIG. 5. It turns out that in order to maintain a good shape of the primary return pulse and a minimum amplitude of the trace oscillations, η »7 is preferably selected in this case.

1. Given the type and size of the disruptive ones mentioned Influences that make exact coordination of the generator impossible, the amplitude of the resulting trailing oscillations is as possible kept small. Or, for a certain permitted amplitude of the trailing oscillations, there can be significantly larger deviations from

It should be clear that the above-mentioned measurement of f / S maintains two essential advantages.

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f 0 <r and fy are allowed in relation to their exact tuning value. 2, ·. The spread of f / 3 in the different copies of one and the same production series caused by the tolerances and production spread have practically no influence on the amplitude of the downward oscillations, this is in contrast to when £ / 3 · for example in the middle between a maximum and a Is the minimum, for example f / S =. Then the production variations in f / 3 will have a great influence on the amplitude of the trailing oscillations.

It should be noted that the curve indicated by ER in Fig. 3 is intended for a network which, when the switch is closed, contains a certain amount of ohmic losses in the oscillating circuit formed by E, SW, L2, C2 (quality factor Q approximately equal to '2G) . It now turns out that when these losses are reduced, the maxima "of the ER curve become higher and the minima lower. By reducing the ohmic losses, in particular by reducing the copper and iron losses in the transformer 1, with the above-mentioned optimum * measurement of f / 3 a further reduction of the trailing vibrations can be obtained. In the case of not optimally dimensioned £ / Z , such a measurement assumption just leads to an increase in the sheeting vibrations. · ■ ·

Therefore, in Kombin.at.ion with one of the Qaalitätsfaktor Q is an optimum ^ benefit of correct Be'messung zu.erhalten of f / 3 'fj / 3 selected Hinlaufkreises preferably larger than 25 * examples proper dimensioning before,' the. Q «■ 25 becomes in the case of a relatively large (5 ? ·) Deviation _; 'of the ratio f £ / f oc compared to the optimal · ^: value of this ratio * a vibration energy ratio (3R) of only titwa ^ ^ found. The mentioned quality factor can easily be. exponential increase in the in-flight oscillations during the

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Determine the follow-up time.

Both on either side of a minimum ■%>. lying maxima in the ER curve lie at about t / 3 - »and. The distance between a minimum and an adjacent maximum is therefore • * 7. From Fig. 3 it is further apparent that in order to be sufficiently in a minimum of the ER curve f A no more than about '_ + ^ r

'may deviate from the optimal value ~ = r. When watching TV

arrangements, where T - V can be about 55 / US, the permissible deviation from f / 3 is about + _ 2.3 kHz. From the above it follows that for f / 3 an absolute (in kHz) and not a relative (in '/ <} accuracy is required.

In the exemplary embodiment according to FIG. 4, corresponding parts are given the same reference numerals as in FIG. 1. Compared to the embodiment according to Fig. 1, Fig. 4 still zusStzliche transformer winding 5, which with one side of the + pole of the supply voltage source and with 7 the other hand via a parallel-LC-circuit i6 _t 17 to the collector electrode of transistor 12 is connected is. The substitute circuit diagram is shown in FIG. L3 and C3 therein essentially represent the inductance of the coil 16 and the capacitance of the capacitor 17, while L4 represents the true inductance between the windings 4 and 3. As already explained in the current patent application (IHN. 557β), the network according to Pif. 5 <* ^re i resonance frequencies f oil , ff and ty , and an almost vibration-free follow-up time can be obtained by choosing these frequencies in such a way that all three of them correspond to equation (Hl)! where K is an odd integer, for example K equals 1, 5 and 7.

The network in this case has two frequencies f / S. and

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f Λ ρ, where the input impedance of the network is minimal (θ), and of which the first (f / 3 /) is between foe and f £ and the second (f / 3 _) between f £ and fr. As a result of the disturbing influences mentioned at the beginning, some vibrations will occur again during the Hinalfu time. These oscillations consist of a first component with the frequency f / 3 .. and a second component with a frequency f A ₂ .

If now f λ, within the given limits (+ -. —— τ \ / ι. — 81-t '

is chosen equal to a whole multiple of the reciprocal value of the duration of the follow-up time (with η as an integer) the amplitude of the mentioned first component can be set to a minimum and also by f fi "within the given limits (jh -χ equal to a whole multiple of the the reciprocal value of the duration of the follow-up time can be selected (■ = - ~ r with m as an integer), the amplitude of said second component can be set to a minimum.

Ea, it should be noted that in the equivalent circuit according to FIG Increase of f / 3., "Realized with the same position of the other frequencies by decreasing L3 and L4 and increasing C1 and C3 with a slight adjustment to the rest Elements. An increase of ~ -f / 3 "can be achieved by decreasing C3 and increasing C1 with a small adjustment to the remaining elements carry out. -

3C982E./Q761

Claims (3)

- 44 - him. 6059 Claims My flyback high voltage and sawtooth current generator, in particular for a television display arrangement with switching means that are periodically non-conductive during a flyback time V and conductive during a delay time T - f and with a network with input terminals connected to the switching means, the network containing a transformer with at least one Primary winding, with possibly one or more coils connected to it, through which said sawtooth current flows during the ramp-down time, as well as with a secondary winding to which a rectifier circuit for generating the aforementioned high voltage from the voltage pulses occurring in the secondary winding during the ramp-down time is connected, which Network, also as a result of the leakage inductance present between the transformer windings, has a first resonance frequency f 0t during the flyback time, which at least almost corresponds to the expression entepri <jhti K 2 Γ where K - 1 and 3 is a correction factor which corresponds to the relative reduction in the slope of the sawtooth current at the end of the trace time compared to this slope in the middle of the trace time, as well as a second resonance frequency f V, which, in the above expression, equals an odd integer for K greater than 1, characterized in that said network has a frequency f / $ lying between f «C and fy, the impedance of the network at the input terminals being minimal, which is approximately a whole multiple of the reciprocal value of the duration of the Follow-up time corresponds to. 2. Flyback high voltage and sawtooth current generator after Claim 1, characterized in that the quality fit factor of the generator during the run-out time at the said frequency f / 3 more than 25 is. 309825/0761 47 - ΡΗΝ.6Ο59 3. Return high-voltage and SSgezahnstromgenerator nachAnspruch 1 or 2, wherein said network has a third, between f oL and fy lying'Resonanz frequency f £ during the return time, which almost corresponds to the expression for K equal to an odd integer, characterized in, that the first frequency f / j . lying between t oC and f Z , whereby the impedance of the network at the input terminals is minimal, almost corresponds to a whole multiple of the reciprocal value of the duration of the trace time. 4 · Reverse high voltage and SS gear current generator according to claim 3 »characterized in that the second frequency between ff and f y f / S _? i where the impedance of the network at the input cycle men is minimal, also almost corresponds to a whole multiple of the reciprocal value of the duration of the trace time. 309825/07 61 Blank page