JPS58178602A - Comb line type band-pass filter - Google Patents

Abstract

PURPOSE:To eliminate the need for fitting electrode to form electrostatic capacity in the inside of a housing by providing the titled comb line type band-pass filter with plural resonance elements consisting of rod-like conductors having axial length especially related to resonance wavelength. CONSTITUTION:In the comb line type band-pass filter, the axial length of the resonance elements 41-46 are selected at 1/4 resonance wavelength lambda and the terminal of the opposite side to the earth side of each resonance element is kept at the opened state or the nearly opened state. The adjacent resonance elements are mutually coupled through nagnetic fields generated around the respective resonance elements by resonance current flowing into the resonance elements 41-46. The band-pass filter with the prescribed transmission characteristics is constituted by setting up the magnetic field coupling factor between respective resonance elements to a prescribed value.

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a combline bandpass filter for microwaves. Hereinafter, the bandpass filter will be abbreviated as BPF.

FIG. 1 is a cross-sectional view showing an example of a conventional microwave combline type BPF, where 1 is a housing, 2 is an input/output coaxial terminal, and 3 is a sectional view showing an example of a conventional microwave combline type BPF.
is an input/output coupling rod, 4 is a resonant rod, and 5 is an electrode forming a load capacitor. In this BPF, the axial length of the resonant rod 4 is selected to be the resonant wavelength h, and the electrode 5 attached to the tip and the housing 1 The electrical length of the resonant rod 4 is made equal to the resonant wavelength of 4 due to the capacitance between the resonant rods, thereby enabling resonance and electric field coupling with adjacent resonant rods. However, when providing the required capacitance between the electrode 5 and the housing 1, the gap between the electrode 5 and the housing 1 becomes narrower, degrading the withstand voltage characteristics, and the electrode 5 and the housing become weaker due to changes in ambient temperature. Since a change in the size of the gap between the two leads to a large change in the capacitance between them, it is unsuitable for high-power BPFs for transmission, for example, because it has disadvantages such as not being able to obtain stable electrical characteristics. .

The present invention has a simple configuration, excellent withstand voltage characteristics, stable electrical characteristics without being affected by changes in surroundings or R degree, and is suitable for microwave combline type BPs suitable for high power applications, etc.
The purpose is to realize F.

FIG. 2 is a sectional view (B in FIG. 3) showing one embodiment of the present invention.
-B sectional view), Fig. 3 is an AA sectional view of Fig. 1, and in both figures, 1 is an electromagnetic shielding case, 21 and 22 are input/output coaxial terminals, and 31 and 32 are input/output couplings. An element, for example, a strip line.

Instead of coupling using a strip line, the input/output coupling may be performed by tap coupling, loop coupling, coupling using capacitance between a ribbon-shaped conductor and an iterator, or the like. 41 to 4 are resonant elements made of rod-shaped conductors, and the width of the housing 1 (H in FIG. 3) and the diameter of the resonant elements 41 to 4I are relatively small (for example, 0.005 mm) compared to the large resonant wavelength.
1 to 0.2 persons), the axial length of the resonant elements 41 to 4b is selected to be 4, which is the larger of the resonant wavelength, and the width H of the housing 1 and the diameter of the resonant elements 41 to 4b are compared to the resonant wave length. If it is relatively large, the axial length of the resonant elements 41 or 4 is appropriately shortened to the same Qt as the resonant wavelength, and the end of each resonant element opposite to the ground side is electrically opened or opened. It is kept close to.

In the comb-line type BPF' of the present invention configured as described above, adjacent resonant elements are magnetically coupled to each other by the magnetic field generated around each resonant element by the resonant current flowing through the resonant elements 41 to 4, and the magnetic field between each resonant element is By determining the magnetic field coupling coefficient as described below, it is possible to configure a BPI" having a maximal flank type transmission characteristic or a BPF' whose passband has a Chebyshev type transmission characteristic.

Magnetic field coupling coefficient between mutually adjacent resonant elements, that is, coupling coefficient M+ between resonant elements 41 and 42, 2% 4. ; Coupling coefficient MQ3 between 1 and 43; Coupling coefficient M5 between 4s and 4. B is Lh: the amount of magnetic field loss, and C: the distance between the center axes of mutually adjacent resonant elements. This is a curve diagram showing the relationship with the resonant wavelength (axis), where the curve 8 represents a case where H is relatively large compared to the resonance wavelength, and the curve 8 represents a case where H is relatively small compared to the resonance wavelength.

Next, the coupling coefficient M' between each adjacent resonant element expressed by the load Q or center circumference, the number of skins, the 3 dB lower frequency bandwidth of the transmitted signal, the geometric coefficient related to each resonant element, etc.
ni+I is M', ni,...=(Spring・k!te' Ma...
...(3)2 In the above formula, gx, gK and l are geometric coefficients, 2 to -1 g ==2Bln□π ......(4) n l B10 QL fe QL: Load Q B : 3 dB reduction in transmission signal frequency bandwidth fo: Center frequency coefficient related to resonant element 41 gK=g+t(4
) is obtained by substituting 1 for k in equation (4), and 2 is substituting for k in equation (4) to obtain the geometric coefficient g1 ya related to the resonant element 42, = g,
By determining QL or B, s3, and f, the resonant element 41 and 4. The coupling coefficient M' between +l:li! '+, 2 can be obtained, and in the same manner, 5 is substituted for k in equation (4) to obtain the geometric coefficient gK=g; related to the resonant element 45; By substituting the geometric coefficient L related to the four resonant elements
+++=gI, and QL or BW3 and fo
By setting the resonant elements 45 and 46 from equation (3),
It is possible to obtain the coupling coefficient M' between ++''M'!; and b.

Next, set the resonance wave key, determine the width H of the casing 1, and determine the center axis distance 01 between the resonant elements 41 and 42 (from this, from equation (1), the coupling coefficient M between the resonant elements 4I and 42 1 = "Ml-" can be obtained, and by similarly determining the central axis spacing C between the resonant elements 45 and 4&, the coupling coefficient M between the resonant elements 4s and 4& can be set to 2 from equation (1). ++ ” M5j can be obtained, and the size of the coupling coefficient M6. A maximally flat n-type BPF can be constructed by making the coupling coefficient M' of each stage correspond to the magnitude of ++, and its transmission characteristics are expressed by the following equation.

L (, iB) = lOmg, (1+x3″)...
...(5) Δf: Find the detuning frequency geometric coefficient of the transmission signal from the center frequency f from equations (6) and (7) below, and use the same method as above to make the passband Chebyshev shape. It is possible to configure a BPF whose characteristics and attenuation range are Wakuna-type characteristics.

However, the allowable voltage standing wave ratio in the S2 passband is π 13K = γ + 81n7 In this case, the geometric coefficient g1 related to the resonant element 4- is (
6) Substitute 1 for k in the equation, and find the geometric coefficient g associated with the 4 resonant elements by substituting 2 in the equation (7). Similarly, the related geometric coefficients g3 to gL are also expressed by kl of equation (7).
Find by substituting. Note that (6) is used only when calculating g+ according to equation 1.

Next, the transmission characteristics of a BPF with a passband having Chebyshev type characteristics and an attenuation area b (Wagner type) can be expressed as a 7-missing equation by giving the word voltage standing wave ratio S in the passband and the order n. I can come out.

...(8) Tn(X): Chebysyev's polynomial x＞
For I, TJx) = co8h'(n caBh-'x) At a normalized frequency of 3 dB below the transmission signal, (7
), it is necessary to hold that 3 decipe Jl of the transmission signal < the reduced normalized frequency x
3 can be found, so from By + 3 = Bwr aX3, find the 3 dB lower frequency width BIV3 of the transmission signal, find the load Q (QL) from , and use these values and (6)
By substituting the geometric coefficient g obtained from the equation and equation (7), gK, and I into equation (3), the coupling coefficient M' between each resonant element is obtained.
1, set the resonant wavelength size and the width H of the housing 1, and appropriately select the center axis spacing C of each resonant element (1
) By making K+1 coincide with M' in equation (3) to within 1, it is possible to construct a BPP whose passband has a Chebyshev type characteristic and whose attenuation pole has a Wagner type characteristic.

In this BPF, for example, by indirectly coupling two resonant elements and two openings, it is possible to construct a BPF having a Chebyshev type characteristic in the passband and an attenuation pole in the attenuation range.

6 shown with a dotted line in FIG. 3 is an indirect coupling #lIi circuit consisting of a coaxial line or a strip line, etc. 71 and 71 are coupling elements of the resonant elements 4 and 45 and the indirect coupling line 6, which is a loop or a resonant element. 42 and 4≦.

Among the signals transmitted through the main circuit of the resonant elements 41, t'L45, and Q, the signals whose frequencies are higher than the passband lead the phase of the voltage and current by 90 degrees in each resonant circuit consisting of the resonant elements 4I to 4&, and Since the phase advances by 270 degrees in the phase shift circuit formed between the coupling elements 71 and 4, the phase of the signal in the resonant circuit consisting of the resonant element 42 and the phase of the signal in the resonant circuit consisting of the resonant element 4 are in phase. When A is formed as a loop, if the polarities of both loops are set appropriately, the connection from the resonant circuit consisting of the resonant element 4- to the resonant element 45 via the indirect coupling circuit consisting of the coupling element 71, A and the line 6 is possible. The phase of the signal transmitted to the two resonant circuits produces a phase difference of 180 degrees between the two resonant circuits. Therefore, by adjusting the degree of coupling of the coupling elements 71 and 71 to equalize the magnitude of the signal transmitted through the indirect coupling circuit and the magnitude of the signal transmitted through the main circuit, an attenuation pole is generated at the frequency position of this signal. I can do it.

Among the signals transmitted through the main circuit, signals with frequencies lower than the passband have a phase delay of 90 degrees in each resonant circuit, and a phase delay of 27,070 degrees in the phase shift circuit formed between each resonant circuit, so the resonant element 4λ The phases of the signals in both the resonant circuits consisting of the resonant circuits 7I and 4s are in phase, and the phases of the signals transmitted through the indirect coupling circuit are opposite. By making the magnitudes of both signals transmitted through the coupling circuit equal, an attenuation pole can be produced at the frequency position of this signal.

The frequency position of the seven attenuation poles that determine the degree of coupling between the coupling elements 7I and 7 moves away from the passband, and conversely, when the degree of coupling is increased, the frequency position of the attenuation pole moves closer to the passband.

The transmission characteristics of the polarized BPF configured in this way are expressed by the following equation.

When the order n is 6 as in this example, that is, when n is an even number, when n is an odd number, Frequency difference Rc: means to take the real part ■, : means to take the imaginary part If the frequency position of the attenuation pole is relatively far from the passband, the passband has Chebyshev type characteristics, the attenuation range has a coupling characteristic that is almost equal to BPF', which has a Wagner-type characteristic, and if the attenuation pole is located at a frequency position relatively close to the passband, there will be a difference between the theoretical value and the experimental value due to the influence of the indirect coupling loop or capacitance. This generally occurs, and appropriate correction is required to obtain the above-mentioned ff1i, ρ3, etc.

FIG. 5 is a sectional view (BB sectional view in FIG. 6) showing another embodiment of the present invention, FIG. 6 is a sectional view taken along AA in FIG. 5, and FIG. In each figure, 8 is a partition wall made of a conductor, and other symbols are the same as in FIGS. 2 and 3.

In this embodiment as well, the width of the housing 1 (H in FIG. 6) and the resonant elements 41 to 41. If the diameter of the resonant elements 4- to 4b is relatively small, the axial length of the resonant elements 4- to 4b is selected to be 4 to the resonant wavelength, and the width H of each unit 1 and the diameter of the resonant element group to 4- If the wavelength is relatively large, the axial length of the resonant elements 4I to 4b is appropriately made shorter than the resonant wavelength 4, and the end of each resonant element opposite to the ground side is electrically open or nearly open. The configuration is such that adjacent resonant elements can be magnetically coupled to each other in a cascade connection.The geometric coefficient is determined from equation (4) or equations (6) and (7), and the resonant wavelength is determined by Determine the width of the body 1 (H in Figure 6), adjust the center axis spacing of the resonant elements 4I to 4b, and add +1 to the coupling coefficient MK in equation (1) to M' in equation (3). +1, it is possible to construct a BPF having the same transmission characteristics as in the previous embodiment, but in this embodiment, the resonant cables 41 to 4b are arranged in a U-shape, so that the transmission characteristics are different from those in the previous embodiment. The entire structure can be made compact.

In this embodiment, for example, as shown in the cross-sectional view in FIG. As shown in the cross-sectional view in the figure, by indirectly coupling between the resonant elements 42 and 45 via the loop IO, it is possible to prevent the generation of five-attenuation poles. Therefore, there is no need to use a coaxial line or the like as in the previous embodiment, and indirect coupling can be performed by a coupling element inserted in the hole of the partition wall 8, so that the structure can be made simple from this point as well.

In each of the above embodiments, six resonant elements are used. However, the number of resonant elements is not limited to six, and the present invention can be implemented by increasing or decreasing the number of resonant elements as appropriate. (6, O and 7 in Figure 3) and indirect coupling elements (9 in Figure 8 and 10 in Figure 9) are also resonant elements 42 and 4s.
Instead of providing it between the resonant cables 41 and 4b, or 4
It may also be provided between C and 45 and between 4I and 46 (in short, if it is between two resonant elements or an integral multiple of the resonant elements in a cascade connection relationship) The present invention can be implemented by appropriately selecting the location and number of indirect coupling circuits or elements.

As is clear from the above description, in the BPF of the present invention, the axial length of each resonant element is selected to be close to the resonant wavelength Z, and the end opposite to the ground side is electrically opened to connect adjacent resonant elements to each other. Since it is configured to create magnetic field coupling, there is no need to attach an electrode to the tip of the resonant element to form a capacitance between it and the housing as in the conventional case. It is possible to provide a sufficiently large gap between the
Therefore, the withstand voltage between the tip of the resonant element and the casing can be sufficiently increased, and stable electrical characteristics can be obtained regardless of changes in ambient temperature, making it suitable, for example, as a BPF for high power use.

[Brief explanation of the drawing]

FIG. 1 is a sectional view showing a conventional combline type bandpass filter, FIGS. 2 and 3 are sectional views showing an embodiment of the present invention, and FIG. 4 is a sectional view showing the filter of the present invention. Figures 5 to 9 are cross-sectional views showing other embodiments of the present invention; 3: Input/output coupling rod, 31 and 3
2: Input/output coupling element, 4: Resonant rod, 41 to 4F:
Resonant element, 5: electrode, 6: indirectly coupled line, 7I, 7a,
9 and 10: indirect coupling element; 8: partition wall. Figure 1 Figure 2 Figure 3 Figure 4 Figure 5 Figure 1 Figure 6 Figure 7 Figure 8 Figure 9

Claims (1)

(Claims: (+)) A combline type bandpass filter comprising a plurality of resonant elements each made of a rod-shaped conductor having an axial length of approximately 4 times the resonant wavelength. (2) a plurality of resonant elements made of rod-shaped conductors having an axial length of 1/4 times the resonant wavelength, and among the plurality of resonant elements,
A combline type bandpass filter comprising a circuit or an element that indirectly couples two resonant elements separated by two or an integer multiple of the resonant elements. (3) A combline type bandpass filter according to claim 1 or 2, in which a plurality of resonant elements are arranged in a line. (4) A combline type bandpass filter according to claim 1 or 2, wherein a plurality of resonant elements are arranged to form a fold-back signal transmission path.