JPH1172515A - Broad-band analog insulation circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To obtain a floated broad-band signal on the non-floating side. SOLUTION: Multiplication circuits 20 and 60 are arranged both on the left floating side and on the right non-floating side of insulation transformers 11 and 12 and a high frequency sinusoidal signal is supplied from an oscillator 18. The multiplication 20 performs a multiplication between an input signal applied to input terminals 1a and 1b and a high frequency sinusoidal signal 18 applied to terminals 3a and 3b. The results are multiplied by the high frequency sinusoidal signal 18 at the multiplication circuit 60 to obtain an input signal restored to output terminals 2a and 2b through low pass filters 67 and 68. Light or a radio wave can be used in place of the insulation transformers 11 and 12. Two sets of circuits as illustrated are used and the phase difference of the high frequency sinusoidal signal 18 between the two systems is set to 90 deg. to add outputs of the two multiplication circuits 60. This can eliminate low pass filters 67 and 68.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a wideband analog isolation circuit. Specifically, in the case of a floating measurement for measuring a potential difference between two points where neither point is the ground potential, a range from DC to several hundred megahertz (MHz) is used to insulate a grounded voltmeter from a measuring object. It is an object of the present invention to provide an improved isolation circuit for isolating a wideband analog signal.

[0002]

2. Description of the Related Art Floating measurement for measuring a potential difference between two points which are not grounded is often required in the fields of motors, power supplies and the like. However, the floating measurement point is often at a high potential, and it is not easy to measure the potential difference between two points of the floating measurement object safely and accurately. In particular, when a minute differential signal is superimposed on a large voltage amplitude having the same phase, or when the signal to be measured has a high frequency, accurate measurement has been difficult.

[0003] Such a problem often occurs at the time of measurement with an oscilloscope, but conventionally, the oscilloscope is measured in a so-called floating state without grounding.
However, when the oscilloscope is not grounded, the following problem occurs.

[0004] The first problem is that there is a possibility that the outer metal part of the measuring instrument will have the same potential as the object to be measured, resulting in electric shock.

A second problem is that the measured signal waveform is linked due to the ground capacitance of the oscilloscope, or at the moment when the measured point is touched by the probe, the measured point which is floating in a direct current and / or high frequency is changed. As a result of being momentarily grounded through the ground capacitance, the circuit under test on the floating side may be damaged.

In order to prevent such inconvenience, it is necessary to use an oscilloscope grounded, and to electrically insulate the oscilloscope from the measurement point on the floating side with an analog insulation circuit. Conventional analog insulation circuits generally use a synchronous detection circuit that multiplies an analog signal by a square wave clock, modulates the signal, multiplies the signal again by the square wave clock, and demodulates the signal.

FIGS. 13 and 14 show an example of a conventional analog insulation circuit of this kind and waveforms of respective parts thereof. The input voltage of FIG. 14A applied to the input terminal 1 of FIG. 13 is input to the modulator 202 via the input buffer 201. The square wave clock of (b) from the terminal 19 that obtains the output of the oscillator 18B passes through the insulating transformer 12 and becomes the modulation clock of the modulator 202. The modulator 202 multiplies the input signal of (a) by the modulation clock of (b) and outputs the result.

[0008] The output of the modulator 202 shown in (c) is input to the demodulator 203 through the insulating transformer 11. The square wave clock output from the oscillator 18B is supplied to the demodulator 203
Becomes the demodulation clock. The demodulator 203 multiplies the output of the modulator 202 shown in (c) by the demodulated clock shown in (b) and outputs the result.

The output of the demodulator 203 shown in FIG.
Since it includes a spike-like noise component of the frequency component of the square-wave clock of (b), it is input to the low-pass filter 204. The output of the low-pass filter 204 passes through the output buffer 205 to obtain the output signal of (e) at the output terminal 2. The floating modulation side and the non-floating demodulation side, which is often grounded to the ground, are insulated by the isolation transformers 11 and 12, so that analog signals can be isolated. Note that the use of two types of GND (ground) symbols on the floating side and the non-floating side in FIG. 13 means that each GND is insulated.

In the analog isolation circuit based on synchronous detection using the square wave clock as described above, since the multiplication with the square wave clock at the time of modulation and demodulation is performed, the modulation clock and the demodulation clock have sag and overshoot. Must not be a square wave.

Therefore, it is required that the square wave clock generated by the oscillator 18B does not include sag and overshoot, and that the square wave characteristic is not deteriorated by the insulating transformer 12. Since this type of oscillator 18B and insulating transformer 12 operate only at a few hundred kHz at most, the analog signal bandwidth that can be handled is about 100 kHz, which is sufficiently lower than the repetition frequency of the square wave clock, and a high-speed waveform is generated. It was impossible to measure.

As another analog isolation circuit, there is one using a photocoupler. However, the band in which the analog signal can be accurately isolated by the photocoupler is at most 1 MHz, which is insufficient for high-speed waveform observation.
A broadband analog isolation circuit using a photocoupler and an isolation transformer has been proposed in the following document.

Reference 1. US Patent 5,517,15
4 “SPLITPATH LINEAR ISOLATIONCIRCUIT APPARATUS
AND METHOD "

The technique disclosed in Document 1 is DC-10
Insulate the low band of about 0kHz with a photo coupler,
A high frequency band of about kHz to 100 MHz is insulated by an insulating transformer and crossover is performed so that the entire frequency characteristic becomes flat, thereby isolating analog signals of DC to about 100 MHz.

[0015]

The conventional analog isolation circuit using synchronous detection or a photocoupler can only provide a frequency bandwidth of about 100 kHz to about 1 MHz, and therefore cannot cope with higher-speed signal observation. was there.

The analog isolation circuit disclosed in Document 1 still has the following problem.

The first problem is that since the bandwidth of the photocoupler is at most 1 MHz, the crossover frequency can be increased to only about 100 kHz at most, and the low cutoff frequency of the insulating transformer cannot be increased. Therefore, there is a problem that the insulating transformer is not downsized, and the insulating transformer that can be used up to low frequencies becomes large, so that generally the high frequency characteristics are not good. Therefore, it is difficult to further increase the frequency bandwidth.

The second problem is that low-frequency and high-frequency bands are separately insulated and cross-over, so that cross-over distortion occurs and accurate adjustment of low-frequency and high-frequency gains is required. It is necessary.

The present invention has been made in view of such an unsolved problem, and does not use an insulating transformer having a low cut-off frequency, which is an obstacle to widening the band, and uses a waveform from DC to high frequency. An object of the present invention is to provide a wideband analog isolation circuit capable of isolation without distortion.

[0020]

A floating-side multiplication circuit for multiplying a floating-side analog input signal and a floating-side high-frequency sine-wave signal, an output signal of the floating-side multiplication circuit and a non-floating-side high-frequency sine-wave signal , A low-pass filter for removing the high-frequency component from the output of the multiplier and restoring the analog input signal, and isolating the floating side from the non-floating side, A multiplying circuit having the output of the multiplying circuit as an input to the non-floating side multiplying circuit; an insulating means for a multiplying signal; an oscillator for oscillating a high-frequency sine-wave signal; High frequency sine wave signal and Floating side while insulating against non floating side, provided with high-frequency sine-wave insulating means for applying to the multiplying circuit of the floating-side and non-floating side Te.

With this configuration, the conventional synchronous detection type analog isolation circuit performs balanced modulation and demodulation with a square wave, whereas the analog isolation circuit of the present invention uses a high frequency sine wave signal on the floating side and the non-floating side. By performing multiplication, DC to 100 MHz (MH)
z) It is possible to insulate the floating side from the non-floating side while transmitting the analog signal up to the higher frequency. Also, since the insulating means only needs to pass a high-frequency signal, waveform distortion does not occur. further,
Since the insulating transformer transmits a high-frequency signal, there is no need to use a large core material, and the insulating transformer can be made compact.

[0022]

FIG. 1 shows an embodiment of the present invention. The same components as those in FIG. 13 are denoted by the same reference numerals. The use of two types of GND symbols and power supply symbols means that the floating side and the non-floating side are insulated, as in the case of FIG.

In FIG. 1, the differential input terminals 1a and 1b on the floating side are connected to differential inputs of a multiplying circuit 20, respectively. Terminals 3a and 3b of multiplication circuit 20
Is connected to the differential output of the secondary winding of the conversion transformer 16 for converting an unbalanced input biased to the floating-side voltage V _B1 to a balanced output at the middle point, and supplied from the primary winding side. A high frequency sine wave is applied. This high-frequency sine wave is oscillated by the oscillator 18 on the non-floating side, and is supplied via the insulating transformer 12.

The analog signal applied to the terminals 1a and 1b and the high-frequency sine wave signal applied to the terminals 3a and 3b are multiplied in a multiplying circuit 20, and the result of the multiplication is obtained at the terminals 4a and 4b. The result of this multiplication is an insulating transformer 11 that insulates the floating side from the non-floating side via two emitter followers constituted by transistors 21 and 22 and constant current sources 23 and 24 between the floating side power supply V _CC and GND. Is applied to the primary side.

The multiplication circuit 60 on the non-floating side has the same configuration as the multiplication circuit 20 on the floating side.
The input terminals 6a and 6b are connected to two terminals of the isolation transformer 11 whose middle point is biased to the non-floating side voltage _VB2.
The next winding is connected and the result of the multiplication obtained on the floating side is applied.

The terminals 8a and 8b of the multiplying circuit 60 are connected to the differential output of the secondary winding of the conversion transformer 17 for converting the middle point to an unbalanced output biased to the non-floating side voltage V _B3 . A high-frequency sine wave supplied from the primary winding side is applied from a terminal 19 for obtaining an output of the oscillator 18.

The result of the multiplication on the floating side applied to the terminals 6a and 6b and the high-frequency sine wave signal applied to the terminals 8a and 8b are multiplied in a multiplication circuit 60, and the result of the multiplication is obtained at the terminals 9a and 9b. Is obtained. The result of this multiplication is obtained by the transistors 61 and 62 and the constant current sources 63 and 6 between the non-floating side power supply V _CC and GND.
The high-frequency components are removed by low-pass filters (LPFs) 67 and 68 through two emitter followers constituted by 4 and input terminals 1a and 1 are connected to output terminals 2a and 2b.
b reproduces the input signal applied to b.

FIG. 2 shows waveforms at various points in the circuit of FIG. 2A shows the differential input voltage waveform between the differential input terminals 1a and 1b, and FIG. 2B shows the output waveform of the oscillator 18, and the high-frequency sine wave signal at the output terminal 19 of the oscillator 18 is The signal is input to the balanced / unbalanced conversion transformer 16 through the insulating transformer 12. Balance / unbalance conversion transformer 16
Supplies a high frequency sine wave signal with inverted polarity to the terminals 3a and 3b, respectively. The multiplication circuit 20 calculates the multiplication result of the differential input signal and the high frequency sine wave signal as shown in FIG.
a and 4b, and the output has a waveform that is balanced-modulated with a high-frequency sine wave signal. Output signal is transistor 2
It is input to the primary winding side of the insulating transformer 11 through an emitter follower composed of the constant current sources 23 and 24 and the constant current sources 23 and 24.

The differential input signal applied between terminals 1a and 1b is E _A , the high-frequency sine wave signal at terminal 19 is E _B ,
Assuming that the result of the multiplication on the floating side between the terminals 4a and 4b is E _C , the following relationship exists.

[0030] _{E A = f (t) (} 1) E B = sin (ωt) (2) E C = E A × E B = f (t) sin (ωt) (3)

The signal E _C of the resulting expression in the secondary winding of the isolation transformer 11 (3) of the terminal 6a, are input to the multiplier circuit 60 through 6b. On the other hand, the high-frequency sine wave signal of the formula (2) output from the oscillator 18 is input to the balanced / unbalanced conversion transformer 17, and the conversion transformer 17 is connected to the terminals 8a and 8b.
And supplies the high-frequency sine wave signal of which the polarity is inverted to the multiplication circuit 60 via the. Similarly to the multiplication circuit 20, the multiplication circuit 60 also outputs the differential voltage between the terminals 6a and 6b and the terminal 8a
The result of multiplication of the high frequency sine wave signal between
2 and 9b (FIG. 2 (d)).

The output signal E _D = E _C × E _B of the multiplication circuit 60 obtained between the terminals 9a and 9b is expressed as follows. E _D = f (t) sin ² (ωt) = (1/2) f (t) (1−cos (2ωt)) (4)

When the frequency component of the input signal f (t) is lower than the frequency of the high-frequency sine wave signal sin (ωt), the cut-off frequency of the low-pass filters 67 and 68 is set lower than 2ωt. ) Is 0, and only the term of (1/2) f (t) remains. This is obtained between the output terminals 2a and 2b, and the input signal is reproduced (FIG. 2). (E)). The input terminals 1a, 1b and the output terminals 2a,
Although 2b is completely insulated by the insulating transformer 11, analog signal isolation in a wide band from DC to high frequency is possible.

FIG. 3 is a circuit diagram showing a specific example of the oscillator 18 which is a component of FIG. The output of the voltage controlled oscillator 94 is input to a prescaler 95, and the prescaler 95
Divides the frequency to a frequency at which the phase difference detector 91 can operate.
The phase of the divided signal and the signal of the reference oscillator 90 are compared by a phase difference detector 91, and an analog signal is output to a charge pump 92 in accordance with the phase difference. Charge pump 9
The output signal of No. 2 passes through the loop filter 93 and is input to the frequency control terminal of the voltage controlled oscillator 94.

This loop is called a phase locked loop (PLL circuit), and the frequency of the voltage controlled oscillator 94 is locked to the frequency accuracy of the reference oscillator 90. The output of the voltage controlled oscillator 94 is buffered by a buffer 96, and an unnecessary frequency component is cut by a filter 97 as necessary, and output to a terminal 19. In such a PLL circuit, the frequency at the terminal 19 is 1 GHz to 2 GHz according to the recent technology.
It can be easily realized up to about GHz. Also, A
When a high-speed oscillator is used as a sampling clock, as in the case of a digital storage oscilloscope equipped with a D converter, this may be used.

When the oscillator 18 of FIG. 1 outputs a sine wave of about 1 GHz, the conversion transformers 16 and 17 for balance and unbalance or the insulation transformers 11 and 12 can be made compact. For example, the conversion transformers 16 and 17 for balance / unbalance can use high-frequency baluns for wireless use, but very small surface mount components are commercially available. In the case of the insulating transformers 11 and 12, a certain gap is required between the primary winding and the secondary winding in order to secure the dielectric strength, but a wire is wound around a small toroidal core. It can be easily configured.

FIG. 4 shows a specific circuit example of the multiplication circuit 20 on the floating side in FIG. This is known as a Gilbert type multiplication circuit, and can be easily integrated into an IC.

In FIG. 4, the differential input terminals 1a and 1b are respectively connected to the bases of transistors 31 and 32 of the differential current amplifying stage, and a negative feedback resistor is provided between the emitters of the transistors 31 and 32. 41 and 42 are connected. The midpoint of the resistors 41 and 42, constant current source connected to the negative power supply V _EE floating side is connected.

This constant current source is composed of transistors 37, 38 and resistors 45, 46, 47. Resistance 45, 4
6, 47 determine the current value of the current mirror. The transistors 37 and 38 form a current mirror. Cascode-connected transistors 33, 34, 3
Since the elements 5 and 36 are required to have the same characteristics, it is desirable that the Gilbert type multiplication circuit 20 be realized by a monolithic IC.

The emitter of the transistors 33 and 35 is cascode-connected to the collector of the transistor 31, and the transistor 3 is connected to the collector of the transistor 32.
4,36 emitters are cascode-connected. The bases of the transistors 33 and 35 are each a transistor 3
It is connected to 4,36 bases.

The collectors of the transistors 33 and 34 are connected to the collectors of the transistors 36 and 35, respectively.
Is connected to the collectors of the transistors 33 and 36. A load resistor 44 is connected to the collectors of the transistors 33 and 36. The middle point of the load resistors 43 and 44 is connected to the floating-side positive power supply V _CC . The transistors 31 to 38 constitute a Gilbert type multiplication circuit 20 on the floating side. The Gilbert-type multiplication circuit 20 outputs the result (product) of multiplication of the signal voltage between the terminals 1a and 1b and the high-frequency sine wave voltage between the terminals 3a and 3b.

The collectors of the transistors 34 and 35 and the collectors of the transistors 33 and 36 are connected to terminal 4 respectively.
a, 4b are connected to the bases of the transistors 21, 22 (FIG. 1). The collectors of the transistors 21 and 22 are respectively connected to the floating-side positive power supply V _CC, and the emitters of the transistors 21 and 22 are connected to the constant current sources 23 and 24, respectively, to form emitter followers. The emitters of the transistors 21 and 22 are further connected to the primary winding of the insulating transformer 11.

FIG. 5 shows a specific circuit example of the multiplying circuit 60 on the non-floating side in FIG. This is known as a Gilbert type multiplication circuit with the same configuration as in FIG. 4 and can be easily integrated into an IC.

In FIG. 5, the differential input terminals 6a and 6b are respectively connected to the bases of transistors 71 and 72 of the differential current amplifying stage, and a negative feedback resistor is provided between the emitters of the transistors 71 and 72. 81 and 82 are connected. The midpoint of the resistors 81 and 82, constant current source connected to the negative power supply V _EE floating side is connected.

This constant current source is composed of transistors 77 and 78 and resistors 85, 86 and 87. Resistance 85, 8
6, 87 determine the current value of the current mirror. The transistors 77 and 78 form a current mirror. Cascode-connected transistors 73, 74, 7
Since the elements 5 and 76 are required to have the same characteristics, it is desirable that the Gilbert type multiplication circuit 60 be realized by a monolithic IC, and the same multiplication circuit 20 as an IC can be used.

The emitter of transistors 73 and 75 is cascode-connected to the collector of transistor 71, and the collector of transistor 72 is connected to the collector of transistor 72.
4,76 emitters are cascode-connected. The bases of the transistors 73 and 75 are respectively the transistor 7
It is connected to 4,76 bases.

The collectors of the transistors 73 and 74 are connected to the collectors of the transistors 76 and 75, respectively.
The collectors of the transistors 73 and 76 are connected to a load resistor 84, and the middle point between the load resistors 83 and 84 is connected to the floating-side positive power supply V _CC . The transistors 71 to 78 constitute a Gilbert-type multiplication circuit 60 on the floating side. The Gilbert-type multiplication circuit 60 outputs the result (product) of multiplication of the signal voltage between the terminals 6a and 6b and the high-frequency sine wave voltage between the terminals 8a and 8b.

The collectors of transistors 74 and 75 and the collectors of transistors 73 and 76 are connected to terminal 9 respectively.
a, 9b are connected to the bases of the transistors 61, 62 (FIG. 1). The collectors of the transistors 61 and 62 are connected to the floating-side positive power supply V _CC, and the emitters of the transistors 61 and 62 are connected to constant current sources 63 and 64, respectively, to form emitter followers. The emitters of transistors 61 and 62 are connected to output terminal 2 via low-pass filters 67 and 68, respectively.
a, 2b.

FIG. 6 shows that the input signal in FIG. 1 is a step waveform, the frequency of the oscillator 18 is 1 GHz, and the cutoff frequency of the low-pass filters (LPF) 67 and 68 is 100 MHz.
It is an example of the simulation in the case of. 5 on the time axis
A scale of ns interval is provided.

(A) is a differential input waveform between the input terminals 1a and 1b of FIG. 1, (b) is a waveform between the terminals 4a and 4b for obtaining the output of the multiplication circuit 20 of FIG. 1, and (c) is a multiplication. Circuit 60
(D) shows a differential output waveform between the output terminals 2a and 2b. As can be seen from FIG. 6, the floating side (the insulating transformer 11 of FIG. 1)
It can be seen that an analog signal can be obtained from DC to high frequency without distortion while insulating the non-floating side (the left side of FIGS. 1 and 12) from the non-floating side (the right side of the insulating transformers 11 and 12 in FIG. 1).

[0051]

Embodiment 1 FIG. 7 shows a circuit diagram of Embodiment 1 of the present invention. 7, the same components as those in FIG. 1 are denoted by the same reference numerals. In the configuration of FIG. 7, a second system including a multiplication circuit 20B and a multiplication circuit 60B is added to the configuration of FIG. 1 in addition to the first system.

The multiplication circuit 20B is the same as the multiplication circuit 20 and is shown in FIG. However, the terminals 3a and 3b in FIG. 4 (or FIG. 1) are replaced with 3c and 3d, and the terminals 4a and
4b is replaced with 4c and 4d. The multiplication circuit 60B is the same as the multiplication circuit 60 and is shown in FIG.
However, the terminals 6a and 6b in FIG. 5 (or FIG.
6d, the terminals 9a, 9b are replaced with 9c, 9d. The high-frequency sine wave signal output from the oscillator 18 is phase-shifted by 90 degrees by the phase shifter 65, and then input to the multiplication circuit 60B as a high-frequency cosine wave signal via a balance / unbalance conversion transformer 17B. The high-frequency sine wave signal output from the oscillator 18 passes through the insulating transformer 12, is shifted in phase by 90 degrees by the phase shifter 25, and is converted as a high-frequency cosine wave signal through the balance / unbalance conversion transformer 16B. The signal is input to the multiplication circuit 20B.

The inputs of the multiplying circuit 20B are input terminals 1a,
1b, the multiplying circuit 20B has a differential input signal,
The result of multiplication with the high frequency cosine wave signal obtained by shifting the high frequency sine wave signal by 90 degrees, that is, the product is output.
The output signal of the multiplying circuit 20B includes an emitter follower including a transistor 21B and a constant current source 23B connected to a terminal 4c, and a transistor 22 connected to a terminal 4d.
B, passes through the emitter follower composed of the constant current source 24B, and is input to the multiplication circuit 60B via the insulating transformer 11B. The multiplying circuit 60B multiplies the output signal of the multiplying circuit 20B by the high-frequency cosine wave signal obtained by shifting the high-frequency sine wave signal by 90 degrees, that is, outputs the product to the terminals 9c and 9d.

By connecting the terminals 9c and 9d for obtaining the output of the second system to the terminals 9a and 9b for obtaining the output of the first system, the output signals of the multiplication circuits 60 and 60B are added. The added output signal is output from the transistor 61
And constant current source 63 and transistor 62 and constant current source 64
Through two emitter followers consisting of
a and 2b.

FIG. 8 is a waveform diagram obtained by performing simulations on the waveforms of the respective parts of the circuit of FIG. 7 under the same conditions as in FIG. 6, that is, with the frequency of the oscillator 18 being 1 GHz. The time axis is marked at intervals of 5 ns. The voltage axis is an arbitrary scale.

(A) is a differential input waveform between the terminals 1a and 1b, and (b) is the terminals 4a and 4 for obtaining the output of the multiplication circuit 20.
b, the voltage waveform between the terminals 4c and 4d for obtaining the output of the multiplying circuit 20B, and the voltage waveform between the terminals 9a and 9b for obtaining the output of the multiplying circuit 60. A single output waveform of the first multiplication circuit 60 when the output of the second multiplication circuit 60B is set to zero),
(E) Terminals 9c and 9d for obtaining the output of the multiplication circuit 60B
(However, a single output waveform of the second system multiplication circuit 60B when the output of the first system multiplication circuit 60 is set to zero), (f) is (d) and (e) Are the differential output waveforms of the output terminals 2a and 2b obtained by adding the above waveforms.

[0057] Voltage waveform from (a) (f) (E a from E _f) is expressed as follows using equations. _{E a = f (t) (} 5) E b = E a × sin (ωt) = f (t) sin (ωt) (6) E c = E a × cos (ωt) = f (t) cos (ωt (7) E _d = E _b × sin (ωt) = f (t) sin ² (ωt) (8) E _e = E _c × cos (ωt) = f (t) cos ² (ωt) (9) E _f = f (t) × (sin ² (ωt) + cos ² (ωt)) = f (t) (10)

Equation (10) reproduces the input waveform of equation (5) and does not include high-frequency components, so that low-pass filters 67 and 68 are unnecessary. In a real circuit, due to the finite frequency bandwidth and the nonlinearity of the circuit,
The waveform (f) contains a slight noise component, but when the noise component is a problem, the low-pass noise
By adding the filters 67 and 68, it is possible to remove the noise component and extract only the desired signal component f (t).
In this case, the low-pass filters 67 and 68 used in FIG.
, A low-pass filter having a gentler cutoff characteristic having less influence on the signal component f (t) can be used.

[0059]

Second Embodiment FIG. 9 is a circuit diagram of a main part of a second embodiment of the present invention. Components corresponding to those in FIG. 1 are denoted by the same reference numerals, and therefore, differences from FIG. 1 will be described. In FIG. 9, the multiplication circuit 20 and the multiplication circuit 60 are connected by capacitors 101 and 102. The capacitor 103 is connected between the oscillator 18 and the conversion transformer 16 for balance / unbalance. Further, the capacitances of the capacitors 101, 102, and 103 are set as small as possible within a range where the signal of the frequency of the oscillator 18 passes without being attenuated. In this case, the capacitors 101, 102, 10
It will be apparent that isolation is provided for signals below the reduced cutoff frequency of 3. It is also clear that this circuit can be applied to the first embodiment shown in FIG.

[0060]

Third Embodiment FIG. 10 is a circuit diagram showing a main part of a third embodiment of the present invention. Components corresponding to those in FIG. 1 are denoted by the same reference numerals, and therefore, differences from FIG. 1 will be described. In FIG. 10, the multiplication circuit 20 and the multiplication circuit 60
Are connected by two pairs of antennas 121 and 124 and 122 and 125. The oscillator 18 and the conversion transformer 16 for balance / unbalance are connected by a pair of antennas 126 and 123. It will be apparent that such a configuration increases the insulation distance, thereby improving the dielectric strength. It is also clear that this circuit can be applied to the first embodiment shown in FIG.

[0061]

Fourth Embodiment FIG. 11 is a circuit diagram showing a main part of a fourth embodiment of the present invention. Components corresponding to those in FIG. 1 are denoted by the same reference numerals, and therefore, differences from FIG. 1 will be described. In FIG. 11, the output of the multiplying circuit 20 is sent to light transmitting drivers 131 and 132 via transistors 21 and 22 forming emitter followers, and the light emitting elements 136 and 1 are output.
From 37, the light signal is sent to the light receiving elements 146 and 147 by the optical fibers 156 and 157, respectively. The signals received by the light receiving elements 146 and 147 are received by the light receiving amplifier 14.
1, 142, and amplified at the terminals 6a,
6b.

On the other hand, the high-frequency sine wave signal output from the oscillator 18 is applied to the light transmission driver 134 and the light emitting element 139
From the light receiving element 1 through the optical fiber 159
It is sent to 49. The signal received by the light receiving element 149 is amplified by the light receiving amplifier 144 and applied to the balance / unbalance conversion transformer 16.

[0063]

Fifth Embodiment FIG. 12 is a circuit diagram of a main part of a fifth embodiment of the present invention. Components corresponding to those in FIG. 1 are denoted by the same reference numerals, and therefore, differences from FIG. 1 will be described. In FIG. 12, the output of the multiplying circuit 20 is sent to light transmitting drivers 131 and 132 via transistors 21 and 22 forming emitter followers, and the light emitting elements 136 and 1 are output.
From 37, optical signals are sent to the light receiving elements 146 and 147 by spatial transmission, respectively. Light receiving element 146,1
The signal received at 47 is amplified by the light receiving amplifiers 141 and 142 and applied to the terminals 6a and 6b of the multiplication circuit 60.

On the other hand, the high frequency sine wave signal output from the oscillator 18 is applied to the light transmission driver 134 and
, And sent to the light receiving element 149 by spatial transmission. The signal received by the light receiving element 149 is
4 and is applied to a balance / unbalance conversion transformer 16.

[0065]

As described above, according to the present invention, a simple circuit configuration based on a Gilbert-type multiplication circuit suitable for making a monolithic IC can be used to convert a floating side direct current to 100 megahertz (MHz) or more. It has become possible to restore analog signals over a wide band. Further, the circuit of the present invention can be constituted by a small transformer without using a large core for low frequency, which has been an obstacle to miniaturization and broadening of the band, and thus has an advantage that it is suitable for miniaturization and broadening of the band. In addition, there is an advantage that there is no occurrence of crossover distortion, which is a problem in the conventional method (Reference 1) in which the low frequency region and the high frequency region are insulated by separate elements and the respective frequency characteristics are crossed over. Further, for example, 1 GHz to 2 G
If modulation is performed at a radio frequency of Hz, insulation at antenna intervals can be achieved, and analog signal insulation with a very high withstand voltage over a wide band can be realized by using an optical signal. Therefore, the effect of the present invention is extremely large.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing an embodiment of the present invention.

FIG. 2 is a waveform chart showing waveforms of respective parts for illustrating the principle of FIG. 1;

FIG. 3 is a circuit configuration diagram of an oscillator that is one of the components of FIG.

FIG. 4 is a circuit diagram in a case where a multiplication circuit on the floating side, which is one of the components in FIG. 1, is implemented by a Gilbert type multiplication circuit.

FIG. 5 is a circuit diagram in the case where a non-floating side multiplication circuit, which is one of the components in FIG. 1, is implemented by a Gilbert type multiplication circuit.

FIG. 6 is a waveform chart obtained by simulating the waveform of each unit with respect to the step input of FIG. 1;

FIG. 7 is a circuit diagram showing Embodiment 1 of the present invention.

FIG. 8 is a waveform chart in which waveforms of respective parts with respect to the step input of FIG. 7 are obtained by simulation.

FIG. 9 is a circuit diagram showing a main part of a second embodiment of the present invention.

FIG. 10 is a circuit diagram showing a main part of a third embodiment of the present invention.

FIG. 11 is a circuit diagram showing a main part of a fourth embodiment of the present invention.

FIG. 12 is a circuit diagram showing a main part of a fifth embodiment of the present invention.

FIG. 13 is a circuit diagram showing a conventional example.

FIG. 14 is a waveform chart showing waveforms at various parts in FIG.

[Explanation of symbols]

1, 1a, 1b input terminal 2, 2a, 2b output terminal 3a-d, 4a-d, 6a-d, 8a-d, 9a-d
Terminals 11, 12 Insulation transformer 18, 18B Oscillator 16, 17 Conversion transformer 19 Terminal 20, Multiplication circuit 21, 22 Transistor 23, 24 Constant current source 25 Phase shifter 31-38 Transistor 41-47 Resistance 60 Multiplication circuit 61, 62 Transistor 63 , 64 constant current source 65 phase shifter 67, 68 low pass filter (LPF) 71-78 transistor 81-87 resistor 90 reference oscillator 91 phase difference detector 92 charge pump 93 loop filter 94 voltage controlled oscillator 95 prescaler 96 buffer 97 Filter 101-103 Capacitor 111,112 Resistance 121-126 Antenna 131,132,134 Light transmission driver 136,137,139 Light emitting element 141,142,144 Light receiving amplifier 146,147,149 Light receiving Element 156, 157, 159 Optical fiber 201 Input buffer 202 Modulator 203 Demodulator 204 Low-pass filter 205 Output buffer

Claims (14)

[Claims] A floating-side multiplying means for multiplying a floating-side analog input signal by a floating-side high-frequency sine wave signal to obtain a floating-side multiplication result; Signal isolation means (11, 101, 102, 121, 122, 124, 1) for isolating the result of the multiplication on the floating side as an input signal on the non-floating side while insulating them from each other.
25, 131, 136, 156, 146, 141, 13
2, 137, 157, 147, 142) multiplying the non-floating side input signal by the non-floating side high-frequency sine wave signal to obtain a non-floating side multiplication result. (60); a low-pass filter (67, 68) for removing a high-frequency component from the multiplication result on the non-floating side to restore the analog input signal on the floating side; and for oscillating a high-frequency sine wave signal. Oscillation means (18)
And for insulating the floating side from the non-floating side so as to obtain the high-frequency sine wave signal from the oscillating means (18) as the floating-side high-frequency sine wave signal and the non-floating-side high-frequency sine wave signal. High frequency sine wave insulating means (12, 103, 12
6,123,134,139,159,149,14
4) A wideband analog isolation circuit including: 2. The multiplication means (2) on the floating side.
0) and the multiplication means (6
2. The wideband analog isolation circuit according to claim 1, wherein 0) comprises a Gilbert type multiplication circuit. 3. The input signal insulating means and the high frequency sine wave insulating means are respectively provided with insulating transformers (11,
2. The broadband analog isolation circuit according to claim 1, wherein the circuit is configured as in (12). 4. The input signal insulating means and the high frequency sine wave insulating means are respectively provided with capacitors (101,
2. The broadband analog isolation circuit according to claim 1, wherein said analog isolation circuit comprises: 5. The antenna (1) wherein said input signal insulating means and said high frequency sine wave insulating means are opposed to each other.
21. The wide-band analog isolation circuit according to claim 1, wherein the analog isolation circuit is composed of: 21, 124, 122, 125, 126, 123). 6. The input signal insulating means and the high-frequency sine wave insulating means each include an optical fiber (156,
157, 159) by inserting and receiving optical fibers (131, 136, 156, 1) using optical fibers for transmitting and receiving light oppositely.
46, 141, 132, 137, 157, 147, 14
2. The wide-band analog isolation circuit according to claim 1, wherein the analog isolation circuit comprises: 2,134,139,159,149,144). 7. The light transmitting and receiving means (131, 13), wherein the input signal insulating means and the high-frequency sine wave insulating means are opposed to each other by inserting spaces to transmit and receive light.
6,146,141,132,137,147,14
2. The wideband analog isolation circuit according to claim 1, wherein said circuit is comprised of: 2,134,139,149,144). 8. A floating-side first multiplying means (20) for multiplying a floating-side analog input signal and a floating-side high-frequency sine wave signal to obtain a result of the floating-side first multiplication; First input signal insulation means (11, 101, 102) for insulating the floating side and the non-floating side and using the result of the first multiplication on the floating side as a first input signal on the non-floating side; , 121,
122, 124, 125, 131, 136, 156, 1
46, 141, 132, 137, 157, 147, 14
2) multiplying the non-floating side first input signal and the non-floating side high-frequency sine wave signal to obtain a non-floating side first multiplication result. A floating-side second multiplying means for multiplying the floating-side analog input signal by the floating-side high-frequency cosine wave signal to obtain a floating-side second multiplication result; A second input signal insulating means (11B, 101B) for insulating the floating side and the non-floating side and using the result of the second multiplication on the floating side as a second input signal on the non-floating side. , 102,12
1,122,124,125,131,136,15
6,146,141,132,137,157,14
7, 142), and multiplying the non-floating-side second input signal by the non-floating-side high-frequency cosine wave signal to obtain a non-floating-side second multiplication result. Multiplication means (60B), and oscillation means (18) for oscillating a high-frequency sine wave signal
The high-frequency sine wave signal from the oscillating means (18) is converted to the floating-side high-frequency sine wave signal, the non-floating-side high-frequency sine-wave signal, the floating-side high-frequency cosine wave signal, and the non-floating-side high-frequency cosine. High frequency sine wave insulating means (12, 25, 65, 103, 126, 1) for insulating the floating side from the non-floating side to obtain a wave signal.
23, 134, 139, 159, 149, 144). 9. The first and second floating-side circuits
Multiplication means (20, 20B) and the first and second multiplication means (60, 60B) on the non-floating side.
9. The wideband analog isolation circuit of claim 8, wherein B) comprises a Gilbert-type multiplication circuit. 10. The wideband analog insulation circuit according to claim 8, wherein said first and second input signal insulation means and said high-frequency sine wave insulation means are respectively constituted by insulation transformers (11, 11B, 12). . 11. The wideband analog insulation circuit according to claim 8, wherein said first and second input signal insulation means and said high frequency sine wave insulation means are each constituted by a capacitor (101, 102, 103). 12. The antenna according to claim 1, wherein said first and second input signal insulation means and said high-frequency sine wave insulation means are opposed to each other by antennas (121, 124, 122, 125, 1).
26. The broadband analog isolation circuit according to claim 8, wherein said circuit is comprised of: 13. An optical fiber for transmitting and receiving light, wherein said first and second input signal insulating means and said high frequency sine wave insulating means face each other by inserting optical fibers (156, 157, 159). Light transmitting and receiving means (131,
136, 156, 146, 141, 132, 137, 1
57, 147, 142, 134, 139, 159, 14
9. The wideband analog isolation circuit according to claim 8, wherein said circuit is comprised of: 14. The first and second input signal insulating means and the high-frequency sine wave insulating means respectively include a light transmitting and receiving means (131, 136) formed by a space for transmitting and receiving light by opposing a space. 146,141,132,13
7,147,142,134,139,149,14
9. The wideband analog isolation circuit according to claim 8, wherein said circuit is configured as in 4).