JPH11127206A - High frequency power amplifier - Google Patents

Abstract

PROBLEM TO BE SOLVED: To make a device compact and inexpensive by simplifying a structure without degrading a performance of a phase control circuit for compensating a nonlinear distortion. SOLUTION: A sample timing signal generation part 36 of a high frequency power amplifier 10 makes a sample timing signal Ts generate in a timing (I=0) in which an input baseband signal vector of a burst rising passes a Q axis. In an A/D converter 34, a sampling is applied to a feedback baseband signal I' by using this sample timing signal Ts and a sample signal I's is acquired. In a phase control circuit 38, it becomes possible to estimate a phase difference Δθon the basis of the sample signal I's, and a nonlinear distortion can be appropriately compensated by obtaining the real amount P of phase control for a phase shifter 32. Thus, since the number of the A/D converters 34 becomes small, a structure is simplified and can be made compact and inexpensive.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a high-frequency power amplifier, and more particularly, to a high-frequency power amplifier of a transmitter used for wireless communication such as a digital portable telephone.

[0002]

2. Description of the Related Art Conventionally, in digital wireless communication such as a digital cellular phone, quaternary PSK (Phase Shift Keying).
When a linear modulation system such as 16QAM (Quadrature Amplitude Modulation) or the like is used, the requirements for the linearity of the high-frequency power amplifier have become severe. For example, as one method of linearizing a high-frequency power amplifier, there is a Cartesian loop type negative feedback amplifier that compensates for nonlinear distortion by demodulating the output of the power amplifier and performing negative feedback in the form of a baseband signal. FIG. 4 shows this conventional high-frequency power amplifier 5.
0 is a block diagram showing a schematic configuration of the O.0. In FIG. 4, a terminal 52 has an input baseband signal I, a terminal 5
4 is supplied with an input baseband signal Q. Further, the subtracters 56 and 58 input the modulated signals Ix and Qx obtained by subtracting the feedback baseband signals I ′ and Q ′ from the input baseband signals I and Q, respectively, to the quadrature modulator 60. The quadrature modulator 60 quadrature-modulates the carrier signal of the angular frequency ωc generated by the oscillator 70 described later by the modulation signals Ix and Qx to obtain a modulation wave S represented by the following equation. S = Ix cos ωct + Qx sin ωct This quadrature modulated wave S is a power amplifier 6 having nonlinear distortion.
2 and becomes a transmission signal SA,
Although part of the transmission signal SA is radiated, a part of the transmission signal SA is branched by a coupler or the like and supplied to the attenuator 66. The attenuator 66 outputs a quadrature modulated wave S obtained by attenuating the transmission signal SA to a predetermined level.
B is supplied to a quadrature demodulator 68. The quadrature demodulator 68 demodulates the quadrature modulated wave SB using the demodulated carrier signal generated by the oscillator 70 and changing the phase of the carrier signal of the angular frequency ωc by the phase shifter 72, and returns the feedback baseband signals I ′ and Q ′. Get. The feedback baseband signals I 'and Q' output from the quadrature demodulator 68 are affected by the nonlinear distortion of the power amplifier 62.
And Q ′ are supplied to the subtractors 56 and 58 as described above to perform a negative feedback, thereby compensating for the nonlinear distortion.

As described above, in the high-frequency power amplifier 50 that performs negative feedback, the quadrature modulated wave SB is delayed as compared with the transmission signal SA due to the loop length of the negative feedback circuit, the frequency characteristics of the power amplifier 62, and the like. Have different carrier phases. Here, if the phase amount of the phase shifter 72 is fixed, and if the delay amount changes due to a change in the temperature characteristic of the power amplifier 62 or a change in the load of the antenna 64, as shown in FIG. A feedback baseband signal vector, which is a composite vector of feedback baseband signals I ′ and Q ′, rotates around a phase with respect to an input baseband signal vector, which is a composite vector of I and Q. As a result, the distortion compensation characteristics of the negative feedback amplifier have deteriorated. Therefore, in the conventional Cartesian loop type negative feedback amplifier (high frequency power amplifier 50), as shown in FIG. 4, the input baseband signals I and Q and the feedback baseband signals I 'and Q' are respectively converted into A / A signals. D converter 7
4, 76, 78, and 80, are converted into digital signals by A / D conversion, and the phase difference Δθ between the two is calculated based on the following equation. Δθ = tan −1 (Q ′ / I ′) − tan −1 (Q / I) In the phase control circuit 82, the phase of the phase shifter 72 is set to Δ
By outputting a control signal P that changes by θ,
The distortion compensation characteristic deterioration of the negative feedback amplifier due to the phase rotation was reduced.

[0004]

However, in such a conventional high-frequency power amplifier 50, the input baseband signals I and Q and the feedback of the input baseband signals I and Q are reduced in order to reduce the deterioration of the distortion compensation characteristic of the negative feedback amplifier due to the phase rotation. After separately A / D converting the baseband signals I ′ and Q ′,
Since the phase difference Δθ was calculated, the four A / D converters 7
4,76,78,80 are required, and each A / D
The conversion timings of the converters must also be accurately matched, and the amount of calculation for calculating the phase difference Δθ becomes large, resulting in high costs and a large circuit scale. there were. The present invention has been made in view of the disadvantages of the related art,
An object of the present invention is to reduce the size and cost of a Cartesian loop type negative feedback amplifier for compensating for nonlinear distortion by simplifying the configuration of the phase control circuit without deteriorating performance. It is an object of the present invention to provide a high-frequency power amplifier capable of performing the above. Another object of the present invention is to provide a high-frequency power amplifier that can simplify a circuit configuration by using a sample timing signal from which a phase difference Δθ can be estimated.

[0005]

According to a first aspect of the present invention, there is provided a subtractor for obtaining a modulation signal by subtracting a feedback baseband signal from an input baseband signal, an oscillator for generating a carrier signal, and the carrier signal. A quadrature modulator that modulates the modulated signal with the modulation signal to obtain a modulated wave; a power amplifier that amplifies the modulated wave; and a quadrature demodulation that quadrature demodulates a part of an output signal of the power amplifier to obtain the feedback baseband signal. A phase shifter for changing the phase of the carrier signal to input to the quadrature demodulator, and a phase control circuit for controlling the phase shifter, wherein a feedback base from the quadrature demodulator is provided. An A / D converter for digitizing an in-phase component or a quadrature component of a band signal; and a sample signal capable of estimating a phase difference between the input baseband signal and the feedback baseband signal. And a sample timing signal generator for supplying a sample timing signal to be generated by the A / D converter. The phase control circuit performs the input baseband signal based on the sample signal output from the A / D converter. And the phase difference of the feedback baseband signal is estimated to control the phase of the phase shifter.
According to this, a subtractor subtracts a feedback baseband signal from an input baseband signal to obtain a modulation signal, and a carrier wave signal from an oscillator is modulated by a modulation signal in a quadrature modulator to obtain a modulation wave, A power amplifier amplifies the modulated wave, a quadrature demodulator performs quadrature demodulation on a part of the output signal of the power amplifier to obtain a feedback baseband signal, and a phase shifter changes the phase of the carrier signal to perform quadrature demodulation. The phase shifter is controlled by a phase control circuit. Therefore, when the in-phase component or the quadrature component of the feedback baseband signal from the quadrature demodulator is digitized by the A / D converter, the phase difference between the input baseband signal and the feedback baseband signal is determined by the sample timing signal generator. Since the sample timing signal for generating the estimable sample signal by the A / D converter is generated, the circuit configuration for compensating for the nonlinear distortion without deteriorating the performance is simplified, and the configuration is small and inexpensive. be able to.

According to a second aspect of the present invention, in the high-frequency power amplifier according to the first aspect, the sample timing signal generating section includes at least one of an in-phase component and a quadrature component of the input baseband signal in a transmission burst rising section. Is generated so as to generate a sample timing signal that coincides with the timing at which is zero.
According to this, the sample timing signal generated by the sample timing signal generator is set to coincide with the timing at which at least one of the in-phase component and the quadrature component of the input baseband signal becomes zero in the transmission burst rising section. And the phase difference Δθ can be estimated by the phase control circuit, and the circuit configuration can be simplified.

[0007]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An embodiment of the present invention will be described below in detail with reference to FIGS. FIG. 1 is a block diagram illustrating a schematic configuration of a high-frequency power amplifier 10 according to the present embodiment. In FIG. 1, a high-frequency power amplifier 10 includes subtractors 16 and 18, a quadrature modulator 20,
Power amplifier 22, antenna 24, attenuator 26, quadrature demodulator 28, oscillator 30, phase shifter 32, A / D converter 34,
Sample timing signal generator 36, phase control circuit 38
And so on. The high-frequency power amplifier according to the present embodiment is constituted by a Cartesian loop type negative feedback amplifier. First, an input baseband signal I (in-phase component) is input to a terminal 12, and an input baseband signal Q (quadrature component) is input to a terminal 14. Subtractors 16 and 1
Numeral 8 is for obtaining modulated signals Ix and Qx by subtracting feedback baseband signals I 'and Q' to be described later from input baseband signals I and Q, respectively. Then, the modulated signals Ix and Qx are input to the quadrature modulator 20 at the next stage. The quadrature modulator 20 quadrature-modulates a carrier signal having an angular frequency ωc generated by an oscillator 30, which will be described later, with modulation signals Ix and Qx to obtain a modulation wave S represented by the following equation. S = Ix cos ωct + Qx sin ωct The power amplifier 22 amplifies the quadrature modulated wave S and outputs the transmission signal SA, but has nonlinear distortion. The transmission signal SA output from the power amplifier 22 is radiated from the antenna 24 and the transmission signal SA
Is branched by a coupler or the like and supplied to the attenuator 26.

[0008] The attenuator 26 generates a quadrature modulated wave SB in which the transmission signal SA is attenuated to a predetermined level, and supplies it to the quadrature demodulator 28. The quadrature demodulator 28 demodulates the quadrature modulated wave SB with the demodulation carrier signal generated by the oscillator 30 and changing the phase of the carrier signal of the angular frequency ωc by the phase shifter 32, thereby returning the feedback baseband signals I ′ and Q ′. Is what you get. The feedback baseband signals I ′ and Q ′ output from the quadrature demodulator 28 are affected by the nonlinear distortion of the power amplifier 22 described above. Non-linear distortion is compensated by supplying negative feedback to the devices 16 and 18. As described above, when an attempt is made to compensate for nonlinear distortion by applying negative feedback in the high-frequency power amplifier 10, the orthogonal modulation wave SB is delayed compared to the transmission signal SA due to the loop length of the negative feedback circuit, the frequency characteristics of the power amplifier 22, and the like. However, when the carrier wave phases of the two are different from each other and the amount of delay changes due to a change in the temperature characteristic of the power amplifier 22 or a change in the load of the antenna 24, the feedback baseband signal I ′ is added to the input baseband signal vector.
As described in the description of the related art, the feedback baseband signal vector, which is a composite vector of Q and Q ′, rotates around the phase and the distortion compensation characteristic of the negative feedback amplifier deteriorates. Therefore, a feature of the high-frequency power amplifier 10 of the present embodiment is that the feedback baseband signal I ′ is converted into an A / D signal by a sample timing signal Ts at a burst rising edge described later.
The point is that the phase control of the phase shifter 32 is performed using the converted sample signal I's.

Hereinafter, this characteristic configuration will be described. The transmission format of the high-frequency power amplifier 10 according to the present embodiment is a burst transmission as shown in FIG. 2, and the input baseband signal vector at the burst rising is shown in FIG.
It is assumed that a locus as shown in FIG. This premise is based on most TDMA (Time Division Multiple A
ccess) type digital mobile communication wireless transmitters and the like. It is also assumed that the feedback baseband signal vector at the burst rising at this time draws a locus as shown in FIG. Then, FIG.
The phase difference between the phase trajectory of the feedback baseband signal vector shown in FIG. 3A and the phase trajectory of the input baseband signal vector in FIG. Here, the zero-cross timing (that is, the timing when I = 0) at which the input baseband signal vector at the burst rising edge shown in FIG. 3A passes through the Q axis is known. The sample timing signal generator 36 of the high frequency power amplifier 10 shown in FIG. 1 generates a pulse at this timing (see the sample timing signal Ts in FIG. 2). The A / D converter 34 samples the feedback baseband signal I 'using the sample timing signal Ts as a sample timing to obtain a sample signal I's. The phase control circuit 38 can estimate the phase difference Δθ obtained from the above-mentioned sample signal I ′s. That is, since the sign of the sample signal I's indicates the lead and lag of the phase, and the magnitude corresponds to the amount around the phase, the actual phase control amount P can be obtained by weighting the sample signal I's.

As described above, according to the present embodiment, the phase control is performed by using the sample signal obtained by sampling the feedback baseband signal using the sample timing signal Ts generated by the sample timing signal generator 36. It is possible to estimate the phase difference Δθ determined in the circuit 38. Therefore, the conventional example (see FIG. 4)
In this embodiment, four A / D converters 74, 76, 78, and 80 were required, but in the present embodiment, the same characteristics are obtained only by one A / D converter 34 and the sample timing signal generator 36. As a result, it is possible to obtain a small and inexpensive high-frequency power amplifier 10 without performance degradation. Note that, depending on the configuration of the input baseband signal at the rising edge of the burst, the timing at which the feedback baseband signal vector passes through the I axis (that is, Q
= 0) from the sample timing signal generation unit 36,
In this case, by using the signal Q's obtained by sampling the feedback baseband signal Q ', the phase control circuit 3
8, the phase difference Δθ may be estimated, and the same effect as in the above embodiment can be obtained.

[0011]

As described above, according to the first aspect of the present invention, the configuration of the phase control circuit of the Cartesian loop type negative feedback amplifier for compensating for non-linear distortion does not deteriorate the performance. By simplification, it can be made small and inexpensive. An object of the present invention described in claim 2 is to simplify the circuit configuration by using a sample timing signal from which the phase difference Δθ can be estimated.

[Brief description of the drawings]

FIG. 1 is a block diagram showing a schematic configuration of a high-frequency power amplifier according to the present embodiment.

FIG. 2 is a timing chart showing burst transmission and sample timing according to the present embodiment.

3A and 3B are diagrams illustrating a relationship between a baseband signal trajectory and a sample timing in a burst rising section, where FIG. 3A illustrates an input baseband signal trajectory, and FIG. 3B illustrates a feedback baseband signal trajectory.

FIG. 4 is a block diagram showing a schematic configuration of a conventional high-frequency power amplifier.

FIG. 5 is a diagram illustrating a phase difference Δθ between an input baseband signal vector and a feedback baseband signal vector.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 10 High frequency power amplifier 16, 18 Subtractor 20 Quadrature modulator 22 Power amplifier 24 Antenna 26 Attenuator 28 Quadrature demodulator 30 Oscillator 32 Phase shifter 34 A / D converter 36 Sample timing signal generation part 38 Phase control circuit

Claims (2)

[Claims] A subtractor for obtaining a modulation signal by subtracting a feedback baseband signal from an input baseband signal; an oscillator for generating a carrier signal; and a quadrature for modulating the carrier signal with the modulation signal to obtain a modulation wave. A modulator, a power amplifier that amplifies the modulated wave, a quadrature demodulator that obtains the feedback baseband signal by orthogonally demodulating a part of the output signal of the power amplifier, and changing the phase of the carrier signal to change the phase of the carrier signal. In a high-frequency power amplifier having a phase shifter input to a quadrature demodulator and a phase control circuit for controlling the phase shifter, A which digitizes an in-phase component or a quadrature component of a feedback baseband signal from the quadrature demodulator A / D converter, and a sampler for generating, by the A / D converter, a sample signal from which a phase difference between the input baseband signal and the feedback baseband signal can be estimated. And a sample timing signal generator that supplies a timing signal based on the sample signal output from the A / D converter. The phase control circuit determines a phase difference between the input baseband signal and the feedback baseband signal based on the sample signal output from the A / D converter. A high-frequency power amplifier, which estimates and controls the phase of the phase shifter. 2. The apparatus according to claim 1, wherein said sample timing signal generator generates a sample timing signal which coincides with a timing at which at least one of an in-phase component and a quadrature component of said input baseband signal becomes zero in a transmission burst rising section. The high frequency power amplifier according to claim 1, wherein: