JP3865336B2 - High frequency power amplifier - Google Patents

Description

[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a high-frequency power amplifier, and more particularly to a high-frequency power amplifier for a transmitter used for radio communication such as a digital cellular phone.
[0002]
[Prior art]
Conventionally, in the case of using a linear modulation method such as 4-level PSK (Phase Shift Keying) or 16QAM (Quadrature Amplitude Modulation) in digital wireless communication such as a digital cellular phone, the demand for linearity of a high-frequency power amplifier has been severe. .
For example, there is a Cartesian loop type negative feedback amplifier that compensates for nonlinear distortion by demodulating the power amplifier output and applying negative feedback in the form of a baseband signal as one of the linearization techniques of the high-frequency power amplifier. FIG. 4 is a block diagram showing a schematic configuration of this conventional high-frequency power amplifier 50. In FIG. 4, an input baseband signal I is input to the terminal 52, and an input baseband signal Q is input to the terminal 54. Subtractors 56 and 58 input modulated signals Ix and Qx obtained by subtracting feedback baseband signals I ′ and Q ′ from input baseband signals I and Q, respectively, to quadrature modulator 60. The quadrature modulator 60 performs quadrature modulation on a carrier wave signal having an angular frequency ωc generated by an oscillator 70, which will be described later, with modulation signals Ix and Qx to obtain a modulated wave S as represented by the following equation.
S = Ix cos ωct + Qx sin ωct
This orthogonal modulation wave S is amplified by a power amplifier 62 having nonlinear distortion to become a transmission signal SA and radiated from an antenna 64. A part of the transmission signal SA is branched by a coupler or the like and supplied to an attenuator 66. The The attenuator 66 supplies a quadrature modulated wave SB obtained by attenuating the transmission signal SA to a predetermined level to the quadrature demodulator 68. The quadrature demodulator 68 demodulates the quadrature modulated wave SB by using the demodulated carrier signal obtained by changing the phase of the carrier signal having the angular frequency ωc generated by the oscillator 70 by the phase shifter 72, and feedback baseband signals I ′ and Q ′. Get.
Although the feedback baseband signals I ′ and Q ′ output from the quadrature demodulator 68 are affected by the nonlinear distortion of the power amplifier 62, the feedback baseband signals I ′ and Q ′ are subtracted from the subtractors 56 and 56 as described above. The non-linear distortion is compensated by supplying negative feedback to 58.
[0003]
As described above, in the high-frequency power amplifier 50 that performs negative feedback, the quadrature modulation wave SB is generally delayed compared to the transmission signal SA due to the loop length of the negative feedback circuit, the frequency characteristics of the power amplifier 62, and the like, and both carrier phase Are different from each other. Here, assuming that the phase amount of the phase shifter 72 is fixed, if the delay amount changes due to the temperature characteristic variation of the power amplifier 62 or the load variation of the antenna 64, as shown in FIG. A feedback baseband signal vector that is a combined vector of feedback baseband signals I ′ and Q ′ causes a phase shift with respect to an input baseband signal vector that is a combined vector of I and Q. As a result, the distortion compensation characteristic of the negative feedback amplifier has deteriorated.
Therefore, in the conventional negative feedback amplifier of the Cartesian loop type (high frequency power amplifier 50), as shown in FIG. 4, the input baseband signals I and Q and the feedback baseband signals I ′ and Q ′ are respectively A / The D converters 74, 76, 78, and 80 are used to perform A / D conversion to convert them into digital signals, and the phase difference Δθ between them is calculated based on the following equation.
Δθ = tan −1 (Q ′ / I ′) − tan −1 (Q / I)
The phase control circuit 82 outputs a control signal P that changes the phase of the phase shifter 72 by Δθ, thereby reducing the distortion compensation characteristic deterioration of the negative feedback amplifier due to the phase shift.
[0004]
[Problems to be solved by the invention]
However, in such a conventional high-frequency power amplifier 50, the input baseband signals I and Q and the feedback baseband signals I ′ and Q ′ are used to reduce the distortion compensation characteristic deterioration of the negative feedback amplifier due to the phase shift. Since the phase difference Δθ was calculated after each A / D conversion, four A / D converters 74, 76, 78, and 80 are required, and the conversion timing of each A / D converter is also accurate. Since the amount of calculation for calculating the phase difference Δθ must be increased, there is a disadvantage in that the cost is increased and the circuit scale must be increased.
The present invention has been made in view of the disadvantages of the prior art, and an object of the present invention is to configure a phase control circuit of a Cartesian loop type negative feedback amplifier for compensating for nonlinear distortion. It is an object of the present invention to provide a high-frequency power amplifier that can be made small and inexpensive by simplifying the above without degrading the performance.
An object of the present invention is to provide a high-frequency power amplifier that can simplify the circuit configuration by using a sample timing signal that can estimate the phase difference Δθ.
[0005]
[Means for Solving the Problems]
According to the first aspect of the present invention, a subtractor that subtracts a feedback baseband signal from an input baseband signal to obtain a modulation signal, an oscillator that generates a carrier wave signal, and modulates the carrier wave signal by modulating the modulation signal. A quadrature modulator for obtaining a wave; a power amplifier for amplifying the modulated wave; a quadrature demodulator for obtaining a feedback baseband signal by quadrature demodulating a part of an output signal of the power amplifier; and a phase of the carrier signal In a high-frequency power amplifier having a phase shifter that is input to the quadrature demodulator and a phase control circuit that controls the phase shifter, the in-phase component or quadrature component of the feedback baseband signal from the quadrature demodulator An A / D converter for digitizing, and a sample signal for generating a sample signal capable of estimating a phase difference between the input baseband signal and the feedback baseband signal by the A / D converter A phase difference between the input baseband signal and the feedback baseband signal in the phase control circuit based on the sample signal output from the A / D converter. And the phase of the phase shifter is controlled.
According to this, a subtractor subtracts a feedback baseband signal from an input baseband signal to obtain a modulated signal, and a modulated wave is obtained by modulating a carrier wave signal from an oscillator with a modulated signal in a quadrature modulator, The modulated wave is amplified by the power amplifier, a part of the output signal of the power amplifier is orthogonally demodulated by the quadrature demodulator to obtain a feedback baseband signal, and the phase of the carrier wave signal is changed by the phase shifter to perform quadrature demodulation. The phase shifter is controlled by the phase control circuit. Therefore, when the in-phase component or quadrature component of the feedback baseband signal from the quadrature demodulator is digitized by the A / D converter, the phase difference between the input baseband signal and the feedback baseband signal is calculated by the sample timing signal generation unit. Since the sample timing signal for generating the estimable sample signal by the A / D converter is generated, the circuit configuration for compensating the non-linear distortion without degrading the performance is simplified, and the configuration is small and inexpensive. be able to.
[0006]
According to a second aspect of the present invention, in the high-frequency power amplifier according to the first aspect, the sample timing signal generator is configured such that at least one of the in-phase component and the quadrature component of the input baseband signal is zero in the transmission burst rising period. It is characterized in that a sample timing signal that coincides with a given timing is generated.
According to this, the sample timing signal generated by the sample timing signal generation unit matches the timing at which at least one of the in-phase component and the quadrature component of the input baseband signal becomes zero in the transmission burst rising section. The phase difference Δθ can be estimated by the phase control circuit, and the circuit configuration can be simplified.
[0007]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, an embodiment of the present invention will be described in detail with reference to FIGS.
FIG. 1 is a block diagram showing a schematic configuration of a high-frequency power amplifier 10 according to the present embodiment. In FIG. 1, a high-frequency power amplifier 10 includes subtractors 16 and 18, a quadrature modulator 20, a power amplifier 22, an antenna 24, an attenuator 26, a quadrature demodulator 28, an oscillator 30, a phase shifter 32, and an A / D converter. 34, a sample timing signal generator 36, a phase control circuit 38, and the like. The high-frequency power amplifier according to the present embodiment is configured by a Cartesian loop type negative feedback amplifier.
First, the input baseband signal I (in-phase component) is input to the terminal 12, and the input baseband signal Q (quadrature component) is input to the terminal 14.
Subtractors 16 and 18 subtract feedback baseband signals I ′ and Q ′ described later from input baseband signals I and Q, respectively, to obtain modulation signals Ix and Qx. The modulated signals Ix and Qx are input to the next-stage quadrature modulator 20.
The quadrature modulator 20 performs a quadrature modulation on a carrier wave signal having an angular frequency ωc generated by an oscillator 30 to be described later using modulated signals Ix and Qx to obtain a modulated wave S as shown in the following equation.
S = Ix cos ωct + Qx sin ωct
The power amplifier 22 amplifies the quadrature modulated wave S and outputs the transmission signal SA, but has nonlinear distortion. The transmission signal SA output from the power amplifier 22 is radiated from the antenna 24, and a part of the transmission signal SA is branched by a coupler or the like and supplied to the attenuator 26.
[0008]
The attenuator 26 generates a quadrature modulation wave SB obtained by attenuating the transmission signal SA to a predetermined level, and supplies the quadrature demodulator 28 with the quadrature modulation wave SB. The quadrature demodulator 28 demodulates the quadrature modulated wave SB by demodulating the carrier wave signal of the angular frequency ωc generated by the oscillator 30 using the phase shifter 32 to change the phase of the feedback modulated baseband signals I ′ and Q. Is what you get. Although the feedback baseband signals I ′ and Q ′ output from the quadrature demodulator 28 are affected by the nonlinear distortion of the power amplifier 22 described above, the feedback baseband signals I ′ and Q ′ are subtracted as described above. The nonlinear distortion is compensated by supplying negative feedback to the devices 16 and 18.
As described above, if the high-frequency power amplifier 10 performs negative feedback to compensate for nonlinear distortion, the orthogonal modulation wave SB is delayed compared to the transmission signal SA due to the loop length of the negative feedback circuit, the frequency characteristics of the power amplifier 22, and the like. When the carrier phases of the two are different from each other and the delay amount changes due to the temperature characteristic variation of the power amplifier 22 or the load variation of the antenna 24, the feedback baseband signals I ′ and Q ′ are changed with respect to the input baseband signal vector. As described in the description of the prior art, the feedback baseband signal vector, which is a combined vector, causes a phase shift and the distortion compensation characteristics of the negative feedback amplifier deteriorate.
Therefore, the high-frequency power amplifier 10 according to the present embodiment is characterized by the phase control of the phase shifter 32 using the sample signal I ′s obtained by A / D conversion of the feedback baseband signal I ′ with the sample timing signal Ts at the burst rise described later. It is in the point which was made to do.
[0009]
Hereinafter, this characteristic configuration will be described.
The transmission format of the high-frequency power amplifier 10 of the present embodiment is burst transmission as shown in FIG. 2, and the input baseband signal vector at the rising edge of the burst draws a locus as shown in FIG. To do. This premise applies to most TDMA (Time Division Multiple Access) digital mobile communication radio transmitters. Also, assume that the feedback baseband signal vector at the burst rising at this time draws a locus as shown in FIG. Then, the phase difference Δθ is obtained as the phase difference between the phase locus of the feedback baseband signal vector shown in FIG. 3B and the phase locus of the input baseband signal vector shown in FIG.
Here, the timing of the zero crossing when the input baseband signal vector of the burst rising shown in FIG. 3A passes through the Q axis (that is, the timing when I = 0) is known. The sample timing signal generator 36 of the high-frequency power amplifier 10 shown in FIG. 1 generates a pulse at this timing (see the sample timing signal Ts in FIG. 2). The A / D converter 34 described above samples the feedback baseband signal I ′ using the sample timing signal Ts as the sample timing to obtain the sample signal I ′s.
The phase control circuit 38 can estimate the phase difference Δθ obtained from the sample signal I ′s described above. That is, since the sign of the sample signal I ′s represents the phase advance / delay and the magnitude corresponds to the amount around the phase, the actual phase control amount P can be obtained by weighting the sample signal I ′s.
[0010]
As described above, according to the present embodiment, the phase control circuit 38 uses the sample signal obtained by sampling the feedback baseband signal using the sample timing signal Ts generated by the sample timing signal generation unit 36. It is possible to estimate the required phase difference Δθ. For this reason, in the conventional example (see FIG. 4), four A / D converters 74, 76, 78, and 80 are required. However, in this embodiment, one A / D converter 34 and sample timing are used. Since equivalent characteristics can be obtained only by the signal generator 36, the high-frequency power amplifier 10 can be made small and inexpensive without performance deterioration.
Depending on the configuration of the input baseband signal at the rising edge of the burst, in order to obtain the phase difference Δθ, a sample timing signal consisting of the timing at which the feedback baseband signal vector passes through the I axis (that is, the timing at which Q = 0) is used. The phase control circuit 38 may estimate the phase difference Δθ by using the signal Q ′s generated from the sample timing signal generator 36 and in this case using the sampled feedback baseband signal Q ′. The same effect as in the embodiment can be obtained.
[0011]
【The invention's effect】
As described above, according to the invention described in claim 1, by simplifying the configuration of the phase control circuit of the Cartesian loop type negative feedback amplifier for compensating the nonlinear distortion without degrading the performance. , Small and inexpensive. According to the second aspect of the present invention, the circuit configuration can be simplified by using the sample timing signal capable of estimating the phase difference Δθ.
[Brief description of the drawings]
FIG. 1 is a block diagram showing a schematic configuration of a high-frequency power amplifier according to an embodiment.
FIG. 2 is a timing chart showing burst transmission and sample timing according to the present embodiment.
FIGS. 3A and 3B are diagrams showing a relationship between a baseband signal locus and a sample timing in a burst rising section, where FIG. 3A is an input baseband signal locus, and FIG. 3B is a feedback baseband signal locus;
FIG. 4 is a block diagram showing a schematic configuration of a conventional high-frequency power amplifier.
FIG. 5 is a diagram illustrating a phase difference Δθ between an input baseband signal vector and a feedback baseband signal vector.
[Explanation of symbols]
DESCRIPTION OF SYMBOLS 10 High frequency power amplifier 16, 18 Subtractor 20 Quadrature modulator 22 Power amplifier 24 Antenna 26 Attenuator 28 Quadrature demodulator 30 Oscillator 32 Phase shifter 34 A / D converter 36 Sample timing signal generation part 38 Phase control circuit

Claims (2)

A subtractor that subtracts a feedback baseband signal from an input baseband signal to obtain a modulated signal; an oscillator that generates a carrier signal; a quadrature modulator that modulates the carrier signal with the modulated signal to obtain a modulated wave; and A power amplifier that amplifies the modulated wave, a quadrature demodulator that obtains the feedback baseband signal by quadrature demodulating a part of the output signal of the power amplifier, and changing the phase of the carrier signal and inputting it to the quadrature demodulator And a phase control circuit for controlling the phase shifter, a high-frequency power amplifier comprising:
An A / D converter that digitizes the in-phase or quadrature component of the feedback baseband signal from the quadrature demodulator;
A sample timing signal generator for supplying a sample timing signal for generating a sample signal in the A / D converter that can estimate a phase difference between the input baseband signal and the feedback baseband signal;
With
The phase control circuit estimates the phase difference between the input baseband signal and the feedback baseband signal based on the sample signal output from the A / D converter, and controls the phase of the phase shifter. A high frequency power amplifier. The sample timing signal generation unit generates a sample timing signal that coincides with a timing at which at least one of an in-phase component and a quadrature component of the input baseband signal becomes zero in a transmission burst rising section. Item 4. The high frequency power amplifier according to Item 1.

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