JPH09284054A - Digital am demodulator and its method - Google Patents

Abstract

PROBLEM TO BE SOLVED: To obtain a demodulation signal by means of an extremely small level of a distortion even if a signal generated by means of absolute value addition, etc., does not satisfy a sampling condition by adding the whole plural digital signals together by means of an adder, executing level-correction through the use of a level correcting equipment and, then, removing a D.C. components by means of a digital high-pass filter. SOLUTION: The signal is used as an AM signal by inputting a signal from an input terminal 1 and sampling and quantizing it at a sampling frequency in an A/D converter 2. A set of n signals with desired phase differences are generated by a PSN matrix 9 from the discretized AM signal, the absolute values of respective signals are obtained and the n signals whose absolute values are taken are added in an adder 4. Then, the digital high-pass filter 6 removes the D.C. component from the signal which is corrected by a corrected coefficient generator 16 and a level correcting equipment 5, and the signal becomes the envelope of to the AM signal itself. After that, the signal is converted into an analog signal by a D/A converter 7 and an AM demodulating signal which is proportional the envelope of the carrier is outputted to an output terminal 8.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a digital AM demodulator and its method.

[0002]

2. Description of the Related Art For example, in the case of demodulating an AM signal in a receiver such as a wireless communication device, there have been demodulation methods such as envelope detection and square detection. Since these detection circuits are composed of analog elements such as diodes, the operation of the circuits becomes unstable due to changes in ambient temperature and humidity and changes over time, making it difficult to detect modulated signals quantitatively. There was a case. Therefore, recently, with the development of digital signal processing technology, various demodulators and methods for AM signals by digital processing have been proposed.

As a conventional technique (1) for digital AM demodulation,
There is an envelope detection method in which a squared sum square root process is performed to extract a modulated signal proportional to the envelope of the carrier. According to this method shown in FIG. 2, first, an input analog AM signal is converted into a digital AM signal Xin by an analog / digital converter (hereinafter, referred to as an A / D converter), and is converted into a digital AM demodulator from a terminal 20. Generally, the signal input to the A / D converter, the signal input to the D / A converter, and the like, and the sampling frequency (hereinafter, also referred to as sampling frequency) fs are input in the equation (1). You need to be happy with the relationship. However,
Let ts be the sampling time, tc be the carrier period, and fc be the carrier frequency. fc ≦ 2fs ∴ts ≦ tc / 2 (1)

Further, the relation of the equation (1) is satisfied, and the equation (2) is satisfied.
When sampling at the sampling frequency shown in
The signal obtained by squaring the digital AM signal Xin at the multiplier 22 and the signal obtained by squaring the digital AM signal Xin at the multiplier 23 by the digital phase shifter 21 configured by the register D There are q sample points with a π / 2 phase difference of the digital AM signal. However, q is an integer. 4fc = fs (1 + 2q) (2)

Then, by adding these two signals by the adder 24, it is possible to generate the signal Yin which is the square value of the amplitude of the carrier wave fc. Then, the square sum signal Yin is square root-calculated by the square root function unit 25 and square rooted to obtain a digital signal.
The amplitude of the carrier wave of the AM signal can be obtained, and the DC component of the signal after square root removal is removed by the digital high-pass filter 26.
Output from 27, digital / analog converter (hereinafter, D /
It is called A converter. ) To convert to an analog signal,
A signal proportional to the amplitude level of the AM signal wave can be taken out, and a modulated signal proportional to the carrier wave envelope can be obtained.

However, the above-mentioned conventional technique (1) is sufficient in principle but is not practical, and for example, a ROM table or a dedicated digital square root square root device is used in the square root square root calculation of the square root function unit 25. However, in the case of the ROM table, there is a problem that a huge amount of ROM is required to improve the calculation accuracy of the square root expansion, and when a dedicated digital square root square root extractor is used, the square root square root extractor There is also a problem that it takes a long time for the calculation to converge to a predetermined accuracy, and in any case, the calculation accuracy of the square root is
It had a great influence on the demodulation characteristics.

Further, as the prior art (2), a digital AM demodulation system disclosed in Japanese Patent Publication No. 6-18291, which is an improvement of the calculation of the square root square root, which has caused the problem of the prior art (1). There is. As shown in FIG. 3, this method is a method in which the calculation accuracy of the square root square root calculation is performed with a constant accuracy regardless of the signal level of the input signal. The level of the signal Yin is detected by the level detector 30, and the level is set within the allowable error range of the polynomial calculator 32 that performs square root square root extraction.
After level correction with upper bit shifter 31, polynomial calculator
In this method, the square root of the sum of squares is squared at 32, and the lower bit shifter 33 restores the original signal level to obtain a modulated signal.

However, in the above prior art (2), it is necessary to correct the level of the sum-of-squares signal Yin so as to be within the allowable error range of the polynomial calculator 32 in order to improve the calculation accuracy of the square root. However, for example, it is theoretically impossible to put the signal of the sum of squares signal Yin as close as possible to zero, that is, a signal close to 100% modulation within the allowable error range of the polynomial calculator 32. Also, in order to accurately calculate the square sum squared signal Yin that is as close as possible to the level-corrected zero, the square root square root calculation requires a considerable amount of calculation, and in order to actually obtain a high-quality modulated signal, it is limited to zero. Without close 2
The multiply-add signal Yin also needs to be demodulated with high accuracy, resulting in the problem that a considerable amount of calculation is required.

On the other hand, as the prior art (3), a PSN (Phase Shift Networks) different from the digital AM demodulation method by the square root square root calculation method of the above prior arts (1) and (2) is used.
There is a digital SSB demodulation method. This system, shown in Fig. 4, is used for detection of SSB modulated signals, and it detects the sideband (hereinafter also referred to as "sideband") components at frequencies above or below the carrier by shifting to DC. Is.

However, in the above prior art (3), the PSN
The band that can be phase-shifted is limited by this, and beyond this band, an unwanted sideband (image) phase shift occurs in the demodulated signal band, that is, aliasing occurs and sideband suppression of the demodulated signal occurs. The level gets worse. The filter in the latter stage can select the optimum PSN band that suppresses this deterioration, but the group delay characteristic deteriorates to that extent. Therefore, even if these are appropriately traded off, it is difficult to obtain a modulated signal with high S / N and good quality.

Further, as the prior art (4), there is a PSN digital AM demodulation method to which the above-mentioned prior art (3) is applied. This method shown in FIG. 5 is based on the SSB of the prior art (3).
Unlike the signal, the sidebands are symmetrical with respect to the carrier wave, so after performing phase shift processing with PSN of the prior art (3),
Detection is performed by taking an absolute value.

This PSN absolute value detection method can accurately demodulate an AM signal having a large degree of modulation as compared with the digital AM demodulation methods of the above-mentioned prior arts (1) and (2) regardless of the amount of calculation of the calculator. Is.

However, in the prior art (4), even if the signal input to the A / D converter satisfies the condition of the expression (1), the signal generated when the absolute values are internally added is expressed by the expression (1)
Since the signal is input to the D / A converter without satisfying the condition (1), aliasing occurs, and there is a problem that an amplitude distortion signal is superimposed on the demodulation signal.

[0014]

In view of the problems of the various digital AM demodulation methods described above, the present invention improves the PSN absolute value detection method of the prior art (4) and adds the absolute values to generate the PSN absolute value detection method. Signal, etc. does not satisfy the condition of equation (1)
We propose a digital AM demodulator that can obtain a demodulated signal with an extremely small level of distortion even if it is A / A converted.

[0015]

An analog / digital converter, a phase shifter, a first phase coefficient generator, a first multiplier connected to the first phase coefficient generator, and a first phase coefficient generator. A second phase coefficient generator, a second multiplier connected to the second phase coefficient generator, a first multiplier for adding the output of the first multiplier and the output of the second multiplier No. PSN circuit composed of an adder of ## EQU1 ## and an absolute value converter connected to the first adder, and a second of adding respective outputs of the n PSN circuits.
The digital AM demodulator is characterized in that the adder, the level corrector, the digital high-pass filter, and the digital / analog converter are connected in cascade.

Further, in the digital AM demodulator according to claim 1, the input AM signal is first converted by the analog / digital converter.
Of the first digital signal, and a phase shifter generates second and third digital signals having a phase difference of π / 2 from the first digital signal, and the second digital signal is converted into the first digital signal.
N phase coefficients from cos (0) to cos ((n-1) π / n), each of which has a phase difference of π / n, output from the phase coefficient generator of , And the third digital signal is output from the second phase coefficient generator, each of which has a phase difference of π / n from sin (0) to sin ((n-1) π / n ) Up to n phase coefficients are multiplied by n second multipliers, and the output of the n first multipliers and the output of the n second multipliers are respectively multiplied by n. Vector synthesis is performed by the first adder, and n absolute values are taken by the n absolute value generators to generate n fourth digital signals, and the n fourth digital signals are generated as second digital signals. After adding all with the adder of, and correcting the level with the level corrector, remove the DC component with the digital high-pass filter, convert it into an analog signal with the digital / analog converter, and demodulate the signal. Provides a method of demodulating the digital AM demodulator, characterized in that to obtain. Where n
Is a natural number.

[0017]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An embodiment showing a digital AM demodulator and its method of the present invention will be described with reference to the drawings. FIG. 1 is a block diagram showing an embodiment of the present invention, where 1 is an input terminal and 2
Is an A / D converter, 3 is a digital phase shifter, 4 is a digital adder, 5 is a level corrector, 6 is a digital high-pass filter, 7 is a D / A converter, 8 is an output terminal, 9 is a PSN matrix, 10a, 10b and 10c are phase coefficient generators,
11a, 11b, 11c are phase coefficient generators, 12a, 1
2b and 12c are digital multipliers, 13a, 13b and 13
c is a digital multiplier, 14a, 14b and 14c are digital adders, and 15a, 15b and 15c are absolute value digitizers. In the following description, 10a, 10b, 10c
Let the phase coefficient generator 10 be a phase coefficient generator 10, and
The phase coefficient generators 11b and 11c are replaced by the phase coefficient generator 11
, The digital multipliers 12a, 12b, 12c are the digital multipliers 12, and the digital multipliers 13a, 13b, 13c are the digital multipliers 13, 14a, 14b,
The digital adder 14c is a digital multiplier 14, and 1
The absolute value digitizers 5a, 15b, and 15c are referred to as absolute value digitizer 15.

In the wireless communication device, the received analog AM
The signal x (equation (3)) becomes a desired IF frequency after being subjected to signal processing by the front end circuit, the IF circuit, etc., and is input from the input terminal 1 of the present invention, and is input by the A / D converter 2. Digital AM sampled and quantized at sampling frequency fs
It becomes the signal xd. It should be noted that vs is a modulation signal. x = (1 + vs) cos (ωt−ψ) (3)

The digital AM signal xd is supplied to the digital phase shifter 3
Then, there are two digital AM signals I and Q having a phase difference of π / 2 shown in equations (4) and (5). Here, for convenience, ψ
Is zero. I = (1 + vs) cosωt ・ ・ ・ (4) Q = (1 + vs) cos (ωt-π / 2) = (1 + vs) sinωt ・ ・ ・ (5)

Using these signals I and Q, n digital AM signals xdn are generated by equally dividing one cycle of the digital AM signal xd with a phase difference of π / n. As shown in the equation (6), the n digital AM signals xdn represent the desired phase difference from the preceding two orthogonal digital AM signals I and Q and the phase coefficient generators 10 and 11 located at the subsequent stage. It is generated by multiplying the derived phase coefficient signal by multipliers 12 and 13, respectively, and vector-combining them by an adder 14. xd1 = (1 + vs) cos (ωt-0) = Icos (0) π + Qsin (0) π xd2 = (1 + vs) cos (ωt- (π / n)) = Icos (π / n) + Qsin (π / n) xd3 = (1 + vs) cos (ωt- (2π / n)) = Icos (2π / n) + Qsin (2π / n) xd4 = (1 + vs) cos (ωt- (3π / n)) = Icos (3π / n) + Qsin (3π / n) ・ ・ ・ ・ ・ ・ ・ ・ ・ ・ xdn = (1 + vs) cos (ωt-((n-1) π / n)) = Icos ((n-1) π / n) + Qsin ((n-1) π / n) (6)

Next, in the absolute value converter 15, n digital AMs are input.
Each of the signals xdn is set to an absolute value, and this absolute value is taken n
The digital AM signals | xdn | are all added by the digital adder 4 as shown in equations (7) and (8). | Xd1 | + | xd2 | + | xd3 | + | xd4 | + ... + | xdn | = (1 + vs) Σ | Icos (i / n) π + Qsin (i / n) π | (however, i = 0 ~ n-1) ・ ・ ・ (7) = (1 + vs) Σ ｜ cos (ωt- (i / n) π) ｜ (where i = 0 〜 n-1) ・ ・ ・ (8 )

Here, Σ | cos (ωt- (i / n) in equation (8)
π) | terms, f (ωt) = Σ | cos (ωt- (i / n) π)
When the formula is expanded as |, the expansion formula of the function f (ωt) is different when n is an even number and when it is an odd number. When n is an even number, the expansion formula of the function f (ωt) is as follows. [0 <ωt ≦ π / n] f (ωt) = Cmax1 ・ cos (ωt- (π / 2n)) [π / n <ωt ≦ 2π / n] f (ωt) = Cmax1 ・ cos (ωt- (3π / 2n)) [2π / n <ωt ≦ 3π / n] f (ωt) = Cmax1 ・ cos (ωt- (5π / 2n))・ ・ ・ ・ ・ ・ ・ ・ [(N-1) π / n <ωt ≦ nπ / n] f (ωt) = Cmax1 ・ cos (ωt-((n-1) π / 2n)) (however, Cmax1 = 2 ・ Σ ｜ cos ((2j-1) π / 2n) ｜ (j = 0 to n / 2)) (9)

When n is an odd number, the following is obtained. [-π / 2n <ωt ≦ π / 2n] f (ωt) = Cmax2 · cos (ωt) [π / 2n <ωt ≦ 3π / 2n] f (ωt) = Cmax2 · cos (ωt- (π / n) ) [3π / 2n <ωt ≦ 5π / 2n] f (ωt) = Cmax2 ・ cos (ωt- (2π / n)) ・ ・ ・ ・ ・ ・ ・ ・ ・ ・・ ・ ・ ・ ・ [(2n-3) π / 2n <ωt ≦ (2n-1) π / 2n] f (ωt) ＝ Cmax2 ・ cos (ωt-((n-1) π / n)) (However, , Cmax2 = 1 + 2 · Σ | cos (2jπ / 2n) | (j = 1 to (n-1) / 2)) (10)

When n is an even number, the function f (ωt) is expressed by the formula (9) from ωt = π / (2n), 3π / (2n), 5π / (2n), ... The maximum value shown in ωt = 0, π / n, 2π / n, 3π / n, ...
Take the minimum value shown in 2). f (π / (2n)) = Cmax1 (11) f (π / n) = Cmax1 · cos (π / (2n)) (12)

Similarly, when n is an odd number, from the formula (10), ωt = 0, π / n, 2π / n, 3π / n, ... π / (2n), 3π / (2n), 5π / (2n), ...
Take the minimum value shown in 4). f (0) = Cmax2 (13) f (π / (2n)) = Cmax2 ・ cos (π / (2n)) (14)

When n is set to infinity, cos (π in equation (12) is
It can be seen that the term / (2n)) converges to 1 and the minimum value approaches the maximum value Cmax1. That is, when n is increased, the function f (ω
t) approaches a certain value Cmax1 regardless of time t.
The same applies to equation (14).

From the above, f (ωt) is a constant value Cmax1
And approaching Cmax2 means that equation (8) is:
It is understood that the equations shown in 5) and (16) are obtained, and only the modulated wave component and the DC component remain. That is, it can be said that the signal generated by the absolute value addition is almost close to the envelope of the modulated signal. (1 + vs) ・ Cmax1 ・ ・ ・ (15) (1 + vs) ・ Cmax2 ・ ・ ・ (16)

As can be inferred from the equations (15) and (16), when n is increased, these DC components are increased. Therefore, the digital multiplier 5 in the subsequent stage of the adder 4 uses the equation (1
The reciprocal of the maximum value shown in 1) and (13) is multiplied as a correction coefficient C shown below to correct the amplitude level of the original digital AM signal. C = 1 / Cmax1 = 1 / (2 ・ Σ ｜ cos (2j-1) π / 2n) ｜) (n is even: j = 0 to n / 2) ・ ・ ・ (17) C = 1 / Cmax2 = 1 / (1 + 2 ・ Σ ｜ cos (2j) π / 2n) ｜) (n is odd: j = 1 to (n-1) / 2) (18)

The signal corrected by the correction coefficient generator 16 and the digital multiplier 5 has a direct current component removed by a digital high-pass filter and becomes the envelope of the digital AM signal itself. After that, a digital / analog converter (hereinafter D / A
It is called a converter. ), It is converted into an analog signal and the AM demodulation signal y proportional to the envelope of the carrier wave is output to the output terminal 8.

However, since n is finite, the function f (ωt)
Varies with time, and the signal generated by the absolute value addition changes slightly from the envelope. Function f
As is clear from the equation (8), (ωt) can be considered as a signal generated by the absolute value addition in the non-modulation. The width varying from the envelope depends on the varying width of the function f (ωt), and will be considered as unmodulated for convenience sake from the following.

Therefore, the aliasing signal generated when the sampling condition is not satisfied is divided into its amplitude and frequency and derived. First, FIG. 6 shows a waveform diagram when the absolute values of the above n digital AM signals are added.
From Fig. 6, the waveform with the absolute value added is 2 times the carrier frequency fc.
It is a half-wave sine wave that repeats at an n-fold frequency fb, and this amplitude level is calculated by the correction coefficient C
Taking into account that the maximum amplitude level of the signal at the time of demodulation is normalized to 1 and the amplitude level of the half-wave sine wave is α, α is given by 19). α = (maximum value−minimum value) / correction coefficient = 1−cos (π / (2n)) (19)

When n is set to infinity, α shown in the equation (19) converges to zero, and the signal generated by the absolute value addition is detected as the envelope of the digital AM signal itself. I understand. However, it is unrealistic to add an infinite number of digital AM signals, and it is considered that α should be at a level where there is no practical problem.

On the other hand, the frequency of the half-wave sine wave generated by the absolute value addition becomes 2n times the carrier frequency, and the condition of the equation (1) cannot be satisfied. This is a form in which there is not more than one sample point in the half cycle of the half-wave sine wave. As a result, the frequency of the half-wave sine wave generated by the addition of the absolute values with respect to the sampling frequency fs becomes a frequency exceeding fs / (4n), so that a phenomenon such as aliasing occurs in the demodulated signal. Further, it is considered that the locus of this sampling point shown in FIG. 6 becomes an aliasing signal, but from FIG. 6, the amplitude of the aliasing signal becomes a value equal to or less than α shown in equation (19).

On the other hand, the half-wave sine wave period tb is the carrier wave period.
Since it is 1 / 2n times tc, tb is given by equation (20),
If the difference between the sampling time ts and the half-wave sine wave period tb is Δt, then Δt is given by equation (21). tb = 2ntc ・ ・ ・ (20) Δt = | ts-tb | ・ ・ ・ (21)

Further, in the half-wave sine wave, since one sample point appears every m cycles, the sample point m is as shown in equation (22), and Δt is accordingly given by equation (2)
It looks like 3). m = {(ts / tb) +0.5} (However, {} indicates the integer part.) (22) Δt = | ts-mtb | (23)

From the above, the period tu of the aliasing signal is
Has the form shown in equation (24), the frequency fu is
It becomes like the formula (25). tu = (tb / Δt) ts ・ ・ ・ (24) fu = fs / (tb / Δt) = 2nfcfs | (1 / fs)-(m / (2nfc)) | ・ ・ ・ (25)

Further, since the poles of this aliasing signal are inverted at the position where it crosses one cycle of the half-wave sine wave, the distortion component of the demodulated signal is also superposed with the harmonic component of the frequency shown in equation (25). It Further, the amplitude level of this harmonic is considered to be equal to or less than the value obtained by multiplying α obtained by the equation (19).

From the above, the distortion component of the demodulated signal is composed only of the aliasing signal and its harmonics, and the distortion level does not depend on the modulation factor from equations (19) and (25). Even with a modulation degree of 100%, it is possible to detect a high-quality modulated signal with a low distortion level.

[0039]

[Embodiment 1] Table 1 shows distortion levels for a carrier when the sampling frequency fs is 48.441 kHz and the carrier frequency fc is 12 kHz and the digital AM demodulation of the present invention is performed.

[0040]

[Embodiment 2] Table 2 shows distortion levels for a carrier when the sampling frequency fs is 48.441 kHz and the carrier frequency fc is 15.138 kHz and the digital AM demodulation of the present invention is performed.

[0041]

[Third Embodiment] Table 3 shows distortion levels for a carrier when the sampling frequency fs is 48.441 kHz and the carrier frequency fc is 1.892 kHz and the digital AM demodulation according to the present invention is performed.

[0042]

According to the digital AM demodulator according to the present invention, in view of the recent advances in digital technology and IC manufacturing technology, the configuration of the circuit and the like is simpler than that of the analog AM demodulator, and A modulated signal of high quality can be stably detected, and a digital AM demodulator can be realized at low cost by forming an IC.

[0043]

[Brief description of drawings]

FIG. 1 is a block diagram showing an embodiment of the present invention.

FIG. 2 is a block diagram showing a conventional technique (1).

FIG. 3 is a block diagram showing a conventional technique (2).

FIG. 4 is a block diagram showing a conventional technique (3).

FIG. 5 is a block diagram showing a conventional technique (4).

FIG. 6 is a waveform diagram showing an embodiment of the present invention.

[Explanation of symbols]

 1 Input Terminal 2 A / D Converter 3 Digital Phase Shifter 4 Digital Adder 5 Level Corrector 6 Digital High Pass Filter 7 D / A Converter 8 Output Terminal 9 PSN Matrix 10a-10c Phase Coefficient Generator 11a-11c Phase Coefficient Generator 12a to 12c Digital multiplier 13a to 13c Digital multiplier 14a to 14c Digital adder 15a to 15c Absolute value digitizer

[Table 1]

[Table 2]

[Table 3]

Claims (2)

[Claims] 1. An analog / digital converter, a phase shifter, a first phase coefficient generator, a first multiplier connected to the first phase coefficient generator, and a second phase coefficient. A generator, a second multiplier connected to the second phase coefficient generator, and a first adder for adding the output of the first multiplier and the output of the second multiplier , N PSN circuits each comprising an absolute value converter connected to the first adder,
A second AM adder for adding the respective outputs of the PSN circuits, a level corrector, a digital high-pass filter, and a digital / analog converter are connected in cascade, and a digital AM demodulator is provided. . 2. The digital AM demodulator according to claim 1, wherein
The input AM signal is converted into a first digital signal by an analog / digital converter, and a phase shifter is used to generate second and third digital signals having a phase difference of π / 2 from the first digital signal. However, each of the second digital signals output from the first phase coefficient generator has a phase difference of π / n.
n phase coefficients from cos (0) to cos ((n-1) π / n) are respectively multiplied by n first multipliers, and the third digital signal is multiplied by the second phase coefficient. The n phase coefficients from sin (0) to sin ((n-1) π / n), each of which has a phase difference of π / n, are output from the generator with n second multipliers. Multiply by
The outputs of the n first multipliers and the outputs of the n second multipliers are vector-combined by the n first adders, respectively, and the n absolute value converters respectively perform absolute vector synthesis. By taking a value, n fourth digital signals are generated, the n fourth digital signals are all added by the second adder, the level is corrected by the level corrector, and then the digital high-pass filter is used. A demodulation method for digital AM demodulation, which removes a DC component and converts it into an analog signal with a digital / analog converter to obtain a demodulated signal.