JPH0884499A - Speed sensorless vector controller for induction motor - Google Patents

Abstract

PURPOSE: To match an estimated motor speed and an actual motor speed by comparing a secondary excitation axial flux reference value outputted from a function generator with an estimated value and then correcting an estimated motor speed based on the error. CONSTITUTION: A subtractor 23 operates the difference between an estimated value of secondary excitation axial flux received from a coordinate converter 10 and a reference value of secondary excitation axial flux received from a function generator 21 thus determining a secondary flux difference. A subtractor 24 corrects a motor speed estimated at a speed adaptive mechanism 7 with a speed error and delivers the estimated motor speed thus corrected to a flux observer 4 of same dimension and an adder 17. Since a secondary flux estimated by the observer 4 of the same dimension is equalized to actual secondary flux of a motor, the estimated motor speed is substantially equalized to actual motor speed after correction even in a low speed operational region thus realizing a highly accurate speed control.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a speed sensorless vector control device for an induction motor, and is devised so that stable speed control can be performed even in operation in a low speed region.

[0002]

2. Description of the Related Art A slip frequency control type vector control method has become popular as a high performance speed control method for induction motors.
A speed sensorless vector control method for controlling this without a speed sensor is known.

FIG. 3 shows a control system of a conventional speed sensorless vector controller for an induction motor, which estimates the actual speed of the induction motor by using a speed adaptive secondary magnetic flux observer.

First, the estimation of the motor speed (rotational angular frequency ω _r ) of the induction motor 1 will be described with reference to FIG.

The voltage equation of the induction motor 1 is given by the following equation (1) when it is represented by d-q axes for observing various quantities from the synchronous rotation coordinate system rotating at the power source angular frequency (ω ₀ ).

[0006]

[Equation 1]

However, v _1d , v _1q ... Primary excitation axis voltage, primary torque axis voltage (V) i _1d , i _{1q on} synchronous rotation coordinates (dq axes) synchronous rotation coordinates (dq axes) ) Above primary excitation axis current, primary torque axis current (A) λ _2d , λ _2q ... Secondary excitation axis magnetic flux on secondary rotation coordinate (dq axis), secondary torque axis magnetic flux (Wb) ω ₀ ………… Power supply angular frequency (rad / sec) ω _r ………… Motor speed (rotational angular frequency, rad / sec) ω _s ………… Slip angular frequency (rad / sec) R ₁ , R ₂ … 1 Next, secondary resistance (Ω) L ₁ , L ₂ … Primary, secondary inductance (H) M ………… Mutual (excitation) inductance (H) Lσ ………… Equivalent leakage inductance (H) (Lσ = (L ₁ L ₂ −M ² ) / L ₂ ) s ………… Time fine molecule (d / dt) And power source angular frequency (ω ₀ ), motor speed (ω _r ),
Relationship slip angular frequency command value (omega _s ^*), and calculates the slip angular frequency command value (omega _s ^*) is represented by the following formula (2). ω ₀ = ω _r + ω _s ^* ω _s ^* = i _1b ^* / i _1a ^*・ τ ₂ …… (2) However, τ ₂ ………… Second-order time constant (τ ₂ = L ₂ / R ₂ ) Subscript (*): Indicates a command value or set value. Now, i _1d ^* = constant …………………………………………………………………………………………………………………………………………………………………………………… （………………………………………………………………………………………………………………………………………………………… of When the primary voltage command values (v _1d ^* , v _2q ^* ) of the synchronous rotation coordinate system (dq axes) are given by the following formula (4) which is a constitutive formula of the digital current controller 3, it becomes as follows.

[0008]

[Equation 2]

When the control is performed so as to satisfy the above equation (4), the primary current detection value i ₁ on the synchronous rotation coordinates (dq axes) is calculated.
^Indicates that the current flows according to the primary current command value i ₁ ^* (i _1d ^* , i _1q ^* ), and the secondary magnetic flux λ ₂ on the synchronous rotation coordinates (d-q axes).
(Λ _2d , λ _2q ) is kept at λ _2d = Mi _1d (constant), λ _2q = 0 …………………… (5).

As a result, the torque (T) of the induction motor 1
Is T = M / L ₂ · (λ _2d · i _1q −λ _2q · i _1d ) = M ² / L ₂ · (i _1d · i _1q ) ………………………… (6) , Secondary magnetic flux λ on synchronous rotation coordinates (d-q axes)
₂ (λ _2d , λ _2q ) and the secondary current i ₂ (i _2d , i _2q ) are non-interfering control vector control.

By the way, as is clear from the above equation (2).
If the speed sensor is not used, the slip angular frequency command
Value ω_s ^*Even if is set, the motor speed ω_rIs unknown
Power source angular frequency ω₀Can't decide
Is the power source angular frequency ω₀Synchronous rotation coordinate (d-
secondary magnetic flux λ on the q-axis)₂(Λ_2d, Λ_2q) Is the above formula (5)
So that the power source angular frequency ω₀To control
Similarly, vector control of non-interference control is realized by
Can be That is, the same-dimensional magnetic flux observer 4
Speed adaptive secondary magnetic flux observer consisting of
Using synchronous rotation coordinates that satisfy the above equation (5)
Secondary magnetic flux λ on (dq axis)₂(Λ_2d, Λ _2q) Estimated
The secondary magnetic flux estimated value λ₂ ^#(Λ_2d ^#, Λ_2q ^#) Based on
Motor speed ω_rEstimate (ω_r ^#)
Equation (2) (ω₀= Ω_r ^#+ Ω_s ^*) From the power angular frequency
Number ω₀And the power source angular frequency ω₀Due to digital current
Vector of decoupling control by controlling the controller 3
Control can be realized.

In the conventional speed sensorless vector control method for the induction motor in the vector control system shown in FIG. 3, in order to detect the actual speed of the induction motor 1 without using the speed sensor, the primary current of the induction motor 1 ( Phase current) i _u , _iv , i _w are detected and the 3-phase-2 phase phase number converter 1
It is assumed that the primary current detection value i ₁ (i _1a , i _1b ) on the stator coordinates (a-b axis) converted in phase number in 4 is obtained. Then, the detected primary current i ₁ and the electric motor 1 on the stator coordinate (ab axis)
The secondary magnetic flux on the stator coordinate (a-b axis) is calculated by the same-dimensional magnetic flux observer 4 that receives the next voltage command value v ₁ ^* (v _1a ^* , v _1b ^* ) and the estimated speed value ω _r ^#. Estimated value λ ₂ ^#
(Λ _2a ^# , λ _2b ^# ) and the primary current estimated value i ₁ ^# (i _1a ^# ,
i _1b ^# ) and the speed adaptation mechanism 7 estimates the primary current i ₁ ^# (i _1a ^# , i _1b ^# ) and the detected primary current i.
₁ (i _1a , i _1b ) is compared with the estimation error signal (i ₁ −i
And a motor speed estimation value (omega _r ^#) calculation estimated by the induction motor 1 speed detected by the adaptive tuning strategy represented by the following formula (7) based on ₁ ^#).

Ω _r ^# = K _p (e _ia λ _2b ^# −e _ib λ _2a ^# ) + K _i ∫ (e _ia λ _2b ^# −e _ib λ _2a ^# ) dt (7) However, e _ia = i _1a −i _1a ^# : estimation error e _ib = i _1b −i _1b ^# : estimation error K _p : velocity estimation part proportional gain K _i : velocity estimation part integration gain Note that the same-dimensional secondary magnetic flux observer 4 and velocity adaptation mechanism are used. 7
For the speed sensorless vector control method of the induction motor 1 that estimates the actual speed of the induction motor by the speed adaptive secondary magnetic flux observer consisting of and, see "Journal of the Institute of Electrical Engineers of Japan, Volume 11, No. 11, 1991" (Kubota, Ozaki, Matsuse, Nakano: "Velocity sensorless direct vector control of induction motor using adaptive secondary magnetic flux observer").

A conventional speed sensorless vector control system for estimating the actual speed of the induction motor using the speed adaptive secondary magnetic flux observer will be described below.

The operation in FIG. 3 (control system configuration)
To explain, digital current in the current controller (ACR)
In the flow controller 3, 1 on the synchronous rotation coordinate (dq axis)
Next voltage command value v₁ ^*(V_1d ^*, V_1q ^*) Is the primary power
Flow command value i₁ ^*(I_1d ^*, I _1q ^*) And the primary current detection value i
₁(I_1d, I_1q) Are equal (i_1d ^*= I_1d, I_1q ^*= I
_1q) Is a conditional expression that enables decoupling control.
It is calculated by the above equation (4).

Primary voltage finger on synchronous rotation coordinates (dq axes)
Command v₁ ^*(V_1d ^*, V_1q ^*) By the coordinate converter 9
Primary voltage command value v on the stator coordinates (ab axis)₁ ^*(V
_1a ^*, V_1b ^*2 phase to 3 phase phase number converter
The number of phases is converted by 15 and the PWM control inverter 2 has three phases.
Phase Primary voltage control command voltage v for each phase_u, V_v, V_wConversion to
The output voltage of each of the three phases of the PWM control inverter 2 is
Control. As a result, the induction motor 1 is driven at the desired motor speed.
Command value (ω_r ^*) According to the speed control. Also power
Angular frequency ω₀Synchronous rotation coordinates (d-q axes) that rotate at
Stator coordinates (ab) fixed to the stator of the induction motor 1
Used for coordinate converters 8 and 9 to convert between
Unit vector (sin θ₀, Cos θ₀) To produce
Basic phase angle of₀(Θ₀= Ω₀t) is calculated as follows.
Can be That is, the slip calculator 5
As shown in equation (2), on the synchronous rotation coordinates (d-q axes)
Primary excitation axis current command value i_1d ^*Primary torque axis current command value
i_1q ^*, And the secondary time constant τ of the induction motor 1₂(= L₂/
R₂Slip angular frequency command value (ω_s
^*) And the speed adaptive secondary magnetic flux observer (4, 7)
Estimated motor speed estimate (ω_r ^#) And obtained from
Power angular frequency (ω₀) Is the integrator 11 for calculating the basic phase angle.
The fundamental phase angle θ₀Seeking
Can be.

As described above, the conventional speed sensor
The torque control method is based on the motor speed (ω_r)
Estimated motor speed
(Ω_r ^#) And slip angular frequency command value (ω_s ^*) And addition
Power source angular frequency ω obtained by₀(Ω ₀= Ω_r ^#+ Ω_s
^*) Deviates from the true value, resulting in a digital current
Power source angular frequency ω in controller 3₀Is calculated based on
Primary voltage command value v for decoupling control₁ ^*of
For displacement and coordinate conversion in the coordinate converters 8 and 9
Basic phase angle used₀Misalignment, eventually coordinate conversion axis misalignment
The vector control of decoupling control is no longer valid.
There is a risk of injury.

This means that the induction motor 1
In the speed control of 1), a serious problem that the torque according to the torque command value cannot be obtained occurs.

Therefore, the applicant of the present application first applied for a speed sensorless vector control system capable of performing good vector control by preventing the deviation of the vector control coordinate axis which is a "lag" in the process of estimating the motor speed. (Japanese Patent Application No. 5-265976).

The technique of Japanese Patent Application No. 5-265976 filed earlier is developed based on the following knowledge.

The "lag" in the estimation process of the estimated motor speed value (ω _r ^# ) of the induction motor 1 is the slip angular frequency command value (ω _s ^* calculated by the slip calculator 5 when viewed from the induction motor 1 side ^. ) Is offset.
(Equation (2) above, ω ₀ = ω _r + ω _s ^* , ω _s = i _1q ^* /
(See i _1d ^*・ τ ₂ )

Therefore, the electric power when the vector control is established is established.
Source angular frequency ω₀The requirement that determines
Estimated value of secondary magnetic flux λ on the mark (dq axis)₂ ^#Torque axis
Minute (λ_2q ^#) Is set to zero (see the above equations (5) and (6))
Therefore, the secondary torque axis magnetic flux estimated value λ_2q ^#Integral (ω_sc
= Kω_i∫λ_2q ^#・ Dt Kω_i: Integral gain)
Slip angular frequency correction value ω_scSlip angular frequency command value
ω_s ^*To (ω_s ^*+ Ω_sc) Allows the motor speed
Degree estimate ω_r ^#Power source angular frequency ω due to “delay”₀Nozomi
Re Δω₀Can be corrected and the coordinate axis is prevented from shifting.
And complete vector control is performed. (See the following formula) ω₀+ Δω₀= Ω_r ^#+ Ω_s ^*+ Ω_sc Δω₀= Ω_sc

FIG. 4 shows an embodiment of Japanese Patent Application No. 5-265976.

In the illustrated control system, the current controller
In the digital current controller 3 in (ACR),
Primary current command value i₁ ^*Primary torque axis current command value (i
_1q ^*) And the primary excitation axis current command value (i_1d ^*) And the primary power
Flow detection value i₁Primary torque shaft current detection value (i_1q) And 1
Secondary excitation axis current detection value (i_1d) Is compared and i_1q ^*= I
_1q, And i_1d ^*= I_1dPWM control so that
On the synchronous rotation coordinate axes (dq axes) that control the inverter 2
Primary voltage command value v₁ ^*(V_1d ^*, V_1q ^*) Is above
It is calculated by the equation (4).

The primary power output from the digital current controller 3
Pressure command value v₁ ^*(V_1d ^*, V_1q ^*) In the coordinate converter 9
Primary voltage command value v on the stator coordinates (ab axis)₁ ^*
(V _1a ^*, V_1b ^*) Is converted to 2 phase-3 phase
The number of phases is converted by the converter 15, and the PWM control inverter
2 primary voltage control command voltage v for each of the three phases_u, V_v, V _w
To control the output voltage of the PWM control inverter 2.
As a result, the induction motor 1 has a desired torque axis current command value.
i_1q ^*The torque is controlled accordingly.

The actual motor speed of the induction motor 1 (ω_r)
Is the estimated value of the motor speed estimated as follows.
ω_r ^#To use. That is, the stator coordinates (a-
primary current detection value i on the (b-axis)₁(I_1a, I_1b) Primary power
Pressure command value v₁ ^*(V_1a ^*, V _1b ^*) And motor speed estimation
Value (ω_r ^#) Input same-dimensional magnetic flux observer 4
And the primary estimated by the same-dimensional magnetic flux observer 4.
Current estimated value i₁ ^#(I_1a ^#, I_1b ^#) And the estimated secondary magnetic flux
λ₂ ^#(Λ_2a ^#, Λ_2b ^#) And the detected primary current i ₁(I
_1a, I_1bBased on the above equation (7)
A speed-adaptive secondary magnetic flux observer consisting of
The motor speed estimate ω_r ^#To estimate.

Then, the power source angular frequency ω ₀ (ω ₀ = ω _r ^# +) caused by the “lag” of the estimated motor speed (ω _r ^# ) in the process of estimating the actual motor speed (ω _r ).
In order to correct the deviation of ω _s ^* ), the secondary magnetic flux estimated value λ ₂ ^# (λ _2a ^# , λ _2b ^# ) on the stator coordinate (ab axis) estimated by the same-dimensional magnetic flux observer 4 is used as the coordinate. Converter 10
Is the secondary magnetic flux λ ₂ ^# (λ
_2d ^# , λ _2q ^# ) and the secondary magnetic flux estimated value λ ₂ ^#
The secondary torque axis magnetic flux estimated value (λ _2q ^# ) is integrated by the slip angle frequency correction integrator 16 and the slip angle frequency correction value ω _sc
(Ω _sc = Kω _s ∫λ _2q ^# · dt) is obtained and added to the slip angular frequency command value ω _s ^* by the adder 17.

To add the slip angular frequency correction value ω _sc to the slip angular frequency command value ω _s ^* , the deviation of the motor speed estimated value (ω _r ^# ) is corrected by the slip angular frequency correction value ω _sc . Therefore, the basic phase angle (θ) due to the deviation of the power source angular frequency (ω ₀ ) determined by the above equation (2)
₀ ) is prevented, the coordinate conversion axis is prevented from being displaced, and accurate vector control of decoupling control is established.

[0029]

By the way, FIG. 3 and FIG.
The previously proposed technique shown in (1) has left the following problems.

Since it is difficult to detect the motor voltage of the induction motor 1 in the vector control device shown in FIGS. 3 and 4, the primary voltage command value v ₁ ^* (v _1a ^* , v _1b ^* ) is actually It is assumed that they match the motor voltage value of. Based on this assumption, the same-dimensional observer 4 uses the primary voltage command values v ₁ ^* (v _1a ^* , v _1b ^* ) and the primary current detection value i.
₁ (i _1a , i _1b ), estimated motor speed value ω _r ^# as input value, estimated primary current i ₁ ^# (i _1a ^# , i _1b ^# ), estimated secondary magnetic flux λ ₂ ^# (λ _2a ^# , Λ _2b ^# ) is estimated.

However, there is an error between the primary voltage command value v ₁ ^* and the actual motor voltage value (the cause of the error will be described below), and the error becomes larger especially in the low speed region where the output voltage is low. Due to this voltage error, an error occurs between the estimated secondary magnetic flux estimated value λ ₂ ^# and the actual secondary magnetic flux.
For this reason, an error occurs in the estimation calculation in the speed adaptation mechanism 7, and as a result, an error occurs in the motor speed estimated value ω _r ^# and the speed control becomes unstable.

Two factors of the voltage error described above will be described.

<First Voltage Error Factor> The main switching element used in the inverter part of the PWM control inverter 2 has a switching delay. In order to prevent the upper and lower arms from being short-circuited due to this delay time, a short circuit prevention period ( Dead time). Because of this dead time,
The actual motor voltage value and the primary voltage command value v ₁ ^* are different. Therefore, at present, in order to reduce this voltage error, a dead time compensating circuit for delaying the output voltage of the PWM control inverter 2 in accordance with the dead time is provided. However, complete compensation cannot be performed and the voltage error remains. Was there.

<Second Voltage Error Factor> In the PWM modulation command section of the PWM control inverter 2, the primary voltage control command voltages v _u , v _v , v _w are compared with the carrier signal (for example, triangular wave), and the PWM modulation command (base current) depending on size
Is occurring. The main switching element of the inverter section is turned on / off according to the high / low of the PWM modulation command to produce an equivalent sine wave output voltage. When the PWM modulation command section is digitally configured, the command voltages v _u , v _v , and v _w are digitally compared with the carrier signal to generate a PWM modulation command, so that the carrier resolution and the resolution of the voltage calculation value are set. Therefore, an error occurs between the actual motor voltage value and the primary voltage command value v ₁ ^* .

In view of the above-mentioned prior art, it is an object of the present invention to provide a speed sensorless vector control device for an induction motor capable of performing stable speed control even in a low speed operation region.

[0036]

[Means for Solving the Problems] First to solve the above problems
In the configuration of the present invention, the primary current command value (i₁ ^*) And the primary power
Flow detection value (i₁) And power angular frequency (ω₀) And invite
Conductor motive current decoupling control is performed and primary voltage command value
(V₁ ^*) Output current controller (3) and the current control
The primary voltage command value (v₁ ^*) Based
Power converter (2) for controlling the speed of the induction motor and the primary
Current detection value (i₁) And the primary voltage command value (v₁ ^*) And electric
Aircraft speed estimate (ω _r ^#) Respectively and input the primary current
Constant i₁ ^#And secondary magnetic flux estimated value λ₂ ^#Same dimension to estimate
Magnetic flux observer (4) and the same-dimensional magnetic flux observer
Estimated primary current value i which is the output of (4)₁ ^#And secondary magnetic flux
Constant value λ₂ ^#And the primary current detection value (i₁) For each
The estimated motor speed (ω_r ^#) Is estimated and output
Speed adaptive mechanism (7) and the primary current command value (i₁ ^*)of
Based on the excitation axis component and the torque axis component, the slip angle around the motor
Wave number command value (ω_s ^*) For calculating and outputting
(5) and the slip which is the output of the slip calculator (5)
Angular frequency command value (ω_s ^*) Motor speed estimate
(Ω_r ^#) Is added to the control input of the current controller (3)
Power source angular frequency (ω₀) Output adder (17)
Induction motor speed sensorless vector control
Output of the same-dimensional observer (4)
Estimated value of the secondary excitation axis magnetic flux (λ_2a ^#)
Is the secondary excitation axis magnetic flux estimation value (λ_2d ^#) To
A coordinate converter (10) for coordinate conversion, and the induction motor
To be approximately equal to the actual secondary excitation axis magnetic flux value in (1)
Calculated secondary excitation axis reference value (λ_0d) Is for each rotation speed
The estimated motor speed value ω_r ^#According to
Note: Secondary excitation axis magnetic flux reference value (λ_0d) Output function generator
(21) and 2 output from the coordinate converter (10)
Estimated magnetic flux of secondary excitation axis (λ_2d ^#) And the function generator (2
2) Excitation axis magnetic flux reference value (λ_0d) With
The secondary magnetic flux deviation value (Δλ), which is the deviation, is calculated, and the secondary magnetic
The coefficient (K _p), And the multiplication value (K
_p・ Calculate Δλ) and then output from the speed adaptation mechanism (7).
Estimated motor speed (ω)_r ^#) To the multiplication value (K_p
・ Modified motor speed estimated value by subtracting Δλ)
(Ω_r ^#') And calculate this modified motor speed estimate
(Ω_r ^#') Is the same dimension observer (4) and adder (1
7) for sending to the correction calculation unit.

In the configuration of the second aspect of the present invention, in the correction calculation section, the electric motor 2 is operated from the time when the induction motor (1) is started.
In the following the delay time following a magnetic flux is established has elapsed, the operation of outputting a motor speed estimated value (ω _r ^#) subtraction and correct motor speed estimate a multiplication value (K _p · Δλ) from the (ω _r ^# ') It is characterized by starting.

[0038]

In the speed sensorless vector control device, an error occurs between the primary voltage command value v ₁ ^* and the actual motor voltage value in the low speed operation region, and the estimated secondary magnetic flux estimated value λ ₂ ^{# is} calculated. An error occurs between the actual secondary magnetic flux and the actual secondary magnetic flux.
In the present invention, in accordance with the difference between the actual secondary magnetic flux and the secondary magnetic flux estimated value lambda ₂ ^#, by correcting the motor speed estimated value omega _r ^#, the actual motor speed the motor speed estimated value omega _r ^# To match.

[0039]

Embodiments of the present invention will be described below in detail with reference to the drawings. The same parts as those of the conventional technique are designated by the same reference numerals, and duplicate description will be omitted.

FIG. 1 shows an embodiment of the present invention. In the present embodiment, the slip angle frequency correcting integrator 16 is removed from the conventional device shown in FIG. 4, and a function generator 21 and a multiplier 22 are newly added.
And subtractors 23 and 24 are added.

As shown in FIG. 1, the coordinate converter 10 estimates the secondary magnetic flux estimated values λ _2a ^# and λ _2b ^# on the fixed coordinates, which are estimated and calculated by the same-dimensional magnetic flux observer 4, to estimate the secondary magnetic flux on the rotating coordinates. Coordinates are converted to the values λ _2d ^# and λ _2q ^# . Of these, the secondary excitation axis magnetic flux estimated value λ _2d ^# is input to the subtractor 23.

On the other hand, as shown in FIG. 2, the function generator 21 causes the secondary excitation axis magnetic flux reference value λ _0d (value on the rotational coordinate) of the amplitude value determined in advance according to the motor speed estimated value ω _r ^#. ) Is output. This secondary excitation shaft magnetic flux reference value λ _0d is the induction motor 1
Is calculated based on the motor constant, the exciting current, and the exciting inductance, and the secondary excitation axis magnetic flux reference value λ _{0d at} each rotation speed is the actual secondary excitation axis magnetic flux value (on the rotation coordinate). Value)). Therefore, the secondary excitation shaft magnetic flux reference value λ _0d is the induction motor 1
The rotation speed of is constant from 0 to N ₁ , but when it exceeds N ₁ , it gradually decreases as the rotation speed increases. In addition,
The function generator 21 has a reference value λ corresponding to the motor rotation speed.
Since only need to prepare _0d, this function can be easily calculated.

The subtractor 23 sends the data from the coordinate converter 10.
Estimated value of the secondary excitation axis magnetic flux λ_2d ^#And the function generator 2
Secondary excitation axis magnetic flux reference value λ sent from 1_0dDifference from
The secondary magnetic flux deviation value Δλ is calculated. This secondary magnetic flux
The deviation value Δλ is the secondary excitation shaft magnetic flux estimated value λ_2d ^#And the actual 2
It corresponds to the difference with the magnetic flux value of the secondary excitation axis. Generally secondary
Excitation axis flux estimate λ_2d ^#Is the actual secondary excitation axis magnetic flux value
Tends to be larger than.

The multiplier 22 multiplies the secondary magnetic flux deviation value Δλ by a proportional coefficient K _p (a number of 1 or less) and outputs a multiplication value K _p · Δλ. By multiplying by the proportionality coefficient K _p , the multiplication value K
_p · Δλ is a value indicating the motor speed corresponding to the secondary magnetic flux deviation value Δλ.

The subtracter 24 subtracts the multiplication value K _p .Δλ from the estimated motor speed value ω _r ^# estimated and calculated by the speed adaptation mechanism 7 and outputs the corrected estimated motor speed value ω _r ^# '. This corrected motor speed estimated value ω _r ^# 'is a value obtained by correcting the speed error caused by the deviation between the secondary excitation axis magnetic flux estimated value λ _2d ^# and the actual secondary excitation axis magnetic flux. The corrected motor speed estimated value ω _r ^# 'corrected for the speed error is input to the same-dimensional magnetic flux observer 4 and the adder 17. Therefore, the secondary magnetic flux estimated value λ ₂ ^# estimated and calculated by the same-dimensional magnetic flux observer 4 becomes equal to the actual motor secondary magnetic flux, and the corrected motor speed estimated value ω _r ^# 'estimated and calculated even in the low speed operation region and the actual motor. The speed becomes almost equal, and accurate speed control can be performed.

The subtractor 24 starts the above-described subtraction calculation after the delay time for establishing the secondary magnetic flux of the motor elapses from the time when the induction motor 1 is started. this is,
The correction calculation of the estimated speed at the motor start is performed by the motor 2
This is because it is difficult to understand the state until the next magnetic flux is established, and it is considered that correct correction calculation may not be obtained.

[0047]

As described above in detail with the embodiments, in the first aspect of the present invention, the secondary excitation axis magnetic flux estimated value λ estimated and calculated as the secondary excitation axis magnetic flux reference value λ _0d output from the function generator. comparing the magnitude of the _2d ^#, the error value, is regarded as a value corresponding to the estimated computed motor speed estimated value omega _r ^# error, the motor speed estimated value omega _r Fixed ^# operation and correct motor speed Output the estimated value ω _r ^# '. Since the corrected motor speed estimated value ω _r ^# 'is input to the same-dimensional magnetic flux observer 4, the speed estimation ability is improved even in the low speed operation region.

Further, in the second aspect of the present invention, since the correction calculation of the motor speed is performed after the secondary magnetic flux of the motor is established, it is not necessary to perform the unnecessary correction calculation at the time of starting.

[Brief description of drawings]

FIG. 1 is a configuration diagram showing an embodiment of the present invention.

FIG. 2 is a characteristic diagram showing an output characteristic of a function generator.

FIG. 3 is a configuration diagram showing a conventional technique.

FIG. 4 is a configuration diagram showing a conventional technique.

[Explanation of symbols]

1 Induction motor 2 PWM control inverter 3 Digital current controller 4 Same-dimensional magnetic flux observer 5 Slip calculator 6 Speed controller 7 Speed adaptation mechanism 8 Coordinate converter 9 Coordinate converter 10 Coordinate converter 11 Basic phase angle integrator 16 Slip angular frequency correction integrator 17 Adder 21 Function generator 22 Multiplier 23, 24 Subtractor ω _s Slip angular frequency ω _s ^* Slip angular frequency command value ω _sc Slip angular frequency correction value ω ₀ Power angular frequency ω _r Motor speed ω _r ^* motor speed command value ω _r ^# motor speed estimated value ω _r ^# 'modify motor speed estimate v ₁ 1 primary voltage v _1d 1 primary excitation axis voltage v _1q 1-order torque-axis voltage v ₁ ^* ₁ primary voltage directive The value v _1d ^*, v _1q ^* 1 primary excitation axis voltage command value v _1a ^*, v _1b ^* 1 primary torque-axis voltage command value _{i u, i v, i w} 1 primary current i ₁ 1 primary current detection value i _1d, i _1a 1 primary excitation axis current detection value _1q, i _1b 1 primary torque axis current detection value i ₁ ^* 1 primary current command value i _1d ^* 1 primary excitation axis current command value i _1q ^* 1 primary torque axis current command value i ₁ ^# 1 primary current estimated value i _1a ^# Primary excitation shaft current estimated value i _1b ^# Primary torque shaft current estimated value λ ₂ Secondary magnetic flux λ _2d , λ _2a Secondary excitation shaft magnetic flux λ _2q , λ _2b Secondary torque shaft magnetic flux λ ₂ ^# Secondary magnetic flux estimated value λ _2d ^# , λ _2a ^# Secondary excitation axis flux estimation value λ _2q ^# , λ _2b ^# Secondary torque axis flux estimation value λ _0d Secondary excitation axis flux reference value Δλ Secondary flux deviation value v _u , v _v , v _w Primary voltage control command voltage θ ₀ Basic phase angle

Claims (2)

[Claims] 1. A primary current command value (i₁ ^*) And primary current detection
Outgoing price (i₁) And power angular frequency (ω₀) And enter induction
The primary voltage command value (v₁
^*), And a primary voltage command value (v) that is the output of the current controller (3).
₁ ^*), A power converter that controls the speed of the induction motor
(2) and the primary current detection value (i₁) And the primary voltage command value (v₁ ^*)When
Estimated motor speed (ω _r ^#) For each
Flow estimate i₁ ^#And secondary magnetic flux estimated value λ₂ ^#Estimate the same
Dimensional magnetic flux observer (4) and the primary power output from the same dimensional magnetic flux observer (4)
Flow estimate i₁ ^#And secondary magnetic flux estimated value λ₂ ^#And primary current detection
Outgoing price (i₁), And estimate the motor speed (ω
_r ^#) And a speed adaptation mechanism (7) that estimates and outputs the primary current command value (i₁ ^*) Excitation axis component and torque axis component
Based on the slip angular frequency command value (ω_s ^*Play)
A slip calculator (5) that calculates and outputs, and a slip angular frequency finger that is an output of the slip calculator (5).
Command value (ω_s ^*) To the estimated motor speed (ω_r ^#) Is added
Power source angular frequency which is the control input of the current controller (3)
(Ω₀) Output adder (17), and an induction motor speed sensorless vector control device comprising:
Where the stator coordinates are the output of the same-dimensional observer (4)
Upper secondary excitation axis magnetic flux estimated value (λ_2a ^#) On the synchronous rotation coordinate
Secondary excitation axis magnetic flux estimated value (λ_2d ^#) Coordinate to convert
The converter (10) and the induction motor (1) actual secondary excitation shaft magnetic flux value is almost
Secondary excitation axis reference value (λ_0d)But
It is set for each rotation speed, and the estimated motor speed ω
_r ^#According to the secondary excitation shaft magnetic flux reference value (λ_0d) Is output
Function generator (21) for performing the secondary excitation axis magnetic flux output from the coordinate converter (10)
Estimated value (λ_2d ^#) And output from the function generator (21)
Secondary excitation axis magnetic flux reference value (λ_0d) And 2
The secondary magnetic flux deviation value (Δλ) is calculated, and this secondary magnetic flux deviation value (Δ
The coefficient (K _p), And the multiplication value (K_p・ Δλ)
Electric power output from the speed adaptation mechanism (7)
Aircraft speed estimate (ω_r ^#) To the multiplication value (K_p・ Δλ) reduced
The corrected motor speed estimated value (ω_r ^#')
Therefore, this modified motor speed estimate (ω_r ^#')
Correction calculation unit to send to buzzer (4) and adder (17)
And a speed sensorless induction motor.
Vector controller. 2. The induction motor (1) in the correction calculator.
After the delay time when the secondary magnetic flux of the motor is established has elapsed from the time when the motor started, the estimated motor speed (ω _r ^# )
The speed sensorless vector control device for an induction motor according to claim 1, wherein the operation for outputting the corrected motor speed estimated value (ω _r ^# ') by subtracting the multiplication value (K _p · Δλ) from is started.