JPH0833382A - Driving apparatus of brushless motor - Google Patents

Abstract

PURPOSE:To obtain a driving apparatus which smoothly changes over a driving current given to driving coil and reduces vibrations when a motor is driven by a method wherein a counter electromotive-force detector which detects a counter electromotive force generated in the driving coil so as to be waveform- shaped is provided with a mask-signal generator which generates a signal used to mask a phase. CONSTITUTION:Voltage waveforms U, V, W and output signals P1 to P6 from a mask-signal generator 12 are input to a counter-electromotive-force detector 10, and U1, V1 and W1 are output so as to be inputted to a trapezoidal-wave current composition device 11. The trapezoidal-wave current composition device 11 generates current outputs IPL1 to IPL3, IPU1 to IPU3, and a trapezoidal motor-position signal is obtained. Then, a current distribution circuit 13 sequentially supplies a driving current to driving coils 1 to 3, and a motor is turned. Thereby, since only the signal of the counter electromotive force according to the rotation of the motor can be detected, the driving current which drives the driving coils can be changed over smoothly, and it is possible to reduce vibrations when the motor is turned.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a brushless motor drive device for smoothly switching a drive current.

[0002]

2. Description of the Related Art In recent years, office automation equipment using floppy disks, hard disks, etc., video tape recorders,
Brushless motors are used in drive devices for consumer equipment such as headphone tape recorders. These brushless motors generally use a two-phase or three-phase half-wave drive system or a full-wave drive system, but this type of brushless motor uses a position detection element such as a hall element for detecting the position of the rotor. Has been.

On the other hand, attempts have conventionally been made to reduce the number of position detecting elements. For example, the energization state of each motor drive coil (hereinafter referred to as drive coil) is switched by the output signal of a self-propelled three-phase multivibrator. , A method using a drive circuit that switches the energization state to each drive coil by utilizing the power generation waveform that appears in the drive coil among the three-phase drive coils that have been de-energized after the rotor has rotated (Japanese Patent Laid-Open No. 5-58200).
0-72113).

[0004]

However, in the above-mentioned conventional structure, the current is rapidly switched to each drive coil, so that there is a problem that unnecessary vibration or noise or electric noise due to spike pulse is generated. Was there. In order to solve such a problem, there is a means for connecting a capacitor to a drive coil in parallel to dull the drive current, but it is necessary to make the capacitor have a large capacity, and it is almost not built in the IC. It is impossible. However,
When the capacitor is externally attached to the external terminal of the IC, it is necessary to provide a capacitor for each drive coil of each phase, which causes a problem of an increase in mounting area as the number of peripheral components increases.

Further, in the above-mentioned conventional structure, since the energization switching signal is created from the back electromotive force of the drive coil, spike noise generated at the time of energization switching of the drive coil is mixed into the energization switching signal to cause malfunction. Since the energization switching signal is generated irrespective of the phase of the motor at the time of starting, there is a problem that the detection signal and the energization switching signal are not synchronized and the starting characteristics are not stable.

The present invention solves the above problems by smoothing the switching of the drive current for driving each drive coil,
An object of the present invention is to provide a drive device for a brushless motor that reduces vibration when the motor is driven.

[0007]

To achieve this object, a brushless motor drive apparatus according to the present invention provides a plurality of drive transistors for supplying current to a plurality of phases of motor drive coils and a motor torque command signal. Generates torque command signal generation circuit, current distribution circuit that sequentially supplies current to drive transistors by output signal of torque command signal generation circuit, trapezoidal wave current synthesizer that outputs trapezoidal wave current to current distribution circuit, and motor drive It is equipped with a back electromotive force detector that detects the back electromotive force generated in the coil and shapes the waveform, and a mask signal generator that outputs a signal that masks a part of the signal waveform to the back electromotive force detector. The generation circuit has a clamp function.

[0008]

With this configuration, since the mask signal generator is provided, the surge pulse superimposed on the waveform of the back electromotive force can be ignored and only the back electromotive force signal according to the rotation of the motor can be detected. Switching of the drive current for driving the coil becomes smooth, and vibration during motor rotation can be reduced. In particular, the effect of reducing the vibration of the motor at the time of starting the motor, which is apt to generate spike noise, is great.

[0009]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS A brushless motor drive device according to an embodiment of the present invention will be described below with reference to the drawings.

FIG. 1 is a circuit configuration diagram of a brushless motor drive device, showing an example applied to a three-phase full-wave current drive motor. In FIG. 1, 1, 2, 3 are drive coils, 4, 5, 6 are discharge side output transistors, 7,
Reference numerals 8 and 9 are output transistors on the suction side, 10 is a back electromotive force detector for detecting back electromotive force generated in the drive coil, and 11
Is a trapezoidal wave current synthesizer that generates a trapezoidal wave current that smoothes rising and falling by a signal obtained by the counter electromotive force generator 10, and 12 is a mask that generates a mask signal to be input to the counter electromotive force detector 10. The signal generator 13 receives the trapezoidal wave current output from the trapezoidal wave current synthesizer 11 as an input and determines a base current supplied to the discharge side output transistors 4, 5, 6 and the suction side output transistors 7, 8, 9 A distribution circuit, 14 is a torque command signal generating circuit for generating a motor torque command signal, and 15 is a resistor for current detection. Reference numerals 22, 23, 24, and 25 denote a charge / discharge control circuit, a charge / discharge circuit, a voltage / current conversion circuit, and a trapezoidal wave current switching circuit, which constitute the trapezoidal wave current combiner 11. U1, V1, and W1 are output signals from the back electromotive force detector, and P1 to P6 are output signals from the mask signal generator 12.

The operation of the brushless motor driving device constructed as described above will be described below with reference to FIG. 1 and FIG.

FIG. 2 is a diagram showing signal waveforms of various parts during steady rotation in the embodiment of FIG. In FIG. 2, U, V, and W are voltage waveforms of the back electromotive force of the drive coils 1, 2, and 3, and N is a voltage waveform of the drive coils 1, 2, and 3 at the neutral point.

First, the voltage waveforms U, V, W, N and the output signals P1 to P6 of the mask signal generator 12 are applied to the counter electromotive force detector 1.
Input to 0, process, and output U1, V1, W1. Output signals U1, V1, W1 from the counter electromotive force detector 10
Is input to the trapezoidal wave current combiner 11. The trapezoidal wave current combiner 11 has current outputs IPL1 to IPL3 and IPU1 to IP.
Output U3. This trapezoidal wave current output IPL1 to IPL
3, IPU1 to IPU3 are input to the current distribution circuit 13, where the current distribution ratio is set. By inputting the output current from the current distribution circuit 13 controlled by the torque command signal generation circuit 14 to the suction side output transistors 7, 8, 9 and the discharge side output transistors 4, 5, 6, the drive coils 1, 2, Drive coil current conduction waveforms IU, IV, IW having a slope for smoothing the rising and falling of the current conduction waveform 3 are formed.

Next, a specific example of the back electromotive force detector 10 in FIG. 1 will be described with reference to FIG. FIG. 3 is a specific circuit configuration diagram of the counter electromotive force detector, and FIG. 4 is a signal waveform diagram of each part of the specific example shown in FIG. Although the voltage waveform U of the counter electromotive force is shown in FIG. 4, the same operation is performed for both V and W.

As shown in FIG. 4, a spike pulse is generated in the voltage waveform U when the current is switched. The voltage waveform U and the neutral point N are input to the comparator 16 to obtain the output UO. Noise pulses detected by the UO at the phase where the spike pulse is generated are N1 to N4. If the output UO is directly input to the trapezoidal wave current combiner 11 as a motor position signal, the motor drive current and the motor phase do not match and the motor does not operate normally. Therefore, the output UO is input to the NAND circuit 17 and the inverter (hereinafter referred to as INV circuit) 18. The mask signal P4 waits high before and after the phase in which the output UO changes from low to high, and when the UO becomes high, the NAND circuit 17
Output goes low. Then, the input of the NAND circuit 19 forming the RS flip-flop becomes low, and NA
The output U1 of the ND circuit 19 becomes high. Next, the mask signal P1 waits high before and after the phase where UO changes from high to low. When UO becomes low, the output of INV18 becomes high and the output of the NAND circuit 20 becomes low. The input of the NAND circuit 21 becomes low, and the output U1 of the NAND circuit 19 becomes low. And the noise pulse N
U1 does not go low because the mask signal P1 is low in the phases generated by 1 and N2. Similarly, U1 does not go high because the mask signal P4 is low in the phase where N3 and N4 are generated. In this way, the spike signal generated in the counter electromotive force by the mask signal can be removed, and the correct phase of the counter electromotive force can be detected.

In the motor drive apparatus of the above embodiment, the counter electromotive forces U, V, W and the mask signals P1 to P6 are used to convert the counter electromotive forces U, V, W into U1, V1, W1.
1, output current IPU1 controlled by the phase of W1
3, IPL1 to 3 are generated, and trapezoidal motor position signals of IU, IV, and IW are obtained. Then, the current distribution circuit 13 sequentially supplies the drive current to the drive coils 1, 2, and 3,
The motor rotates. Therefore, according to the present embodiment, it is possible to configure a full-wave drive type motor drive device that allows a current flowing in the motor to flow in both directions without providing a motor position detection element such as a hall element.

Next, a first specific example of the trapezoidal wave current combiner 11 in FIG. 1 will be described with reference to FIGS. 5 and 6.

FIG. 5 is a circuit configuration diagram of a first specific example of the trapezoidal wave current combiner, and FIG. 6 is an operation waveform diagram of each part of the circuit configuration of FIG. As shown in FIG. 5, the trapezoidal wave current combiner 11 includes a charge / discharge control circuit 22, a charge / discharge circuit 23, a voltage / current conversion circuit 24, and a trapezoidal wave current switching circuit 25. Outputs U1, V1 and W1 of the back electromotive force detector 10 input to the trapezoidal wave current combiner 11 are input to the charge / discharge control circuit 22 and processed as follows. A charge / discharge control signal CHG1 is obtained by an exclusive OR of U1 and W1. Similarly, U
Charge or discharge control signal C by exclusive OR of 1 and V1
The charge / discharge control signal CHG3 is obtained by HG2 and the exclusive OR of V1 and W1. The charge / discharge control signals CHG1, CHG2, CHG3 are input to the charge / discharge circuit 23. The charge / discharge circuit 23 includes individual charge / discharge circuits 26, 2
7 and 28, an example of which is shown in FIG. The charge / discharge circuit 26 includes a charge current source Io, a discharge current source 2Io, and a SW.
1, a capacitor C1 having a capacitance C, pnp transistors Q1 and Q2 connected to the charging current source Io and forming a current mirror circuit, and an npn transistor Q forming a current mirror circuit connected to the discharge current source 2Io.
3 and Q4. The charging / discharging circuit 27, 2
8 also has a circuit configuration similar to that of the charge / discharge circuit 26.

The charging / discharging circuit 26 operates as follows. When the charge / discharge control signal CHG1 is low, SW1 is turned off and the capacitor C1 is charged with the constant current Io. When the charge / discharge control signal CHG1 is high, SW1 is turned on and the capacitor C1 is discharged with the constant current Io. The voltage generated in the capacitor C1 becomes like the slope voltage VSL1. Charge / discharge circuits 27 and 28 are also CHG2 and CH, respectively.
The same operation is performed for G3, and the slope voltage VSL2, V
Generate SL3.

Output signals VSL1, VS of the charge / discharge circuit 23
L2 and VSL3 are input to the voltage / current conversion circuit 24 and generate output currents ISL1, ISL2 and ISL3. The output currents ISL1, ISL2, ISL3 are input to the trapezoidal wave current switching circuit 25 and processed as follows. ISL
The sum of 1 and ISL3 to IPU1 when U1 is high,
When U1 is low, it is distributed to IPL1 and IS
Add the sum of L1 and ISL2 to IPU2 when V1 is high, V
Distribute to IPL2 when 1 is low, ISL2 and ISL3
Is added to IPU3 when W1 is high and IPL when it is low
Distribute into 3.

As described above, the output currents IPU1-3, I from the output signals U1, V1, W1 of the back electromotive force detector 10 are output.
PL1 to PL3 can be obtained. Further, according to the present embodiment, the rises of the output currents IPU1 to 3 and IPL1 to 3 are determined by the charge / discharge control signals CHG1 to CHG3, so that they do not rise sharply. Further, the timing of the fall of the output currents IPU1 to 3 and IPL1 to 3 causes a phase difference with the outputs U1, V1 and W1 of the counter electromotive force detector 10 due to variations in the charging current or the discharging current of the charging / discharging circuit 23. Usually, these circuits are accurately manufactured in the same integrated circuit (hereinafter referred to as an IC), so that variations in charging current and discharging current do not pose a problem.
Further, the output currents IPU1 to 3 and IPL1 to 3 rise,
Since the falling phase is not affected by the capacitors C of the charging / discharging circuits 26, 27, 28, variations in the capacitors C of the charging / discharging circuits 26, 27, 28, which are usually external to the IC, can be ignored.

As described above, in the present embodiment, the switching of the currents supplied to the drive coils 1, 2 and 3 is performed very smoothly, the spike voltage accompanying the switching of the drive currents is reduced, and the motor is driven. Vibration and noise can be reduced.

Next, a second specific example of the trapezoidal wave current combiner 11 in FIG. 1 will be described with reference to FIGS. 12 and 13. 12 is a circuit configuration diagram of a second specific example of the trapezoidal wave current combiner, and FIG. 13 is a signal waveform diagram of each part of FIG.

As shown in FIG. 12, the trapezoidal wave current combiner 1 is used.
1 is a charge / discharge control circuit 50, a charge / discharge circuit 51, a voltage / current conversion circuit 52, a current shaping control circuit 55, and a current shaping circuit 5.
3 and a trapezoidal wave current switching circuit 54.
Outputs U1, V1, W1 of the back electromotive force detector 10 input to the trapezoidal wave current combiner are input to the charge / discharge control circuit 50,
It is processed as follows. The outputs U1 and W1 are input to the AND circuit 56, the outputs U1 and V1 are input to the AND circuit 57, and the outputs V1 and W1 are input to the AND circuit 58.
When the outputs of the circuits 56, 57 and 58 are input to the NOR circuit 59, the charge / discharge control signal CSIG shown in FIG. 13 is obtained from the output of the OR circuit 59. When the output of the NOR circuit 59 is input to the INV circuit 60, the charge / discharge signal CSIGB is obtained from the output of the INV circuit 60. Charge / discharge control signals CSIG, CS
The IGB is input to the charge / discharge circuit 51. Charge / discharge circuit 51
Is a current source 61, 62, 63, 64, a switch SW2,
It is composed of SW3 and capacitors 65 and 66. The current sources 61 and 63 are the current Io, and the current sources 62 and 64 are the current 2
It is a current source of I0. The capacitance of capacitors 65 and 66 is C
S1 and CS2, and CS1 = CS2. SW2
SW3 is on when the input signal is high.

The charging / discharging circuit 51 operates as follows.
The charge / discharge control signal CSIG is input to SW2. When the charge / discharge control signal CSIG is high, SW2 is turned on and the capacitor 65 is discharged with the constant current Io. Charge / discharge control signal C
When SIG is low, SW2 turns off and capacitor 65
Is charged with a constant current Io. The voltage generated in the capacitor 65 has a slope voltage VS1 shown in FIG.
The charge / discharge control signal CSIGB is input to SW3. When the charge / discharge control signal CSIGB is high, SW3 is turned on and the capacitor 66 is discharged with the constant current Io. When the charge / discharge control signal CSIGB is low, SW3 is turned off and the capacitor 66 is charged with the constant current Io. The voltage generated in the capacitor 66 becomes the slope voltage VS2 shown in FIG.

Output signals VS1 and VS2 of the charge / discharge circuit 51
Is input to the voltage-current conversion circuit 52, and the output current IS1,
IS2 is input to the current shaping circuit 53.

The current shaping control circuit 55 is the EX-OR circuit 7
7, 78, 79, AND circuits 80, 81, 82, 83,
It is composed of 84 and 85. The output signals U1, V1, W1 of the counter electromotive force detector 10 are input to the current shaping control circuit 55. The EX-OR circuit 77 outputs the output signals V1 and W1 to the EX
The -OR circuit 78 receives the output signals U1 and W1, and the EX-OR circuit 79 receives the output signals U1 and V1. AND
The circuit 80 outputs the output signal of the EX-OR circuit 77 and the output signal V.
1 and the AND circuit 81 is an EX-OR circuit 77.
Of the AND circuit 8 and the output signal W1
2 receives the output signal of the EX-OR circuit 78 and the output signal W1 as input, the AND circuit 83 receives the output signal of the EX-OR circuit 78 and the output signal U1 as input, and the AND circuit 84 outputs E.
The output signal of the X-OR circuit 79 and the output signal U1 are input, and the AND circuit 85 receives the output signal of the EX-OR circuit 79 and the output signal V1. AND circuits 80, 81,
The output signals of 82, 83, 84, 85 are U shown in FIG.
S1, US2, VS1, VS2, WS1, WS2, which are input to the current forming circuit 53.

The current forming circuit 53 is a transistor 69, 70, 71, 7 forming a first current mirror circuit.
2, transistors 73, 74, 75, 76 forming the second current mirror circuit, and switches SW4, S
It is composed of W5, SW6, SW7, SW8 and SW9.
The switch outputs a current to the GND side when the input signal is high and to the trapezoidal wave current switching circuit 54 when the input signal is low. The output current IS1 of the voltage-current conversion circuit 67 is supplied to the transistor 69 which constitutes the first current mirror circuit. The transistor 7 has a current substantially equal to the output current IS1.
It is output from 0, 71, 72. Similarly, when the output current IS2 of the voltage-current conversion circuit 68 is supplied to the transistor 73, a current substantially equal to the output current IS1 is generated by the transistor 7.
It is output from 4,75,76. US1 for SW4,
US2 is input to SW5, VS1 is input to SW6, VS2 is input to SW7, WS1 is input to SW8, and WS2 is input to SW9. The output current of the transistor 70 is the output signal US1.
Is output to GND when is high, and the output current IUS1 is output to the trapezoidal wave current switching circuit 54 when the output signal US1 is low. The transistor 74 outputs the output current IUS2 to the trapezoidal wave current switching circuit 54. This output current I
US1 and IUS2 are shown in FIG. Similarly, the output currents of the transistors 71, 72, 75, and 76 are SW, respectively.
Signals VS1 and W input to 5, SW6, SW8 and SW9
Output current IVS1, controlled by S1, VS2 and WS2
Input to the trapezoidal wave current switching circuit 54 as IWS1, IVS2, and IWS2.

The trapezoidal wave current switching circuit 54 inputs the output signals U1, V1 and W1 of the counter electromotive force detector 10. In the trapezoidal wave current switching circuit 54, the output current IUS1,
IUS2 sum to UPL1 when U1 is low, U1
When is high, it switches to IPU1. Also, the sum of the output currents IVS1 and IVS2 is IP when V1 is low.
L2 is switched to IPU2 when V1 is high, and similarly, the sum of output currents IWS1 and IWS2 is switched to IPL3 when W1 is low and to IPU3 when W1 is high.

As described above, output currents IPU1-3, I from the output signals U1, V1, W1 of the counter electromotive force detector 10 are output.
PL1 to PL3 can be obtained. Trapezoidal wave current synthesizer 11
The charging / discharging circuit 23 of No. 1 was normally composed of three capacitors which are external to the IC, but the charging / discharging circuit 5 of this embodiment is not shown.
In the case of 1, the number of external capacitors is 2, and the number of parts can be reduced. Further, as in the above-described embodiment, the output currents IPU1 to 3 and IPL1 to 3 are set to rise and fall timings according to the output signals U1, V1 and W1.
Output currents IPU1-3, IP at the timing of switching
Since L1 to L0 are 0, they do not switch abruptly.

As described above, in the present embodiment, the switching of the currents supplied to the drive coils 1, 2 and 3 is performed very smoothly, so the spike voltage associated with the switching is reduced, and the brushless motor with less vibration and noise is generated. Can be realized.

Next, a first specific example of the mask signal generator 12 will be described with reference to FIGS. 7 and 8.
FIG. 7 is a circuit configuration diagram of a first specific example of the mask signal generator,
FIG. 8 is an operation waveform diagram of FIG.

First, the output signal VSL of the charge / discharge circuit 23.
1, VSL2 and VSL3 are input to the mask signal generator 12. In the mask signal generator 12, the comparators 29, 30,
31 compares VSL1 and VSL3 to C31
CSL is compared with VSL1 and VSL2.
2 is compared with VSL3 to obtain C23. Further AND
In the circuits 32, 33 and 34, CLK1 and C2 are obtained by the logical product of the charge / discharge control signals CHG1, CHG2 and CHG3 obtained by the charge / discharge control circuit 22 and C31, C12 and C23.
Obtain LK2 and CLK3. These signals are applied to the OR circuit 35.
To obtain CLK. When this signal CLK is used as a clock signal for the hexadecimal ring counter 36 to operate at the rising edge, P1, P2, P3, P4, P5 and P6 are obtained. If the counter electromotive force detector 10 is configured so as to detect the fall of the counter electromotive force U1 only when P1 of the hexadecimal ring counter 36 stands by at high, the phase of 30 degrees before and after the fall of U1 is detected. Only detect. Similarly, when P2 is high and W1 rises,
3 is high to detect the fall of V1, P4 is high to detect the rise of U1, P5 is high to detect the fall of W1, and P6 is high to detect the rise of V1. In this way, it is possible to generate a mask signal that can be detected every 30 degrees before and after the rise and fall of the back electromotive force. Moreover, since the order of signals that can be detected is determined, malfunctions at startup are prevented and startup characteristics are improved.
It is possible to prevent abnormal operation.

FIG. 9 shows an operation waveform diagram when the rotation speed of the motor is slow. Also in this case, the back electromotive force is detected only by the phase of 30 degrees before and after the rise and fall of the back electromotive force, and the phase relationship between the mask signal and the back electromotive force does not change even if the motor rotation speed is small.

As described above, according to this embodiment, a mask signal having a constant phase relationship can be created without adjustment even when the rotation speed of the motor changes. Further, since the drive current and the mask signal are created by the charge / discharge signal, no special delay circuit for creating the mask signal is required.

Next, a second specific example of the mask signal generator will be described with reference to the drawings.

The second specific example of the mask signal generator makes it possible to arbitrarily determine the phase of the mask signal with respect to the rise and fall of the back electromotive force.

FIG. 10 is a circuit configuration diagram of a second specific example of the mask signal generator, and FIG. 11 is an operation waveform diagram of each part of FIG. In the mask signal generator shown in FIG. 7, the phase of the mask signal is always detected only 30 degrees before and after the rise and fall of the back electromotive force, and the phase of the mask signal is fixed, but this embodiment arbitrarily reverses the phase. The phase with respect to the rise and fall of the electromotive force can be set. For the sake of simplicity, the number of rotations of the motor is constant, and the charging current Io and the discharging current 2Io of the capacitors C of the charging / discharging circuits 27, 28, 29 forming the trapezoidal wave current combiner 11 shown in FIG. It will be explained on the condition that it is not present. At this time, the maximum values of the outputs VSL1, VSL2, VSL3 from the charge / discharge circuit 23 also become constant, and this is referred to as VSLP. Further, the sum of the currents converted by the voltage-current converter 24 does not change depending on the phase, and if ISL1 + ISL2 + ISL3 = Io, then Io is constant.

When Io × R = Vref, in the comparators 37, 38 and 39, Cref1 is obtained from Vref and VSL1 and Cref is obtained from Vref and VSL2.
Cref3 is output from ref2 and Vref and VSL3. When the value of the resistor R40 is set so that the reference voltage Vref becomes Vref = Io × R = VSLP / 2, Cref1
The high phase is 30 degrees before and after the rise and fall of U1. When the resistance R40 is set in this way, the phase of the mask signal is the same as that of the embodiment shown in the circuit configuration of FIG. 7 and the signal waveform diagram of FIG.
It becomes possible to detect the rising and falling of 1 and W1 in the phase of 30 degrees before and after.

Next, the operation when the value of the resistor 40 is changed will be described with reference to FIG. When the resistance 40 is set to 2R / 3, the reference voltage becomes VSLP / 3, and at this time, the phase in which Cref1 is high is 20 degrees before the rise and fall of U1, and 40 degrees after the fall. The same operation is performed thereafter, and the phase of the mask signal is the output signals U1, V of the counter electromotive force detector 10.
It is possible to detect the leading and trailing edges of W1 and W1 at a phase of 20 degrees before and 40 degrees after. In this way ISL1, ISL
Since the reference voltage can be set arbitrarily by taking the sum of 2 and ISL3, the phase of the mask signal can be set to U1, V of the counter electromotive force.
1, W1 can be arbitrarily set.

In the above description, in order to simplify the description, the charging / discharging circuits 26, 27, 2 are operated at a constant motor rotation speed.
It was premised that the capacitor C of 8, the charging current Io, the discharging current 2Io, etc. did not vary. When this condition is not satisfied, there is a variation, but the phase of the mask signal can still be arbitrarily set with respect to the back electromotive forces U1, V1, W1. Further, in this embodiment, the phase of the mask signal can be fixed regardless of the rotation speed of the motor.

An embodiment for smoothing the switching of the drive current applied to the drive coils 1, 2, 3 will be described below.

The operation of the charge / discharge circuit 23 will be described in detail. FIG. 14 shows an example of the charge / discharge circuit. FIG. 15 is a signal waveform of each part.

The charge / discharge circuit 26 includes a pnp transistor Q.
1, a current mirror circuit composed of Q2, a constant current source Io for determining its operating current, and an npn transistor Q
3, a current mirror circuit composed of Q4, a constant current source 2Io for determining its operating current, and a switch means S for controlling switching of the current mirror circuit (Q3, Q4).
It is composed of W1 and a capacitor C1 having a capacitance value C.

Here, when the operating state of the transistors is in the active region, it is assumed that the h _FE of the transistors Q1 to Q4 is very large and the saturation current Is is equal to Q1, Q2 and Q3, Q4. . Then, the collector current of Q2 in the active region becomes I0 and the collector current of Q2 becomes 2Io. In the operation of the charge / discharge circuit, C1 is charged by the collector current of Q2 when SW1 is in the off state. Assuming that the maximum slope voltage is VSLMAX, the power supply voltage is VCC, and the saturation voltage of Q2 is Vce (sat), when VCC-Vce (sat)> VSLMAX, Q2 does not saturate, so if the Early effect is ignored, the collector current of Q2 is It becomes the constant current Io. On the other hand, S
When W1 is turned on, the capacitance C1 is discharged by the collector current (pull-in current) of the transistor Q4, which is larger than the discharge current of the transistor Q2. Then, the slope voltage is lowered and enters the saturation region of the transistor Q4, saturation of the transistor increases as the collector potential drops, h _FE of the transistor Q4 is decreased accordingly, the collector current decreases. Then, the output signals U1, V1, W1 of the back electromotive force detector fall,
The discharge is not completed in the rising phase, and
As shown in, the phase at which discharge is complete is delayed. Then, the reference voltage of the voltage-current conversion circuit 24 is set to the saturation voltage Vce of Q4,
When the slope voltage becomes Vce, the output currents IPU1 to IPU3, IP are output when the current is set to zero.
At the fall of L1 to IPL3, it does not become zero but changes abruptly, which causes vibration and noise of the motor.
Therefore, an embodiment for solving this problem will be described below.

FIG. 16 is a circuit configuration diagram of a second specific example of the charging / discharging circuit, and FIG. 17 shows operation waveform diagrams of respective parts. FIG.
The charge / discharge circuit 26 shown in FIG. 4 is provided with a clamp circuit composed of a constant current source I1, npn transistors Q13 and Q14, and a resistor R1 having a resistance value R.
7 and 28 are similarly clamp circuits (transistors Q15-
Q18) is added.

When such a clamp circuit is added, the transistor Q14 is in the off state when the slope voltage VSL1 is high, and is in the on state when the emitter potential thereof is about to fall below I1 × R, and the transistor Q14 has the emitter voltage. The impedance is reduced and clamp operation is performed. And I
The value of 1 × R is set larger than the saturation voltage of the transistor Q4. By doing this, when the transistor Q4 is turned on and the charge of the capacitor C1 is discharged, the collector potential remains at the clamp level (I
It is possible to operate in an active state at all times, and the collector current can be kept constant at 2 Io without depending on the potential of the slope voltage.
Then, the linearity of the sloped portion of the slope voltage is improved, and the minimum potential of the slope voltage becomes approximately I1 × R at the falling and rising phases of U1, V1, and W1 as shown in FIG. The reference voltage of the voltage-current conversion circuit 24 is set to I1
If the operation is performed so that the current becomes zero when the slope voltage is set to I1 × R, the output currents IPU1 to IPU1
At the fall of PU3, IPL1 to IPL3, it becomes almost zero and does not change sharply.

However, even in the second specific example of the charging / discharging circuit shown in FIG. 16, the linearity of the slope portion of the slope voltage is not perfect, and its problem will be described below. In the charge / discharge circuit shown in FIG. 16, the transistor Q14 is kept off during the period from the high potential side of the slope portion of the slope voltage to the lowest potential (clamp level).
If the operation of switching the transistor Q14 to the off state can be realized at the moment when the minimum potential (clamp level) is reached, the linearity of the inclined portion will not be impaired. However, since the emitter current of the transistor Q14 is determined according to the magnitude of the base-emitter voltage VBE of the transistor Q14, the emitter current of the transistor Q14 changes according to the magnitude of the slope voltage. That is, when the slope voltage falls with a slope, the transistor Q
The emitter current of transistor Q14 is V
It changes according to the exponential function of BE, and a half-conducting period occurs. The level of this problematic voltage is in the range from the lowest potential of the slope voltage to a potential higher by about 100 mV. About 100 mV from the lowest potential
If the potential is higher than the above, the emitter current of the transistor Q14 becomes 1/100 or less of the level during the clamp operation,
If the potential is higher than that, there is almost no problem even if the transistor Q14 is considered to be in the off state.

The halfway conduction of the transistor Q14 makes it difficult for the slope voltage waveform VSL1 to fall before reaching the clamp operation period, and the waveform at the portion where the slope portion switches to the clamp level is blunted. At this time, the level difference between the signal CHG1 and the lowest potential (clamp level) at the time of switching from high to low was 20 to 30 mV when the amplitude of VSL1 was 1 V _PP . This level is 50 to 50 in the circuit example of FIG.
Although it is improved to less than 1/2 with respect to 100 mV, the following problem still occurs.

In the subsequent signal processing, the charge / discharge circuit 2
The voltage / current conversion circuit 24 connected to the output of 3 performs voltage / current conversion of the slope voltage waveform protruding from the reference potential with reference to this clamp level, and has a triangular wave shape with a slightly blunted rise and fall. The output current is input to the trapezoidal wave current switching circuit 25. Then, the trapezoidal wave current switching circuit 25 uses the ISL1 and VS corresponding to VSL1.
When a solution of addition with ISL3 corresponding to L3 and U1 are combined to create an output signal of IPU1, a solution of addition with ISL1 corresponding to VSL1 and ISL3 corresponding to VSL3, and an inverted signal of U1 When the output signal of the composite IPL1 is produced by the product of, the switching operation is performed by the counter electromotive force U1.
At that time, since the switching control is performed in a state where the absolute value of both IPU1 and IPL1 does not reach zero, the trapezoidal wave (IPU1,
There is a step near the zero level of IPL1). Then, the pnp transistor 6 and the npn transistor driven by the trapezoidal current (IPU1, IPL1) and the output current of the similar current distribution circuit 13 drive the pnp transistor 6 by the current corresponding to IPU1 and correspond to IPL1. Since it operates so as to drive the npn transistor with the current, a step difference in the current occurs at the time of switching between the operation of the pnp transistor 6 and the operation of the npn transistor, which causes a sudden change in the drive torque of the motor. , Cause uneven rotation of the motor.

The third specific example of the charging / discharging circuit described below solves the above-mentioned problems, and a charging / discharging waveform V is obtained.
The linearity of SL1 to VSL3 is improved, especially the clamp level of the slope voltage and the switching of the slope portion are sharpened, and the driving current of the output pnp transistor and the npn transistor is once set to zero, and then the complementary transistor is formed. The purpose is to moderate the switching of.

FIG. 18 is a circuit configuration diagram of a third example of the charging / discharging circuit, and FIG. 19 shows operation waveform charts of respective parts. The third specific example of the charging / discharging circuit is obtained by adding SW14 to the second specific example (shown in FIG. 16), and the clamp circuit (transistor Q
13, Q14) is operated in synchronization with CHG2. The same applies to the charge / discharge circuits 27 and 28. C
Turn on SW14 only when HG2 is in low phase,
Clamp circuit (transistor Q1
When 4) is operated, the clamp transistor Q14 is cut off surely until the slope voltage decreases to the clamp level, so that an extra current is not given to the charge / discharge current for the capacitor C1. The linearity of the voltage waveform is improved, the switching between the slope portion of the slope voltage waveform and the minimum potential (clamp level) is sharpened, and the counter electromotive force (U1, V1) of each drive coil (1, 2, 3) is increased. , W1), the switching operation of the slope voltage waveform can be performed.

That is, the capacitor C1 is charged with the collector current Io of Q2 when SW1 is off, and the capacitor C1 is charged with the collector current 2I of Q4 when SW1 is on.
Since it is discharged at o, there is no difference between the charging time and the discharging time. SW1 in the charge / discharge circuit 26 when CHG2 is low
4 is turned on, SW15 is turned on when CHG3 is low in the charge / discharge circuit 27, and SW16 is turned on when CHG1 is low in the charge / discharge circuit 28.

FIG. 20 is an enlarged view of the slope voltage in which the period of the phase A shown in FIGS. 15, 17 and 19 is enlarged, and shows comparison of the first, second and third concrete examples. .
Then, with the slope voltage obtained in the embodiment of FIG.
At the timing of the edge of CHG1 to CHG3, a deviation of 50 to 100 mV was generated from the minimum potential of the slope voltage, but in the embodiment of FIG.
The deviation can be suppressed to 0 to 30 mV, and CHG1 to C
In the embodiment of FIG. 18 in which the clamp operation is performed in synchronization with the signal of HG3, the deviation can be suppressed to several mV, and the best performance is obtained in the embodiment of FIG.

In the embodiment shown in FIG. 18, in the subsequent signal processing, the voltage connected to the output of the charge / discharge circuit 23
The current conversion circuit 24 performs voltage / current conversion of the slope voltage waveform protruding from the reference potential with reference to the clamp level, and inputs a sharp switching triangular wave output current to the trapezoidal wave current switching circuit 25. . Then, the trapezoidal wave current switching circuit 25 combines the solution of the addition of ISL1 corresponding to VSL1 with the ISL3 corresponding to VSL3 and the product of U1 to produce the output signal of IPU1, and the ISL1 corresponding to VSL1. The product of the addition solution with ISL3 corresponding to VSL3 and the inverted signal of U1 is combined I
When the output signal of PL1 is produced, it is switched by the counter electromotive force U1. At that time, since the absolute values of both IPU1 and IPL1 are switched and controlled, there is no step near the zero level of the trapezoidal waves (IPU1, IPL1). And the trapezoidal wave current (IPU1, IPL1)
Driven by the output current of the current distribution circuit 13 similar to
The np transistor 6 and the npn transistor are the IPU 1
The pnp transistor 6 is driven by the current corresponding to
Since it operates to drive the npn transistor with the current corresponding to L1, the operation of the pnp transistor 6 and the np transistor
The polarity of the drive current is switched at the time of switching to the operation of the n-transistor, abrupt fluctuation of the drive torque of the motor is eliminated, and uneven rotation and vibration of the motor are eliminated.

Needless to say, the same effect can be obtained by using a clamp circuit other than the above embodiments.

[0057]

In the brushless motor drive device of the present invention, the slope voltage (charging / discharging waveform), which is the fundamental wave for forming the trapezoidal wave, is formed into a polygonal line waveform that sharply switches in synchronization with the back electromotive force. As a result, the switching operation of the complementary output transistors can be smoothed, and the noise and vibration during motor rotation can be reduced.

Further, the present invention can be applied to a motor in which the rotation speed fluctuates because the phase of the mask signal for detecting only the back electromotive force can be determined without adjustment regardless of the rotation speed of the motor.

[Brief description of drawings]

FIG. 1 is a circuit configuration diagram of an embodiment relating to a brushless motor driving device of the present invention.

FIG. 2 is an operation waveform diagram of each part in FIG.

FIG. 3 is a circuit configuration diagram of a specific example of the back electromotive force detector according to the embodiment.

FIG. 4 is an operation waveform diagram of each part of FIG.

FIG. 5 is a circuit configuration diagram of a first concrete example of a trapezoidal wave current combiner according to the embodiment.

6 is an operation waveform diagram of each part of FIG.

FIG. 7 is a circuit configuration diagram of a first specific example of the mask signal generator according to the embodiment.

8 is an operation waveform diagram of each part of FIG.

FIG. 9 is an operation waveform diagram of each part in FIG. 7 when the rotation speed of the motor is slow.

FIG. 10 is a circuit configuration diagram of a second specific example of the mask signal generator according to the embodiment.

11 is an operation waveform diagram of each part of FIG.

FIG. 12 is a circuit configuration diagram of a second specific example of the trapezoidal wave current combiner according to the embodiment.

13 is an operation waveform diagram of each part of FIG.

FIG. 14 is a circuit configuration diagram of a first specific example of the charging / discharging circuit according to the embodiment.

FIG. 15 is an operation waveform diagram of each part of FIG.

FIG. 16 is a circuit configuration diagram of a second specific example of the charging / discharging circuit according to the embodiment.

FIG. 17 is an operation waveform diagram of each part of FIG.

FIG. 18 is a circuit configuration diagram of a third specific example of the charging / discharging circuit according to the embodiment.

FIG. 19 is an operation waveform diagram of each part of FIG.

FIG. 20 shows first to third specific examples of the charge / discharge circuit (FIG. 14,
Figure for comparing the operation of Figure 16 and Figure 18)

[Explanation of reference signs] 1,2,3 Drive coil 4,5,6,7,8,9 Drive transistor 10 Back electromotive force detector 11 Trapezoidal wave current combiner 12 Mask signal generator 13 Current distribution circuit 14 Torque command signal Generator circuit

Claims (3)

[Claims] 1. A plurality of drive transistors for supplying a drive current to a multi-phase motor drive coil, a torque command signal generation circuit for detecting a motor drive current and generating a torque command signal, and a torque command signal generation circuit. A current distribution circuit for sequentially supplying a current according to an output signal to the bases of the plurality of drive transistors, a trapezoidal wave current combiner for outputting a trapezoidal wave current to the current distribution circuit, and a reverse coil generated in the motor drive coil. A drive device for a brushless motor having a back electromotive force detector for detecting an electromotive force and shaping the waveform, and a mask signal generator for outputting a signal for masking a part of a signal waveform to the back electromotive force detector. 2. A plurality of charging / discharging circuits, wherein a trapezoidal wave current combiner converts the output of the back electromotive force detector into a plurality of series of pulses and generates a triangular wave according to the outputs of the plurality of series of pulses, and A plurality of voltage-current conversion circuits for converting the output voltages of the plurality of charging / discharging circuits into currents; an addition circuit for adding the output currents of the plurality of voltage-current conversion circuits to synthesize a trapezoidal wave current; 2. The brushless motor drive device according to claim 1, further comprising a trapezoidal wave current switching circuit that switches and supplies the trapezoidal wave current to the bases of the suction side drive transistor and the discharge side drive transistor according to the output of the electromotive force detector. 3. A plurality of charging / discharging circuits, wherein the charging / discharging circuit converts the output of the back electromotive force detector into a plurality of series of pulses and generates a triangular wave according to the outputs of the plurality of series of pulses; 3. The brushless motor drive device according to claim 1, further comprising a circuit for clamping the pulse output signal according to claim 1.