JPH07303026A - Voltage controlled current source circuit, equivalent inductance circuit using the circuit and active filter circuit using them - Google Patents

Abstract

PURPOSE:To provide a voltage controlled current source circuit from which influence due to temperature can be eliminated. CONSTITUTION:The current source circuit I3 of a first differential amplifier circuit which comprises this voltage controlled current source circuit is constituted of a current mirror which transfers a current on a first resistor R19 from a fixed voltage source to an output side setting as a reference side current. The current source circuit 14 of a second differential amplifier circuit is constituted of a sub-voltage controlled current source circuit 25 which applies the voltage of prescribed times the voltage between the base and emitter of a transistor Q14 to a normal input terminal and connects an inverse input terminal to a normal output terminal and also, connects a second resistor R23 between the inverse input terminal and the ground and sets the inverse output terminal as an output terminal.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a temperature-compensated voltage controlled current source circuit, an equivalent inductance circuit using the same,
And an active filter circuit using them.

[0002]

2. Description of the Related Art FIG. 8 shows a circuit diagram of a second-pass filter circuit known as a conventional active filter circuit. This circuit is a circuit called a Sallen-Key circuit, which is configured in a monolithic integrated circuit without external parts, and includes resistors R1 to R3, capacitors C1 and C2, and an amplifier 3 having a voltage amplification factor K. It is composed of a positive feedback circuit consisting of. Reference numeral 1 is an input terminal and 2 is an output terminal. In this circuit, the input / output characteristics are obtained, and then the center frequency ω ₀ of the second-pass filter is obtained. Let Vi be the input voltage and Vo be the output voltage.

First, assuming that S = jω, the input / output characteristics are: Vo / Vi = [KS / (R2C2)] / [S ² + S {(1 / R1) (1 / C1 + 1 / C2) + 1 / (R2 · C1) + 1 / (R3 · C2) + (1-K) / (R2 · C2)} + {[1 / (R3 · C1 · C2)] (1 / R1 + 1 / R2)}]・ ・ It becomes (1).

On the other hand, the quadratic bandpass function is expressed by the following general formula, where H is a gain coefficient. Vo / Vi = [H · S · ω ₀ / Q] / [S ² + S · ω ₀ / Q + ω ₀ ² ] (2)

Therefore, ω ₀ ² of the denominator of this equation (2) and
From the denominator term (the term with no coefficient of S ² or S) of the equation (1) corresponding to this, ω ₀ ² = [1 / (R3 · C1 · C2)] (1 / R1 + 1 / R2) · ( 3) is obtained. Here, R1 = R2 = R, C1 = C2 = C
And put a, a ^{_{ω 0 = 2 1/2 /}} (C · R) ··· (4). Thus, it is understood from the equation (4) that the center frequency ω ₀ of the second-pass filter circuit is determined by the time constant of CR.

The temperature coefficient of the capacitor formed inside the integrated circuit becomes zero by making the shape of the capacitor into a MOS type. Similarly, the resistance formed inside the integrated circuit has a positive temperature coefficient, and each coefficient has a positive temperature coefficient. The resistance is the same. Here, assuming that the temperature coefficient of resistance inside the integrated circuit is TC and the change in resistance with temperature is a linear curve characteristic, the variation of the center frequency ω ₀ with respect to the ambient temperature T is obtained from dω ₀ / dT.

First, the resistance value R at the ambient temperature T
(T) is R (T) = TCRT + R (5), where R is the resistance value at a temperature of 0 ° C. Therefore, the variation dω ₀ / dT is as follows.

[0008] _{dω 0 / dT = (d /} dT) [2 1/2 / [C · R ^{(T)]] = - [2 1/2} · C · TC · R] / [C 2 (TC · R ^{· T + R) 2 = -} [2 1/2 · TC] / [CR (TC · T + 1) 2] ··· (6) the center frequency ω of the case where the circuit of Fig. 8 from the equation (6) and an integrated circuit _0, the temperature coefficient of resistance TC, it can be seen to vary in response to temperature.

FIG. 9 is a circuit diagram showing another conventional example and is a diagram showing a biquad filter circuit in which a biquadratic function is modeled. This circuit includes resistors R4 to R13, capacitors C3 to C4, and amplifiers 4 to 7. 8 is an input terminal,
9 to 11 are output terminals. Here, in order to make a comparison with the circuit shown in FIG.
Letting the input voltage of the input terminal 7 be Vi and the output voltage of the output terminal 9 be Vo, the input / output characteristics thereof are obtained, and the center frequency ω ₀ is obtained.

Vo / Vi = − [1 / (R7 · C3)] · [S / {S ² + S (1 / R4 · C3) + R9 / (C3 · C4 · R5 · R6 · R8)}] ... Since (7), in this equation (7), ω ₀ of the above equation (2)
Ω ₀ = R9 / (C3 · C4 · R5 · R6 · R8) (8) is obtained from the term corresponding to R5 = R6 = R8 = R9 =
If R and C3 = C4 = C, then ω ₀ = 1 / CR (9) Thus, also in the circuit of FIG. 9, the center frequency ω ₀ is determined by the CR time constant.

Here, the temperature coefficient of the resistance inside the integrated circuit and the temperature coefficient of the capacitor are the same as in the case of FIG.
Similarly, when dω ₀ / dT is calculated, dω ₀ / dT = (d / dT) [1 / [C · R (T)]] = − [C · TC · R] / [C ² (TC · R · T + R) ² ] = − [TC] / [CR (TC · T + 1) ² ] ... (10). Thus, even in the circuit shown in FIG. 9,
When this is integrated into a circuit, the center frequency ω ₀ varies depending on the temperature due to the temperature coefficient of resistance TC.

[0012]

As described above, in the active filter circuit in which the conventional circuit is formed into a monolithic integrated circuit, as is apparent from the equations (4) and (9), the equation for determining the center frequency ω ₀ is used. Be sure to enter the item "1 / CR".
That is, since the conventional active RC filter circuit has a configuration in which the transfer of energy at the resonance point of the LC filter is complemented by the feedback circuit using the RC network and the active element, the center frequency ω ₀ of the active RC filter circuit is determined. Be sure to enter the item "1 / CR".

When an active RC filter circuit is constructed in a monolithic integrated circuit without using an external element, if a MOS type having an oxide film or a nitride film is used as the dielectric of the capacitor formed therein, the temperature rises. Although the coefficient is zero, the P ⁺ diffusion resistance has a temperature coefficient of 2000 to 500.
Since it reaches as high as 0 ppm / ° C., the center frequency ω ₀ largely depends on this temperature coefficient of resistance, and its value fluctuates depending on the temperature.

The present invention has been made in view of the above points, and an object thereof is to provide a temperature-controlled voltage-controlled current source circuit that requires no external parts, and by utilizing this, external parts. Providing unnecessary equivalent inductance circuit,
Further, it is to provide an active filter circuit which is temperature-compensated by utilizing these.

[0015]

The voltage controlled current source circuit of the first invention comprises a first non-inverting input terminal and a second inverting input terminal, and an operating current is determined by the first current source circuit. A second differential amplifier circuit in which an operating current is determined by the first differential amplifier circuit and the second current source circuit, and the differential output signal of the first differential amplifier circuit is input as a differential input signal. A voltage controlled current source circuit for controlling gain and compensating temperature characteristics according to a ratio of currents of the first current source circuit and the second current source circuit, wherein the first current source The circuit is configured by a current mirror circuit that uses the current flowing from the fixed voltage source to the first resistor as the reference side current and transfers it to the output side, and the second current source circuit is connected to the second non-inverting input terminal. Apply a voltage that is a predetermined multiple of the base-emitter voltage of the transistor, With connecting second inverting input terminal to the non-inverting output terminal connected to a second resistor between ground and the inverting input terminal, and configured in sub-voltage control current source circuit to the output terminal of the inverting output terminal.

The equivalent inductance circuit of the second invention is
Two voltage controlled current source circuits of the first invention are prepared as a first voltage controlled current source circuit and a second voltage controlled current source circuit, and the normal input terminal of the first voltage controlled current source circuit and the above The output terminal of the second voltage controlled current source circuit is commonly connected, and the output terminal of the first voltage controlled current source circuit and the inverting input terminal of the second voltage controlled current source circuit are commonly connected to MO.
It was configured by connecting to one end of an S-type capacitor.

In the active filter circuit of the third invention, a parallel resonant circuit formed by connecting another MOS type capacitor in parallel to the equivalent inductance circuit of the second invention is connected between the output terminal and the ground, and The voltage-controlled current source circuit according to claim 1 is inserted between the output terminal and the input terminal in a negative feedback connection.

An active filter circuit according to a fourth aspect of the invention is such that a series resonance circuit formed by connecting another MOS type capacitor in series to the equivalent inductance circuit of the second aspect of the invention is connected between an output terminal and ground, and The voltage-controlled current source circuit according to claim 1 is inserted between the output terminal and the input terminal in a negative feedback connection.

[0019]

In the voltage controlled current source circuit of the present invention, the positive temperature coefficient due to the first resistance of the first current source circuit is converted into the negative temperature coefficient of the base-emitter voltage of the transistor of the second current source circuit. Is canceled by the difference between the value obtained by multiplying by a predetermined factor and the positive temperature coefficient by the second resistance. As a result, the temperature dependence of the input / output characteristics of the voltage controlled current source circuit is eliminated and temperature compensation is performed, and the characteristics of the equivalent inductance circuit and active filter circuit that use this voltage controlled current source circuit also disappear, and the temperature dependency is eliminated. Will be compensated.

[0020]

【Example】

[Voltage Controlled Current Source Circuit] An embodiment of the present invention will be described below. FIG. 1 is a circuit diagram showing a voltage controlled current source circuit (VCCS) 20 according to an embodiment of the present invention. The transistors Q1 and Q2 are emitter follower transistors whose bases are connected to the input terminals 21 and 22 to form an input circuit. A transistor Q for inputting a current obtained by subtracting the emitter currents of the transistors Q1 and Q2 from the currents of the current source circuits I1 (current value I1) and I2 (current value I2)
A circuit including 3, Q4, emitter resistors R14 and R15, and a current source circuit I3 (current value I3) constitutes a first differential amplifier circuit. Q5 and Q6 are diode-connected transistors as loads of the first differential amplifier circuit. Transistors Q7 and Q8 that use the output signal of the first differential amplifier circuit as a differential input signal, and current source circuit I4 (current value I
The circuit composed of 4) constitutes a second differential amplifier circuit. A diode-connected transistor Q9 is connected to the collector of one transistor Q7 of the second differential amplifier circuit, and current-mirror-connected transistors Q10 and Q11 are connected to the collector of the other transistor Q8. R16 to R18 are load resistors. The current source circuit I5 (current value I5) is connected to the collector of the transistor Q11, and the common connection point of the current source circuit I5 and the collector of the transistor Q11 is connected to the output terminal 23, where the output voltage Vo is obtained. .

In the voltage controlled current source circuit 20, when the differential input voltage Vi is applied between the input terminals 21 and 22,
Assuming that the voltage gain of the emitter followers of the transistors Q1 and Q2 is 1, the transistors Q3 and Q of the first differential amplifier circuit
The above-mentioned input voltage Vi is applied to 4. Here, the transconductance Gm of the first differential amplifier circuit
When 0 is obtained by a π-type equivalent circuit, it becomes as follows. Gm0 = gm / (1 + gm.R _E ) = 1 / [(1 / gm) + R _E ]. (11) where gm: Q3 and Q4 transconductance (= I3 / 2V _T ) V _T : thermal voltage R _E = R14 = R15

Therefore, when the relation between the emitter resistance r _e (= 1 / gm) of the transistors Q3 and Q4 and the above resistance R _E is r _e << R _E , the equation (11) is expressed as Gm0 = 1 / R _E .. (12), so collector current I of transistors Q3 and Q4
c3, Ic4, relations Gm0 is, I3 = Ic3 + Ic4 ·· ( 13) Ic3-Ic4 = Gm0 · Vi = Vi / R E ·· (14) Therefore, from the equation (13), (14), Ic3 = [ I3 / 2] + [Vi / (2 · R _E )] ·· (15) Ic4 = [I3 / 2] − [Vi / (2 · R _E )] ·· (16)

When the emitter sizes of the transistors Q5 and Q6 are the same, the base-emitter voltage V
_{EB 5} and V _{EB 6} are as follows when Is is the reverse saturation current of the transistor. _{V EB 5 = V T · ln} [Ic3 / Is] = V T · ln [ [I3 / 2 + Vi / (2 · R E) ] / Is] ·· (17) V EB 6 = V T · ln [Ic4 / Is] = V _T · ln [[I3 / 2-Vi / (2 · R E) ] / Is] ·· (18)

The transistor Q of the second differential amplifier circuit
7, the base potentials V _B7 and V _B8 of Q8 are expressed by equations (17) and (1
Using 8), it becomes as follows. _{V B7 = Vcc-V EB 5} = Vcc- [V T · ln { [I3 / 2 + Vi / (2 · R E) ] / Is}] ·· (19) V B8 = Vcc-V EB 6 = Vcc- [ V _T · ln {[I3 / 2-Vi / (2 · R E) ] / Is}] ·· (20)

Using these equations (19) and (20), the base-to-base voltage V _BB of the transistors Q7 and Q8 is as follows. V _BB = V _B7 −V _B8 = V _T · ln [{I3 / 2−Vi / (2 · R _E )} / {I3 / 2 + Vi / (2 · R _E )}] ··· (21)

The collector current of the transistor Q9 is -i _0, and the collector current of the transistor Q10 is + i.
When the base-to-base voltage V _BB of the transistors Q7 and Q8 when ₀ is obtained, the result is as follows. V _{BE 7} = V _T · ln [(I4 / 2−i ₀ ) / Is] ··· (22) V _{BE 8} = V _T · ln [(I4 / 2 + i ₀ ) / Is] ··· (23) V _{BB = V BE 7 -V BE 8 =} V T · ln [(I4 / 2-i 0) / (I4 / 2 + i 0)] ·· (24)

Here, by rearranging the equations (21) and (24), the input / output characteristic of the voltage controlled current source circuit 20, that is, the output current i _{0 with} respect to the input voltage Vi is as follows. _{i 0 / Vi = I4 / (} 2 · R E · I3) ·· (25)

[Integrator] An integrator is formed by connecting an internal capacitor Co (capacitance Co) between the output terminal 23 of the voltage controlled current source circuit 20 shown in FIG. 1 and the ground. Then, from the above equation (25), the following is obtained. From Vo / Vi = I4 / (2 · S · Co · R E · I3) ·· (26) This equation (26), the integrator gain, a current source circuit I3,
It can be seen that it is controlled by the ratio of the currents I3 and I4 of I4.

Next, the current source circuit I3 of the voltage controlled current source circuit 20 described above is realized by the circuit shown in FIG.
This circuit is a current mirror connection transistor Q12, Q
13 and a resistor R19 (resistance value R19) that determines the reference current. Vf is a reference voltage. The collector of the transistor Q13 becomes the current output terminal 24.

The collector current, that is, the output current I3 of the transistor Q13 of this circuit is as follows when the base-emitter voltage of the transistor Q12 is V _{BE 1 2} . _{I3 = (Vf-V BE 1} 2) / R19 ·· (27) Therefore, substituting the equation (27) into equation (26), as follows. Vo / Vi = I4 / [2 · S · Co · R E · (Vf-V BE 1 2) / R19] ·· (28)

In this equation (28), the resistance values R _E and R _E
Since 19 is canceled its temperature coefficient, if Ae cancel the temperature coefficient and the current I4 term with the temperature dependence of _{(Vf-V BE 1 2)} , the gain of the integrator will be no temperature coefficient .

Therefore, first, (Vf-V _{BE in} equation (28)
Try to calculate the sensitivity to temperature of _{1 2).} The voltage of the resistor R19 as V _{R1 9,} when put between _{(Vf-V BE 1 2)} = V R1 9, the sensitivity S _T is as follows. _{S T = (T / V R1} 9) (dV R1 9 /dT)=-0.15 ·· (29) which is meant to vary -0.15% per 1% temperature. Here, the collector current of the transistor Q12 is 1
0 .mu.A, the saturation current Is of e ^{-1 6} A, in which the Vf was obtained under the conditions as 5V, the correct temperature coefficient, 505ppm / ℃
Becomes

Next, a circuit as shown in FIG. 3 is used as the current source circuit I4 of the voltage controlled current source circuit 20. This current source circuit I4 causes the above-mentioned (Vf
Cancel the temperature coefficient of -V _{BE 1 2).} Reference numeral 25 is another voltage controlled current source circuit (different from the configuration shown in FIG. 1 as shown in a concrete circuit in FIG. 4), which receives a differential voltage and receives a positive electrode and an absolute value which are equal to each other. Outputs the negative current. Q
14 is a transistor, R20 to R23 are resistors, and 26 is a current output terminal.

In this current source circuit I4, when the input offset voltage and the input bias voltage of the voltage controlled current source circuit 25 are ignored, the current value I4 output from the output terminal 26 is
When a voltage generated across the resistor R23 and V _{R2 3,} as follows. I4 = −V _{R2 3} / R23 ··· (30) Further, here, since the non-inverting output terminal (+) of the voltage controlled current source circuit 25 is connected to the inverting input terminal (−), the negative feedback configuration is provided. , The voltage Va at the non-inverting input terminal (+) is V
a a = V _{R2 3,} therefore, the output current I4 is obtained by the following equation. I4 = -Va / R23 ... (31)

In the voltage controlled current source circuit 20 shown in FIG. 1 of this embodiment, the current source circuit I4 cancels the temperature coefficient of (Vf-V _{BE 1 2} ) as will be described below. As described above, when the resistors R20 to R23 to be used are formed inside the integrated circuit, they all have the same temperature coefficient. The resistance value (R
23) changes, if the non-inverting input terminal (+) of the voltage controlled current source circuit 24 is at the fixed potential Va (when temperature compensation is performed), the potential of the inverting input terminal (-) is also fixed at the same potential. Therefore, the temperature coefficient of the current I _{R2 3} flowing through the resistor R23 becomes equal to the resistance temperature coefficient.

[0036] Thus, the difference between the temperature coefficient of the voltage Va of the resistance temperature coefficient and a non-inverting input terminal of a resistor R23 (+) to match the temperature coefficient of the above _{(Vf-V BE 1 2)} , voltage controlled current If the voltage Va at the non-inverting input terminal (+) of the source circuit 25 is adjusted, the current I _{R2 3} (= I4) flowing through the resistor R23
It is equal to the temperature coefficient of _{(Vf-V BE 1 2)} . (1 / Va) (dVa / dT) = (1 / R23) (dR23 / dR) ... (32)

[0037] Here, when the base-emitter voltage of the transistor Q14 and _{V BE 1 4, Va = Vcc} -nV BE 1 4 ·· (33) However, since it is n = (R20 + R21) / R21, (Vf to carry out the same change to the temperature coefficient of -V _{BE 1 2)} is, (1 / R23) (dR23 / dT) - (1 / nV BE 1 4) (dnV BE 1 4 / dT) = [1 / _{(Vf-V bE 1 2)} ] [d (Vf-V bE 1 2) / dT] wherein it is sufficient to satisfy the ... (34).

As described above, the voltage V of the transistor 14
_{BE 1} multiplied by a coefficient determined by the _fourth resistor R18, the negative temperature coefficient with the R19, to adjust the temperature coefficient of the current I4 by its value, by combining the temperature coefficient of _{(Vf-V BE 1 2)} , FIG. The temperature coefficient of the integrator shown in 1 can be canceled. The current source circuit I shown in FIG.
As for 5 as well, since it is necessary to set I5 = I4 / 2, it is realized by a similar circuit configuration to cancel the temperature coefficient.

In the concrete circuit shown in FIG. 4, the same parts as those in FIG. 3 are designated by the same reference numerals. The transistors Q15 and 16 form a differential circuit, the transistors Q17 and Q18 form an active load of this differential circuit, the transistor Q19 forms a current mirror circuit with the transistor Q18, and the transistors Q20 to Q24 form an output circuit. To do. I6 and I7 are current source circuits whose currents are I6 and I7, and are set to I6 = I7 / 2.

Here, it will be examined whether or not an error occurs in the output current I4 due to the current source circuits I6 and I7.
The input bias current can be considered as a factor causing such an error. First, when the differential circuit is in a balanced state,
The base currents (input bias currents) I _{B2 4} , I _{B2 5} of the transistors Q24, Q25 are
If the current amplification factor of Q25 is h _fe , it becomes as follows. I _{B2 4} = I _{B2 5} = I _B = (I7 / 2) / h _fe ··· (35)

[0041] Then, the input bias current I _B inverting input terminal as an error component (-) to the flow, since the output current I4 from the non-inverted output terminal (+) flows, the voltage V _{R2 3} of resistor R23, the following become that way. _{V R2 3 = R23 (I B} + I4) ·· (36)

Therefore, the output current I4 becomes as follows. _{I4 = (V R2 3 -I B} · R23) / R23 = V R2 3 / R23-I B ·· (37) that is, the output current I4 will decrease by an error bias current I _B. However, since the bias current I _B is as shown in the equation (35), even if I7 / 2 = I4, the error is 1 / h _fe, which is extremely small and can be ignored.

[Equivalent Inductance Circuit] FIG. 5
An equivalent inductance circuit constructed by using the integrator whose temperature coefficient is compensated as described above is shown in Fig. This is composed of a gyrator that equivalently changes capacitance to inductance. In FIG. 5, 27, 28
Is a voltage controlled current source circuit having the same configuration as the voltage controlled current source circuit 20 shown in FIG.
1, Gm2, and their normal input terminals (+) are shown in FIG.
The input terminal 21 and the inverting input terminal (−) of are equivalent to the input terminal 22. The voltage controlled current source circuit 27 and the MOS type internal capacitor Co constitute an integrator. The reference voltage V _ref is applied to the non-inverting input terminal (+) of one of the voltage controlled current source circuits 28 and the inverting input terminal (−) of the other voltage controlled current source circuit 27.
Is applied. First, the input impedance Z viewed from the connection terminal 29 is obtained.

The current flowing through the connecting terminal 29 is i ₁ , the current flowing through the capacitor Co is i ₂ , the voltage applied to the connecting terminal 29 is v ₁ , and the voltage applied to the capacitor Co is v _2.
Then, i ₂ = Gm1 · v ₁ v ₂ = i ₂ / (S · Co) i ₁ = Gm2 · v ₂ , so that Z = v ₁ / i ₁ = S · Co / (Gm1 · Gm2) The equivalent inductance L is as follows. L = Co / (Gm1 · Gm2) Here, if Gm1 = Gm2 = Gm, then L = Co / Gm ² ··· (38).

From the above, the above-mentioned equivalent inductance L
A resonance circuit can be configured by using a MOS type internal capacitor (separate from Co) and an active filter circuit can be configured. For example, a bandpass filter can be constituted by a parallel resonance circuit, and a band elimination filter can be constituted by a series resonance circuit. At this time, a resistance element is required between the signal input terminal and the equivalent inductance L, but the resistance created inside the integrated circuit has a poor absolute value accuracy of ± 10 to 20%, and is realized by the gyrator described above. Since the value of the equivalent inductance L also depends on 1 / Gm, it changes according to the current value I4 of the equation (26). Therefore, in order to keep the Q of the configured active filter constant, it is necessary to make the resistance value between the signal input terminal and the equivalent inductance L proportional to 1 / Gm.

[Bandpass Filter] FIG. 6 shows a bandpass filter constructed in this manner. In FIG. 6A, reference numeral 30 denotes a voltage-controlled current source circuit having a negative feedback configuration that functions as the resistor, and connects the output terminal and the inverting input terminal (-) of the voltage-controlled current source circuit 20 shown in FIG. The configuration described above is used. 31 is an input terminal, 3
2 is an output terminal. Here, a band elimination filter is constructed by connecting a MOS type internal capacitor Ca in parallel to the equivalent inductance L shown in FIG.

Letting the input voltage be Vi and the output voltage be Vo,
Find its transfer function. First, the input impedance Z1 of the parallel resonant circuit viewed from the output terminal 32 is as follows. Z1 = S · L / (S ² · L · Ca + 1) ··· (39) The output voltage Vo is as follows when the transconductance of the voltage controlled current source 30 is Gm3. Vo = Gm3 (Vi-Vo) · Z1 = Gm3 (Vi-Vo) · S · L / (S 2 · L · Ca + 1) ·· (40) Therefore, the transfer function is as follows. Vi / Vo = [S · Gm3 / Ca] / [S ² + S · Gm
3 / Ca + 1 / (L ・ Ca)]

Next, the general formula (2) showing the second-order band pass function and the formula (41) are compared, and the gain coefficient H =
As 1, when obtaining the center frequency ^{_{ω 0 2, Q, ω 0}} 2 = 1 / (L · Ca) ·· (42) Q = (Ca / L) 1/2 / Gm3 since the ... (43) By substituting the equation (38) of the equivalent inductance L into the equations (42) and (43), the following is obtained. ω ₀ ² = Gm ² / (Co · Ca) Therefore, the center frequency ω ₀ is as follows. _{ω 0 = Gm / (Co ·} Ca) 1/2 ·· (44)

This temperature change of the center frequency ω ₀ is zero as follows because the temperature coefficient of the MOS type capacitor in the integrated circuit is zero and the integrator gain is temperature compensated as described above. Become. For _{(dω 0 / dT) = 0} ·· (45) Q, with Q = (Gm / Gm3) ( Co / Ca) 1/2 ·· (46), a temperature change the center frequency omega ₀ of the Q Similarly, dQ
/ DT = 0.

[Band Elimination Filter] FIG. 7 is a block diagram when applied to a band elimination filter. Figure 6
The same reference numerals are given to the same items as the ones that are hidden. Here, the capacitor Ca is connected in series to the equivalent inductance L. Since the input impedance Z2 of the series resonance circuit of this circuit is Z2 = (S ² · L · Ca + 1) / (S · Ca) ··· (47), the output voltage Vo is Vo = Gm3 (Vi-Vo) · Z2 = Gm3 (Vi-Vo ) · (S 2 · L · Ca + 1) / become (S · Ca) ·· (48 ). Therefore, the input / output function is: Vi / Vo = [S ² + 1 / (L · Ca)] / [S ² + S / (Gm3 · L) + 1 / (L · Ca)] ··· (49).

On the other hand, the secondary band elimination function is given by the following general formula. Vi / Vo = [S ² + ω _N ² ] / [S ² + S · ω _N / Q _N + ω _N ² ] _··· (50) By comparing the equations (49) and (50) of the second-order band elimination function,
When determining the center frequency omega _N ² and Q, the ^{_{ω N 2 = 1 / (L}} · Ca) ·· (51) Q = Gm3 · (L / Ca) 1/2 ·· (52). By substituting the equation (38) of the equivalent inductance L into the equations (51) and (52), the following is obtained. _{^{ω N 2 = Gm 2 / (}} Co · Ca) ·· (53) ω N = Gm / (Co · Ca) 1/2 ·· (54)

This temperature change of the center frequency ω _N is because the temperature coefficient of the MOS type capacitor inside the integrated circuit is zero and the integrator gain is compensated as described above, so that dω _N / dT = 0. (55) and the Q of the filter is as follows. Temperature change of Q = (Gm3 / Gm) / (Co · Ca) 1/2 ·· (56) The Q, like the well center frequency d [omega _N as follows. dQ / dt = 0

[0053]

As described above, the output current of the voltage controlled current source circuit of the present invention is temperature-compensated and is not affected by temperature. Therefore, the integrator made up of this voltage-controlled current source circuit and the MOS type capacitor inside the integrated circuit is also temperature-compensated, and the equivalent inductance circuit made up of the integration circuit and another voltage-controlled current source circuit is also temperature-compensated. To be done. Therefore, the resonance circuit created by this equivalent inductance circuit and another MOS-type capacitor inside the integrated circuit is also temperature-compensated, and the active filter circuit configured by using them is also temperature-compensated. An excellent filter circuit can be realized.

[Brief description of drawings]

FIG. 1 is a circuit diagram of an integrator that is a combination of a voltage controlled current source circuit and a capacitor according to an embodiment of the present invention.

FIG. 2 is a block diagram of a current source circuit I3 of FIG.

FIG. 3 is a block diagram of a current source circuit I4 of FIG.

FIG. 4 is a specific circuit diagram of the current source circuit I4 of FIG.

5 is a block diagram of an equivalent inductance configured by using a capacitor different from the integrator of FIG. 1. FIG.

6A is a block diagram of a bandpass active filter circuit using the equivalent inductance of FIG. 5, and FIG. 6B is an equivalent circuit diagram thereof.

7A is a block diagram of a band elimination active filter circuit using the equivalent inductance of FIG. 5, and FIG. 7B is an equivalent circuit diagram thereof.

FIG. 8 is a block diagram of a conventional Sallen-Key circuit.

FIG. 9 is a block diagram of a conventional Biquad filter circuit.

[Explanation of symbols]

1: Input terminal, 2: Output terminal, 3-7: Amplifier, 8: Input terminal, 9-11: Output terminal, 20: Voltage control current source circuit, 21: Forward input terminal, 22: Inversion input terminal, 23 ,
24: output terminal, 25: voltage controlled current source circuit, 26: inverting output terminal, 27, 28: voltage controlled current source circuit, 29:
Connection terminal, 30: voltage control current source circuit, 31: input terminal, 32: output terminal, I3: first current source circuit, 14:
Second current source circuit.

Claims (4)

[Claims] 1. A first differential amplifier circuit having a first non-inverting input terminal and a second inverting input terminal, the operating current of which is determined by a first current source circuit, and a second current source circuit. An operating current is determined, and the second differential amplifier circuit receives the differential output signal of the first differential amplifier circuit as a differential input signal, and comprises the first current source circuit and the second current source circuit. A voltage controlled current source circuit for controlling a gain and compensating a temperature characteristic according to a current ratio of a current source circuit, wherein the first current source circuit is based on a current flowing from a fixed voltage source to a first resistor. A current mirror circuit that transfers this to the output side as a side current is applied, and the second current source circuit is applied to the second non-inverting input terminal by a voltage that is a predetermined multiple of the base-emitter voltage of the transistor. , When connecting the second inverting input terminal to the non-inverting output terminal, Second resistor connected and configured in sub-voltage control current source circuit to the output terminal of the inverting output terminal, the voltage controlled current source circuit, characterized in that between the ground and the inverting input terminal. 2. A voltage control current source circuit according to claim 1 is provided as a first voltage control current source circuit and a second voltage control current source circuit, and the positive voltage control current source circuit is positive. The input terminal and the output terminal of the second voltage controlled current source circuit are commonly connected, and the output terminal of the first voltage controlled current source circuit and the inverting input terminal of the second voltage controlled current source circuit are connected. An equivalent inductance circuit characterized by being commonly connected to one end of a MOS type capacitor. 3. A parallel resonance circuit, which is formed by connecting another MOS type capacitor in parallel to the equivalent inductance circuit according to claim 2, is connected between the output terminal and the ground, and between the output terminal and the input terminal. An active filter circuit, wherein the voltage controlled current source circuit according to claim 1 is inserted in a negative feedback connection. 4. A series resonance circuit, which is formed by connecting another MOS type capacitor in series to the equivalent inductance circuit according to claim 2, is connected between the output terminal and the ground, and is connected between the output terminal and the input terminal. An active filter circuit, wherein the voltage controlled current source circuit according to claim 1 is inserted in a negative feedback connection.