JP3332290B2 - Voltage controlled current source circuit, equivalent inductance circuit using the same, and active filter circuit using the same - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a temperature-compensated voltage-controlled current source circuit, an equivalent inductance circuit using the same,
And an active filter circuit using them.

[0002]

2. Description of the Related Art FIG. 8 shows a circuit diagram of a second-pass filter circuit known as a conventional active filter circuit. This circuit is a so-called Sallen-Key circuit, which is configured without external components in a monolithic integrated circuit, and includes an amplifier 3 having resistors R1 to R3, capacitors C1, C2, and a voltage amplification factor K. And a positive feedback circuit composed of 1 is an input terminal and 2 is an output terminal. In this circuit, the input / output characteristics are obtained, and then the center frequency ω ₀ of the secondary pass filter is obtained. Vi is an input voltage, and Vo is an output voltage.

First, assuming that the input / output characteristics are S = jω, Vo / Vi = [KS / (R2 / C2)] / [S ² + S ｛(1 / R1) (1 / C1 + 1 / C2) + 1 / (R2 · C1) + 1 / (R3 · C2) + (1−K) / (R2 · C2)｝ + {[1 / (R3 · C1 · C2)] (1 / R1 + 1 / R2)}]・ ・ (1)

On the other hand, the secondary bandpass function is represented by the following general formula, where H is a gain coefficient. Vo / Vi = [HS · ω ₀ / Q] / [S ² + S · ω ₀ / Q + ω ₀ ² ] (2)

Therefore, ω ₀ ² of the denominator of the equation (2) and
Than the denominator term in equation (1) corresponding to this (claim no factor of S ² and S), ω ₀ ² = [1 / (R3 · C1 · C2 ) ] (1 / R1 + 1 / R2 ) ·· ( 3) is obtained. Here, R1 = R2 = R, C1 = C2 = C
And put a, a ^{_{ω 0 = 2 1/2 /}} (C · R) ··· (4). Thus, it can be seen from equation (4) that the center frequency ω ₀ of the secondary pass filter circuit is determined by the time constant of CR.

The temperature coefficient of the capacitor formed inside the integrated circuit becomes zero by making the shape of the MOS type. Similarly, the resistance formed inside the integrated circuit has a positive temperature coefficient and the coefficient is The resistance is also the same. Here, assuming that the temperature coefficient of resistance inside the integrated circuit is TC, and that the change in resistance due to temperature is a linear curve characteristic, the variation of the center frequency ω ₀ with respect to the ambient temperature T is obtained from dω ₀ / dT.

First, the resistance value R at an ambient temperature T
(T) is represented by R (T) = TC ・ RT ・ R, where R is the resistance value at a temperature of 0 ° C. (5). Therefore, the variation dω ₀ / dT is as follows.

[0008] _{dω 0 / dT = (d /} dT) [2 1/2 / [C · R ^{(T)]] = - [2 1/2} · C · TC · R] / [C 2 (TC · R ^{· T + R) 2 = -} [2 1/2 · TC] / [CR (TC · T + 1) 2] ··· (6) the center frequency ω of the case where the circuit of Fig. 8 from the equation (6) and an integrated circuit _0, the temperature coefficient of resistance TC, it can be seen to vary in response to temperature.

FIG. 9 is a circuit diagram showing another conventional example, and is a diagram showing a biquad filter circuit in which a biquadratic function is simulated by a circuit. This circuit includes resistors R4 to R13, capacitors C3 to C4, and amplifiers 4 to 7. 8 is an input terminal,
9 to 11 are output terminals. Here, in order to compare with the circuit shown in FIG.
Assuming that the input voltage of the input terminal 7 is Vi and the output voltage of the output terminal 9 is Vo, the input / output characteristics are obtained, and the center frequency ω ₀ is obtained.

Vo / Vi = − [1 / (R7 · C3)] · [S / {S ² + S (1 / R4 · C3) + R9 / (C3 · C4 · R5 · R6 · R8)}] (7) Therefore, in Expression (7), ω _{0 in} Expression (2) is used.
From the term corresponding to ω ₀ = R9 / (C3 · C4 · R5 · R6 · R8) (8), the following is obtained. From this, R5 = R6 = R8 = R9 =
If R, C3 = C4 = C, then ω ₀ = 1 / CR (9) Thus, also in the circuit of FIG. 9, the center frequency ω ₀ is determined by the CR time constant.

Here, the temperature coefficient of the resistance inside the integrated circuit and the temperature coefficient of the capacitor are the same as those in FIG.
Similarly, when dω ₀ / dT is obtained, dω ₀ / dT = (d / dT) [1 / [C · R (T)]] = − [C · TC · R] / [C ² (TC · R · T + R) ² ] = − [TC] / [CR (TC · T + 1) ² ] (10) Thus, in the circuit shown in FIG.
When this is integrated, the center frequency ω ₀ fluctuates according to the temperature by the resistance temperature coefficient TC.

[0012]

As described above, in the active filter circuit which is a monolithic integrated circuit of the conventional circuit, as is apparent from the equations (4) and (9), the equation for determining the center frequency ω ₀ is obtained. The item "1 / CR" is always entered.
That is, the conventional active RC filter circuit has a configuration in which the transfer of energy at the resonance point of the LC filter is supplemented by the feedback circuit using the RC network and the active elements, so that the center frequency ω ₀ of the active RC filter circuit is determined. The item "1 / CR" is always entered.

When an active RC filter circuit is formed in a monolithic integrated circuit without using an external element, a MOS type having an oxide film or a nitride film is used as a dielectric of a capacitor formed therein. Although the coefficient is zero, the P ⁺ diffusion resistance has a temperature coefficient of 2000 to 500.
Since the center frequency ω ₀ reaches 0 ppm / ° C., the center frequency ω ₀ greatly depends on the temperature coefficient of resistance, and its value varies depending on the temperature.

The present invention has been made in view of the above points, and an object of the present invention is to provide a voltage-controlled current source circuit which does not require external components and which is temperature-compensated. Provide unnecessary equivalent inductance circuit,
It is another object of the present invention to provide a temperature-compensated active filter circuit utilizing these.

[0015]

According to a first aspect of the present invention, a voltage controlled current source circuit includes a first non-inverting input terminal and a second inverting input terminal, and an operating current is determined by the first current source circuit. An operating current is determined by a first differential amplifier circuit and a second current source circuit, and a second differential amplifier circuit inputs a differential output signal of the first differential amplifier circuit as a differential input signal A voltage-controlled current source circuit for controlling a gain and compensating for a temperature characteristic by a ratio of currents of the first current source circuit and the second current source circuit, wherein the first current source circuit The circuit is constituted by a current mirror circuit which transfers a current flowing from the fixed voltage source to the first resistor to the output side as a reference side current, and connects the second current source circuit to a second non-inverting input terminal. Apply a voltage that is a predetermined rate times the base-emitter voltage of the transistor, With connecting second inverting input terminal to the non-inverting output terminal connected to a second resistor between ground and the inverting input terminal, and configured in sub-voltage control current source circuit to the output terminal of the inverting output terminal.

[0016] The equivalent inductance circuit of the second invention is as follows.
Two voltage controlled current source circuits of the first invention are prepared as a first voltage controlled current source circuit and a second voltage controlled current source circuit, and a non-inverting input terminal of the first voltage controlled current source circuit and The output terminal of the second voltage-controlled current source circuit is commonly connected, and the output terminal of the first voltage-controlled current source circuit and the inverting input terminal of the second voltage-controlled current source circuit are commonly connected to the MO terminal.
The capacitor was connected to one end of an S-type capacitor.

An active filter circuit according to a third aspect of the present invention is configured such that a parallel resonance circuit formed by connecting another MOS type capacitor in parallel with the equivalent inductance circuit according to the second aspect of the present invention is connected between an output terminal and ground, and The voltage controlled current source circuit according to claim 1 is connected between the output terminal and the input terminal by negative feedback connection.

An active filter circuit according to a fourth aspect of the present invention is arranged such that a series resonance circuit formed by connecting another MOS-type capacitor in series with the equivalent inductance circuit of the second aspect of the invention is connected between an output terminal and a ground, and The voltage controlled current source circuit according to claim 1 is connected between the output terminal and the input terminal by negative feedback connection.

[0019]

The voltage-controlled current source circuit according to the present invention uses the positive temperature coefficient of the first resistance of the first current source circuit as the negative temperature coefficient of the base-emitter voltage of the transistor of the second current source circuit. Is canceled by the difference between the value obtained by multiplying the predetermined magnification and the positive temperature coefficient by the second resistor. As a result, the temperature dependence of the input / output characteristics of the voltage-controlled current source circuit is eliminated and the temperature is compensated.The characteristics of the equivalent inductance circuit and the active filter circuit using the voltage-controlled current source circuit are also eliminated from the temperature dependence. Compensated.

[0020]

【Example】

[Regarding Voltage Controlled Current Source Circuit] An embodiment of the present invention will be described below. FIG. 1 is a circuit diagram showing a voltage controlled current source circuit (VCCS) 20 according to one embodiment of the present invention. The transistors Q1 and Q2 are emitter-follower transistors whose bases are connected to the input terminals 21 and 22 to form an input circuit. A transistor Q that inputs a current obtained by subtracting the emitter currents of the transistors Q1 and Q2 from the currents of the current source circuits I1 (current value I1) and I2 (current value I2).
3, Q4, an emitter resistor R14, R15, and a current source circuit I3 (current value I3) constitute a first differential amplifier circuit. Q5 and Q6 are diode-connected transistors as loads of the first differential amplifier circuit. Transistors Q7 and Q8 which use the output signal of the first differential amplifier circuit as a differential input signal, and a current source circuit I4 (current value I
The circuit consisting of 4) constitutes a second differential amplifier circuit. The diode-connected transistor Q9 is connected to the collector of one transistor Q7 of the second differential amplifier circuit, and the current mirror-connected transistors Q10 and Q11 are connected to the collector of the other transistor Q8. R16 to R18 are load resistors. A current source circuit I5 (current value I5) is connected to the collector of the transistor Q11, and a common connection point between the current source circuit I5 and the collector of the transistor Q11 is connected to the output terminal 23, where an output voltage Vo is obtained. .

In the voltage controlled current source circuit 20, when a differential input voltage Vi is applied between the input terminals 21 and 22,
Assuming that the voltage gain of the emitter followers of the transistors Q1 and Q2 is 1, the transistors Q3 and Q3 of the first differential amplifier circuit
4 is applied with the above-described input voltage Vi. Here, the transconductance Gm of the first differential amplifier circuit
When 0 is obtained by a π-type equivalent circuit, the result is as follows. Gm0 = gm / (1 + gm · R E) = 1 / [(1 / gm) + R E] ·· (11) However, gm: Q3, Q4 of the transconductance _{(= I3 / 2V T) V} T: thermal voltage R _E = R14 = R15

[0022] Therefore, the relationship of the transistors Q3, Q4 emitter resistor r _e (= 1 / gm) and the resistance R _E is, when the r _e << R _E, equation (11), Gm0 = 1 / R _E ··· (12), the collector current I of the transistors Q3 and Q4
The relationship between c3, Ic4, and Gm0 is as follows: I3 = Ic3 + Ic4 (13) Ic3-Ic4 = Gm0 Vi = Vi / R _E (14) Therefore, from the equations (13) and (14), Ic3 = [ I3 / 2] + [Vi / (2 · R _E)] ... (15) Ic4 = [I3 / 2] - a [Vi / (2 · R _E)] ... (16).

Assuming that the transistors Q5 and Q6 have the same emitter size, the base-emitter voltage V
_{EB 5} and V _{EB 6} are as follows when Is is a transistor reverse saturation current. _{V EB 5 = V T · ln} [Ic3 / Is] = V T · ln [ [I3 / 2 + Vi / (2 · R E) ] / Is] ·· (17) V EB 6 = V T · ln [Ic4 / Is] = V _T · ln [[I3 / 2-Vi / (2 · R E) ] / Is] ·· (18)

The transistor Q of the second differential amplifier circuit
7, the base potentials V _B7 and V _B8 of Q8 are _calculated by the equations (17) and (1).
Using 8), it becomes as follows. _{V B7 = Vcc-V EB 5} = Vcc- [V T · ln { [I3 / 2 + Vi / (2 · R E) ] / Is}] ·· (19) V B8 = Vcc-V EB 6 = Vcc- [ V _T · ln {[I3 / 2-Vi / (2 · R _E )] / Is}] (20)

Using these equations (19) and (20), the base-to-base voltage V _BB of transistors Q7 and Q8 is as follows. _{V BB = V B7 -V B8 =} V T · ln [{I3 / 2-Vi / (2 · R E)} / {I3 / 2 + Vi / (2 · R E)}] ·· (21)

The collector current of the transistor Q9 is -i _0, and the collector current of the transistor Q10 is + i
When the voltage V _BB between the bases of the transistors Q7 and Q8 when ₀ is obtained, the result is as follows. _{V BE 7 = V T · ln} [(I4 / 2-i 0) / Is] ·· (22) V BE 8 = V T · ln [(I4 / 2 + i 0) / Is] ·· (23) V BB _{= V BE 7 -V BE 8 =} V T · ln [(I4 / 2-i 0) / (I4 / 2 + i 0)] ·· (24)

Here, when the equations (21) and (24) are arranged, the input / output characteristics of the voltage controlled current source circuit 20, that is, the output current i _{0 with} respect to the input voltage Vi are as follows. _{i 0 / Vi = I4 / (} 2 · R E · I3) ·· (25)

[Integrator] When an internal capacitor Co (capacitance Co) is connected between the output terminal 23 of the voltage-controlled current source circuit 20 shown in FIG. 1 and the ground, an integrator is formed. From the above equation (25), the following is obtained. From Vo / Vi = I4 / (2 · S · Co · R E · I3) ·· (26) This equation (26), the integrator gain, a current source circuit I3,
It can be seen that the current is controlled by the ratio of the currents I3 and I4 of I4.

Next, the current source circuit I3 of the voltage controlled current source circuit 20 is realized by the circuit shown in FIG.
This circuit includes current mirror connection transistors Q12 and Q12.
13, a resistor R19 (resistance value R19) for determining a reference current. Vf is a reference voltage. The collector of the transistor Q13 becomes the current output terminal 24.

The collector current, i.e. the output current I3 of the transistor Q13 in this circuit, when the base-emitter voltage of the transistor Q12 and V _{BE 1 2} as follows. _{I3 = (Vf-V BE 1} 2) / R19 ·· (27) Therefore, substituting the equation (27) into equation (26), as follows. Vo / Vi = I4 / [2 · S · Co · R E · (Vf-V BE 1 2) / R19] ·· (28)

In the equation (28), the resistance values R _E and R
Since 19 is canceled its temperature coefficient, if Ae cancel the temperature coefficient and the current I4 term with the temperature dependence of _{(Vf-V BE 1 2)} , the gain of the integrator will be no temperature coefficient .

Therefore, first, (Vf-V _BE ) in the equation (28)
₁ ) Calculate the sensitivity to temperature in ₂ ). The voltage of the resistor R19 as V _{R1 9,} when put between _{(Vf-V BE 1 2)} = V R1 9, the sensitivity S _T is as follows. _{S T = (T / V R1} 9) (dV R1 9 /dT)=-0.15 ·· (29) which is meant to vary -0.15% per 1% temperature. Here, the collector current of the transistor Q12 is set to 1
0 .mu.A, the saturation current Is of e ^{-1 6} A, in which the Vf was obtained under the conditions as 5V, the correct temperature coefficient, 505ppm / ℃
Becomes

Next, a circuit as shown in FIG. 3 is used as the current source circuit I4 of the voltage controlled current source circuit 20. As will be described later, the above-mentioned (Vf
Cancel the temperature coefficient of -V _{BE 1 2).} Reference numeral 25 denotes another voltage-controlled current source circuit (as shown by a specific circuit in FIG. 4, which is different from the configuration shown in FIG. 1). Outputs the negative current. Q
14 is a transistor, R20 to R23 are resistors, and 26 is a current output terminal.

In the current source circuit I4, if the input offset voltage and the input bias voltage of the voltage controlled current source circuit 25 are ignored, the current value I4 output from the output terminal 26 becomes
When a voltage generated across the resistor R23 and V _{R2 3,} as follows. I4 = −V _{R2 3} / R23 (30) In this case, since the non-inverting output terminal (+) of the voltage control current source circuit 25 is connected to the inverting input terminal (−), a negative feedback configuration is adopted. , The voltage Va of the non-inversion input terminal (+) is V
a a = V _{R2 3,} therefore, the output current I4 is obtained by the following equation. I4 = −Va / R23 (31)

[0035] Now, in the voltage controlled current source circuit 20 shown in FIG. 1 of the present embodiment, as described by the current source circuit I4 to the next, cancel the temperature coefficient of _{(Vf-V BE 1 2)} . As described above, when the resistors R20 to R23 to be used are configured inside the integrated circuit, all the resistors have the same temperature coefficient. The resistance value of the resistor R23 (R
23), when the non-inverting input terminal (+) of the voltage control current source circuit 24 is at the fixed potential Va (when the temperature is compensated), the potential of the inverting input terminal (-) is also fixed at the same potential. since the temperature coefficient of the current I _{R2 3} flowing through the resistor R23 is equal to the resistance temperature coefficient.

[0036] Thus, the difference between the temperature coefficient of the voltage Va of the resistance temperature coefficient and a non-inverting input terminal of a resistor R23 (+) to match the temperature coefficient of the above _{(Vf-V BE 1 2)} , voltage controlled current by adjusting the voltage Va of the non-inverting input terminal of the source circuit 25 (+), the current flowing through the resistor R23 I _{R2 3} (= I4)
It is equal to the temperature coefficient of _{(Vf-V BE 1 2)} . (1 / Va) (dVa / dT) = (1 / R23) (dR23 / dR) (32)

[0037] Here, when the base-emitter voltage of the transistor Q14 and _{V BE 1 4, Va = Vcc} -nV BE 1 4 ·· (33) However, since it is n = (R20 + R21) / R21, (Vf to carry out the same change to the temperature coefficient of -V _{BE 1 2)} is, (1 / R23) (dR23 / dT) - (1 / nV BE 1 4) (dnV BE 1 4 / dT) = [1 / _{(Vf-V bE 1 2)} ] [d (Vf-V bE 1 2) / dT] wherein it is sufficient to satisfy the ... (34).

As described above, the voltage V of the transistor 14
_{BE 1} multiplied by a coefficient determined by the _fourth resistor R18, the negative temperature coefficient with the R19, to adjust the temperature coefficient of the current I4 by its value, by combining the temperature coefficient of _{(Vf-V BE 1 2)} , FIG. The temperature coefficient of the integrator shown in FIG. 1 can be canceled. The current source circuit I shown in FIG.
Since it is necessary to set I5 = I4 / 2 also for 5, the temperature coefficient is canceled by realizing the same circuit configuration.

In the specific circuit shown in FIG. 4, the same components as those in FIG. 3 are denoted by the same reference numerals. Transistors Q15 and Q16 constitute a differential circuit, transistors Q17 and Q18 constitute an active load of the differential circuit, transistor Q19 and transistor Q18 constitute a current mirror circuit, and transistors Q20 to Q24 constitute an output circuit. I do. I6 and I7 are current source circuits with currents of I6 and I7, and are set to I6 = I7 / 2.

Here, it is examined whether or not an error occurs in the output current I4 by the current source circuits I6 and I7.
An input bias current can be considered as a cause of such an error. First, when the differential circuit is in a balanced state,
Transistors Q24, Q25 of the base current (input bias current) I _{B2 4,} I _{B2 5,} the transistor Q24,
Assuming that the current amplification factor of Q25 is h _fe , the following is obtained. _{I B2 4 = I B2 5 =} I B = (I7 / 2) / h fe ·· (35)

[0041] Then, the input bias current I _B inverting input terminal as an error component (-) to the flow, since the output current I4 from the non-inverted output terminal (+) flows, the voltage V _{R2 3} of resistor R23, the following become that way. _{V R2 3 = R23 (I B} + I4) ·· (36)

Accordingly, the output current I4 is as follows. _{I4 = (V R2 3 -I B} · R23) / R23 = V R2 3 / R23-I B ·· (37) that is, the output current I4 will decrease by an error bias current I _B. However, since the bias current I _B is as shown in Formula (35), even at the I7 / 2 = I4, the error is 1 / h _fe is very small, negligible.

[Equivalent Inductance Circuit] FIG.
The equivalent inductance circuit constructed using the integrator whose temperature coefficient is compensated as described above is shown below. This is constituted by a gyrator that makes capacitance equivalent to inductance. In FIG. 5, 27, 28
Is a voltage controlled current source circuit having the same configuration as the voltage controlled current source circuit 20 shown in FIG.
1 and Gm2, each of which has a non-inverting input terminal (+) as shown in FIG.
Of the input terminal 21 and the inverted input terminal (−) correspond to the input terminal 22. An integrator is composed of the voltage control current source circuit 27 and the MOS internal capacitor Co. The reference voltage V _ref is applied to the non-inverting input terminal (+) of one voltage-controlled current source circuit 28 and the inverting input terminal (-) of the other voltage-controlled current source circuit 27.
Is applied. First, the input impedance Z viewed from the connection terminal 29 is obtained.

The current flowing through the connection terminal 29 is i ₁ , the current flowing through the capacitor Co is i ₂ , the voltage applied to the connection terminal 29 is v ₁ , and the voltage applied to the capacitor Co is v ₂
Then, since i ₂ = Gm ₁ · v ₁ v ₂ = i ₂ / (S · Co) i ₁ = Gm ₂ · v ₂ , Z = v ₁ / i ₁ = S · Co / (Gm 1 · Gm 2) The equivalent inductance L is as follows. L = Co / (Gm1 · Gm2) Here, if Gm1 = Gm2 = Gm, L = Co / Gm ² .

From the above, the above-described equivalent inductance L
And a MOS-type internal capacitor (separate from Co) to form a resonance circuit to form an active filter circuit. For example, a band-pass filter can be constituted by a parallel resonance circuit, and a band-elimination filter can be constituted by a series resonance circuit. At this time, a resistance element is required between the signal input terminal and the equivalent inductance L. However, the resistance created inside the integrated circuit has a poor absolute value accuracy of ± 10 to 20%, and is realized by the gyrator described above. Since the value of the equivalent inductance L also depends on 1 / Gm, it changes according to the current value I4 in the equation (26). Therefore, in order to keep the Q of the active filter formed constant, it is necessary to make the value of the resistance between the signal input terminal and the equivalent inductance L proportional to 1 / Gm.

[Regarding Bandpass Filter] FIG. 6 shows a bandpass filter configured as described above. In FIG. 6A, reference numeral 30 denotes a voltage-controlled current source circuit having a negative feedback configuration functioning as the resistor, and connects the output terminal and the inverting input terminal (-) of the voltage-controlled current source circuit 20 shown in FIG. The one having the configuration described above is used. 31 is an input terminal, 3
2 is an output terminal. Here, a MOS-type internal capacitor Ca is connected in parallel with the equivalent inductance L shown in FIG. 5 to form a band elimination filter.

Assuming that the input voltage is Vi and the output voltage is Vo,
Find its transfer function. First, the input impedance Z1 of the parallel resonance circuit viewed from the output terminal 32 is as follows. Z1 = S / L / (S ² · L · Ca + 1) (39) The output voltage Vo is as follows when the transconductance of the voltage control current source 30 is Gm3. Vo = Gm3 (Vi-Vo) · Z1 = Gm3 (Vi-Vo) · S · L / (S 2 · L · Ca + 1) ·· (40) Therefore, the transfer function is as follows. Vi / Vo = [S · Gm3 / Ca] / [S ² + S · Gm
3 / Ca + 1 / (L · Ca)]

Next, the general equation (2) showing the secondary bandpass function is compared with the equation (41), and the gain coefficient H =
As 1, when obtaining the center frequency ^{_{ω 0 2, Q, ω 0}} 2 = 1 / (L · Ca) ·· (42) Q = (Ca / L) 1/2 / Gm3 since the ... (43) By substituting the equation (38) for the equivalent inductance L into the equations (42) and (43), the following is obtained. ω ₀ ² = Gm ² / (Co · Ca) Therefore, the center frequency ω ₀ is as follows. _{ω 0 = Gm / (Co ·} Ca) 1/2 ·· (44)

The temperature change of the center frequency ω ₀ is as follows because the temperature coefficient of the MOS type capacitor in the integrated circuit is zero and the integrator gain is temperature compensated as described above. Become. For _{(dω 0 / dT) = 0} ·· (45) Q, with Q = (Gm / Gm3) ( Co / Ca) 1/2 ·· (46), a temperature change the center frequency omega ₀ of the Q Similarly, dQ
/ DT = 0.

FIG. 7 is a block diagram showing a case where the present invention is applied to a band elimination filter. FIG.
The same components as those in the first embodiment are denoted by the same reference numerals. Here, a capacitor Ca is connected in series to the equivalent inductance L. Since the input impedance Z2 of the series resonant circuit of the circuit, Z2 = a ^{(S 2 · L · Ca +} 1) / (S · Ca) ·· (47), the output voltage Vo, Vo = Gm3 (Vi-Vo ) · Z2 = Gm3 (Vi-Vo ) · (S 2 · L · Ca + 1) / become (S · Ca) ·· (48 ). Therefore, the input / output function is as follows: Vi / Vo = [S ² + 1 / (L · Ca)] / [S ² + S / (Gm3 · L) + 1 / (L · Ca)] (49)

On the other hand, the secondary band elimination function is given by the following general formula. Vi / Vo = compared _{^{[S 2 + ω N 2]}} / [S 2 + S · ω N / Q N + ω N 2] ·· (50) 2 -order band elimination function equations (49) to (50),
When determining the center frequency omega _N ² and Q, the ^{_{ω N 2 = 1 / (L}} · Ca) ·· (51) Q = Gm3 · (L / Ca) 1/2 ·· (52). Substituting the equation (38) for the equivalent inductance L into the equations (51) and (52) yields the following. _{^{ω N 2 = Gm 2 / (}} Co · Ca) ·· (53) ω N = Gm / (Co · Ca) 1/2 ·· (54)

[0052] Temperature changes of the center frequency omega _N is the temperature coefficient of the integrated circuit inside the MOS capacitor is zero, the integrator gain is compensated as in the above description, dω _N / dT = 0 ·· (55), and the Q of the filter is as follows. Temperature change of Q = (Gm3 / Gm) / (Co · Ca) 1/2 ·· (56) The Q, like the well center frequency d [omega _N as follows. dQ / dt = 0

[0053]

As described above, the output current of the voltage controlled current source circuit of the present invention is temperature-compensated and is not affected by temperature. Therefore, the integrator formed by the voltage-controlled current source circuit and the MOS-type capacitor in the integrated circuit is also temperature-compensated, and the equivalent inductance circuit formed by the integrated circuit and another voltage-controlled current source circuit is also temperature-compensated. Is done. Therefore, the resonance circuit created by this equivalent inductance circuit and another MOS-type capacitor inside the integrated circuit is also temperature-compensated, and the active filter circuit formed by using these circuits is also temperature-compensated. An excellent filter circuit can be realized.

[Brief description of the drawings]

FIG. 1 is a circuit diagram of an integrator combining a voltage-controlled current source circuit and a capacitor according to an embodiment of the present invention.

FIG. 2 is a block diagram of a current source circuit I3 of FIG.

FIG. 3 is a block diagram of a current source circuit I4 of FIG.

FIG. 4 is a specific circuit diagram of the current source circuit I4 of FIG.

FIG. 5 is a block diagram of an equivalent inductance formed using the integrator of FIG. 1 and another capacitor.

6A is a block diagram of a bandpass active filter circuit using the equivalent inductance of FIG. 5, and FIG. 6B is an equivalent circuit diagram thereof.

7A is a block diagram of a band elimination active filter circuit using the equivalent inductance of FIG. 5, and FIG. 7B is an equivalent circuit diagram thereof.

FIG. 8 is a block diagram of a conventional Sallen-Key circuit.

FIG. 9 is a block diagram of a conventional Biquad filter circuit.

[Explanation of symbols]

1: input terminal, 2: output terminal, 3 to 7: amplifier, 8: input terminal, 9 to 11: output terminal, 20: voltage controlled current source circuit, 21: forward input terminal, 22: inverting input terminal, 23 ,
24: output terminal, 25: voltage-controlled current source circuit, 26: inverted output terminal, 27, 28: voltage-controlled current source circuit, 29:
Connection terminal, 30: voltage controlled current source circuit, 31: input terminal, 32: output terminal, I3: first current source circuit, 14:
A second current source circuit;

──────────────────────────────────────────────────続 き Continuation of the front page (56) References JP-A-3-14311 (JP, A) JP-A-62-157409 (JP, A) (58) Fields investigated (Int. Cl. ⁷ , DB name) H03H 11/04 H03H 11/54 H03F 3/45

Claims (4)

(57) [Claims] A first differential amplifier circuit having a first non-inverting input terminal and a second inverting input terminal, the operating current of which is determined by a first current source circuit; and a second current source circuit. An operating current is determined, and a second differential amplifier circuit inputs a differential output signal of the first differential amplifier circuit as a differential input signal. The first current source circuit and the second differential amplifier circuit A voltage-controlled current source circuit for controlling a gain and compensating for a temperature characteristic by a current ratio of a current source circuit, wherein the first current source circuit is based on a current flowing from a fixed voltage source to a first resistor. A current mirror circuit which transfers the current to the output side as a side current, wherein the second current source circuit applies to the second non-inverting input terminal a voltage that is a predetermined rate times the base-emitter voltage of the transistor; Connecting the second inverting input terminal to the non-inverting output terminal, Second resistor connected and configured in sub-voltage control current source circuit to the output terminal of the inverting output terminal, the voltage controlled current source circuit, characterized in that between the ground and the inverting input terminal. 2. A voltage controlled current source circuit according to claim 1, wherein two of said voltage controlled current source circuits are provided as a first voltage controlled current source circuit and a second voltage controlled current source circuit. A common input terminal and an output terminal of the second voltage controlled current source circuit are connected in common, and an output terminal of the first voltage controlled current source circuit and an inverted input terminal of the second voltage controlled current source circuit are connected. An equivalent inductance circuit, which is commonly connected to one end of a MOS capacitor. 3. A parallel resonance circuit comprising another equivalent MOS type capacitor connected in parallel with the equivalent inductance circuit according to claim 2 between an output terminal and ground, and between the output terminal and the input terminal. An active filter circuit comprising a voltage-controlled current source circuit according to claim 1 and a negative feedback connection. 4. A series resonance circuit in which another MOS type capacitor is connected in series with the equivalent inductance circuit according to claim 2, connected between an output terminal and ground, and between the output terminal and the input terminal. An active filter circuit comprising a voltage-controlled current source circuit according to claim 1 and a negative feedback connection.