JPH0655030B2 - Instantaneous value control method for load current - Google Patents

Description

The present invention relates to a switching means interposed between a power supply and a load, a means for detecting an instantaneous value of a load current, a detected load current instantaneous value and a current command value, and both are compared. A comparator for outputting a high-level or low-level signal depending on the magnitude relationship between the switching means and a means for driving the switching means on / off by the high-level or low-level signal are provided. The present invention relates to a method for instantaneous value control of a load current, which repeatedly rises and falls between an upper limit level and a lower limit level determined by a circuit constant of a comparator with respect to a current command value.

FIG. 1 is a circuit diagram showing an example of a conventional load current instantaneous value control method. In the figure, 1 is a DC power source, 2 is a switching element, 3 is a freewheeling diode, 4 is a reactor, 5 is a load that generates a back electromotive force such as a DC motor, 6 is a load current detector, 7 is a comparator, Reference numeral 8 is a current command value generator, R _i is an input resistance, and R _f is a feedback resistance.

FIG. 1A is a timing chart showing the changing state of the load current i when the current command value i ^* output from the generator 8 is maintained at a constant value in FIG.

The operation will be described with reference to FIGS. 1 and 1A. First
In the figure, the output voltage of the DC power supply 1 is V _DC . now,
If the switch element 2 is turned on, a current i flows from the DC power supply 1 to the load 5 via the switch element 2 and the reactor 4, and the load 5 generates a counter electromotive force E.

The current command value generator 8 outputs the command value i ^* of the load current i. The instantaneous value (actual value) i of the load current detected by the detector 6 is applied to the non-inverting input terminal (+) of the comparator 7 via the input resistor R _i , while the command value i ^* is applied to the inverting input terminal ( -) Is applied. The comparator 7 compares the two, and as shown in FIG. 1A, when the instantaneous value i reaches the level U by increasing the command value i ^* by Δi, its output changes from the low level to the high level. Invert. As a result, the switch element 2 is turned off. Then, the load current i becomes load 5 → diode 3 →
The closed circuit of reactor 4 → load 5 is circulated and attenuated. Then, when the instantaneous value i of the load current reaches the level L by lowering the command value i ^* by Δi as shown in FIG. 1A, the comparator 7 changes its output from the high level until then to the low level. Turn around. As a result, the switch element 2 is turned on, and the current is again supplied from the load power source 1 to the load 5
And the load current i increases.

In this way, by turning the switch element 2 on and off, the load current i is repeatedly controlled to rise and fall within a range of the command value i ^* of ± Δi and is stably controlled. The range (± Δi) between the upper limit level U and the lower limit level L is determined in relation to each resistance value of the feedback resistance R _f and the input resistance R _i , which is well known.

The conventional instantaneous value control system has the following problems. That is, the cycle in which the instantaneous value i of the load current repeatedly rises and falls between the upper limit level U and the lower limit level L (in other words, the switch frequency at which the switch element 2 repeatedly turns on and off) is the DC voltage V of the DC power supply 1. It varies significantly according to the relationship between _DC and the back EMF of the load 5. This will be described below in an easy-to-understand manner.

Now, assuming that the inductance of the reactor 4 is L, the differential value (rate of change over time) di / di of the load current i when the switch element 2 is on is given by the following equation.

That is, as is apparent from the above equation (1), even when the power supply voltage V _DC is constant, when the magnitude of the back electromotive force E changes, the instantaneous current value i rises toward the upper limit level U. The slope (di / dt) changes.

Similarly, when the switch element 2 is off, the slope at which the current instantaneous value i drops toward the lower limit level L (this is given by setting V _DC to zero in the above equation (1)), When the magnitude of the counter electromotive force E changes, it also changes. This means that the switching frequency of the switching element 2 changes depending on the magnitude of the back electromotive force E.

Turning on / off the switch element 2 causes noise to other devices. If the switching frequency of the switching element is constant, the frequency of the generated noise will also be constant, and it is easy to take measures such as by preparing a filter of that frequency, but the switching frequency changes and the noise frequency also changes to that. Therefore, when it changes, it becomes difficult to take measures against noise.

In order to improve such a problem, in a load current instantaneous value control system, a takt pulse composed of a pulse train having a certain repetition frequency is superimposed on either one of the load current instantaneous value and the current command value, and the comparator is used. The present inventors have previously proposed a control method in which the switch frequency of the switch element is pulled into the repetition frequency of the tact pulse to synchronize the switch frequency of the switch element.
7-184677).

Next, the outline of this proposal will be described.

FIGS. 2 to 4 are explanatory views showing the influence of the magnitude relationship between the DC power supply voltage V _DC and the back electromotive force E due to the load on the switch frequency.

In FIG. 2 (a), when the DC power supply voltage V _DC is large and the counter electromotive force E is smaller than 1/2 thereof, the rising slope of the current i is large as shown by R and the falling slope is small as shown by F. . Therefore, the state in which the current i repeatedly rises and falls between the upper limit U and the lower limit L around the command value i ^* is shown in FIG.
It will be understood that the switch frequency tends to decrease as shown in (b).

FIG. 3 is an explanatory view similar to FIG. 2, but in this case, as shown in FIG. 3 (a), the counter electromotive force E is approximately 1/2 of the DC power supply voltage V _DC. Therefore, the rising slope R and the falling slope F of the current i are the same. At this time, as seen in FIG. 3B, the frequency (switch frequency) at which the current i repeatedly rises and falls between the upper limit U and the lower limit L becomes maximum.

FIG. 4 shows that when the counter electromotive force E is larger than 1/2 of the DC power supply voltage V _DC as shown in (a), the rising slope R of the current i is increased.
Is small and the descending slope F is large. Therefore, the state in which the current i repeatedly rises and falls between the upper limit U and the lower limit L is (b)
In this case, the switch frequency also becomes small as in the case of FIG.

It will be understood from the above that even if the DC power supply voltage V _DC is constant, if the magnitude of the back electromotive force E changes, the switch frequency changes accordingly. Next, the operating principle of the proposed instantaneous value control method will be described with reference to FIGS.

FIG. 5 (a) is a waveform diagram in which the waveform shown in FIG. 2 (b) is enlarged as it is, in other words, when the counter electromotive force E is 1/2 or less of the DC power supply voltage V _DC . It is a wave form diagram which shows the change of the electric current i. Figure 5 (b) is based on the already proposed principle.
It is a waveform diagram showing a pulse train (tact pulse) to be superimposed on the waveform of the current i shown in (a). It will be understood that the tact pulse is composed of positive-polarity pulses P _{1 to} P ₃ and negative-polarity pulses N _{1 to} N ₃ having a constant cycle.

FIG. 5C shows the waveform of the current i shown in FIG.
FIG. 7 is a waveform diagram showing how the current i is inverted when the tact pulse shown in FIG.

In FIG. 5 (c), the current i passes through the lower limit L by superimposing the negative polarity pulse N ₁ at time t ₁ while following the descending process.
Invert immediately. After the reversal, the level of the current i continues to rise, and at the time t ₂ , it coincides with the upper limit U, so that it reverts and starts the descending process. Then, at time t ₃ , the positive pulse P ₁ is superposed, but since it is in the descending process at this time,
Even if the current level exceeds the upper limit U due to the superposition of the positive polarity pulse P ₁ , inversion does not occur.

Next, at time t ₄ , when the negative polarity pulse N ₂ is superimposed, the level of the current i exceeds the lower limit L, so the current i is inverted. Similarly, the waveform of the current i is as if
Repeat the reversal as if you were stuck to the upper limit U. The repetition cycle of this inversion is the negative polarity pulse N _{1 to} N ₃
It will be easily understood that it is synchronized with the repetition cycle of.

FIG. 6 is a waveform diagram similar to FIG. 5, but FIG. 6 (a)
Is a waveform diagram in which the waveform shown in FIG. 4B is enlarged as it is, in other words, the counter electromotive force E is 1 / _DC of the DC power supply voltage V _DC .
It is a wave form diagram which shows the change of the electric current i when it is 2 or more.
FIG. 6 (c) shows the reversal situation of the current i when the tact pulse shown in FIG. 6 (b) is superimposed on the waveform shown in FIG. 6 (a).
Will be described with reference to.

In FIG. 6 (c), the current i in the ascending process is changed to the time t ₁
When the positive polarity pulse P ₁ is superposed, the current level exceeds the upper limit U, and the current i immediately reverses and starts the descending process. Next, at time t ₂ , the current i
Since the level of reaches the lower limit L, it reverses again and starts the rising process. At time t ₃ , the negative polarity pulse N ₁ is superposed and the current level exceeds the lower limit level, but the current is in the process of rising, so it is not reversed. Then at time t ₃ ,
When the positive polarity pulse P ₂ is superposed, the current level exceeds the upper limit U, so that the current i in the ascending process is immediately inverted and enters the descending process. Similarly, the waveform of the current i is
The reversal is repeated as if the lower limit L was stuck. It will be easily understood that the repetition cycle of this inversion is synchronized with the repetition cycle of the positive polarity pulses P _{1 to} P ₃ .

As described above, when the magnitude of the back electromotive force E is 1/2 or less of the power supply voltage V _DC , the negative tact pulse effectively acts, and when it is 1/2 or more, the positive polarity is positive. Tact pulse of effectively works. The frequency of the tact pulse is
It is preferable to set the switch frequency slightly higher than the expected maximum value of the switch frequency (see FIG. 5 (a) or FIG. 6 (a)) which is formed when the tact pulse is not applied.

FIG. 7 is a circuit diagram showing a proposed instantaneous value control method based on the above principle. In the figure, the difference from the conventional circuit configuration shown in FIG. 1 is that a tact pulse is input from the tact pulse generator 9 to the non-inverting input terminal (+) of the comparator 7 via the input resistor R _i , This is a point that the current instantaneous value i detected by the detector 6 is superimposed. There is no other difference.

The circuit operation will no longer need to be explained. The tact pulse from the tact pulse generator 9 has the same result even if it is superimposed not on the non-inverting input terminal (+) of the comparator 7 but on the inverting input terminal (-), that is, the current command value i ^*. Is obtained.

By the way, the instantaneous value control method according to the already proposed as described above is
The magnitude of the back electromotive force E due to the load is 1/2 of the power supply voltage V _DC .
When it is more than or equal to 1/2 or less, the switch frequency is pulled in to the frequency of the tact pulse and is well synchronized, At the same time, that is, when the rising slope and the falling slope of the load current i are almost the same as shown in FIGS. 3 (a) and 3 (b), the current i is set to either the upper limit U or the lower limit L. Since it does not tend to cling to either, it may not cling to either and the switch frequency may not be synchronized with that of the tact pulse. If the width between the upper limit U and the lower limit L is set too wide, a dead zone occurs in which the current instantaneous value i does not follow the change in the current command value i ^*, which causes a problem of poor control response.

On the other hand, if the width between the upper limit U and the lower limit L is made too small, the switch element will be turned on and off at an irregular frequency exceeding the tact frequency.

In any case, like this, In this case, the proposed instantaneous value control method has a drawback in that the switch frequency is difficult to be drawn into that of the tact pulse.

The present invention has been made to solve the above-mentioned drawbacks of the prior art, and accordingly, the object of the present invention is to At the same time, it is another object of the present invention to provide a method for controlling the instantaneous value of the load current, which can easily pull the switch frequency into that of the tact pulse.

The essential point of the configuration of the present invention is that a tact signal including at least a sawtooth wave that repeats at a constant cycle is used as the tact pulse.

Next, the operating principle of the present invention will be described with reference to the drawings. FIG. 8 is an explanatory diagram of the operating principle of the present invention. The same figure (a) is shown in FIG.
The same waveform diagram as (b), in other words, back electromotive force FIG. 8 (b) is a waveform diagram showing the change of the current i when it is (DC power supply voltage V _DC ), FIG. 8 (b) is a waveform diagram of the tact signal tact used according to the principle of the present invention, and FIG. 8 (c) is , Tact signal tac
Upper limit U = (i when t is superimposed on the current command value i ^*
* + Tact + ^Δi) and a waveform diagram showing the lower limit L = waveforms ^{(i * + tact-Δi)} .

In FIG. 8 (c), the current i reverses from rising to falling at the k ₁ point, reverses from falling to rising at the k ₂ point, and then k ₃ , k ₄ ...
It will be seen that similar inversions are repeated at each point of. That is, the current i repeats the inversion in the form of clinging to the lower limit L, so that the switch frequency is synchronized with the frequency of the tact signal tact.

FIG. 9 is a circuit diagram showing an embodiment of the present invention. The circuit configuration shown in the figure is different from the previously proposed method shown in FIG. 7 in that the waveform of the tact signal output from the tact pulse generator 9A is different, and the tact signal is different from the current command value i. ^It is added to ^* , there is no other change.

The circuit operation of this embodiment will no longer be described.

FIG. 10 is a circuit diagram showing another embodiment of the present invention. The figure shows an embodiment in which the present invention is applied to a load circuit capable of reversible regeneration such as four-quadrant operation of a DC machine (back electromotive force load 5).

In the figure, 2A to 2D are switch elements and 7 respectively.
A and 7B are comparators, 13A to 13D are switch element driving circuits, 14 is a dead band delta setter, 15 is a sign inverter, and 16 to 19 are adders, respectively.
Is.

The operation of the four switch elements 2A to 2D will be briefly described. For example, when it is desired to rotate the load 5 in the forward direction, the lower right switch element 2D is turned on to connect the negative electrode of the load 5 to the negative electrode of the DC power supply 1, and the upper left lower switch elements 2A and 2B, that is, the left arm are switched. Then, the switching control as shown in FIG. 8C is performed. Therefore, when it is desired to reverse the load 5, the switch 2B under the left arm is turned on, and the right arms 2C, 2D switch up and down to control the current. In this way, it is easier to understand that the arm that controls the forward rotation and the reverse rotation is determined, and the control can be smoother than the case where the left and right arms control the switching simultaneously.

However, it is a problem which arm should switch when the speed is neither normal rotation nor reverse rotation near zero. It is the role of the dead band δ setter 14 to adjust this. When the dead zone δ is increased to a positive value, a bias as the dead zone δ is given to the inputs of the comparators 7A and 7B through the adders 19, 17 and 18, so that the current command value i ^* and the actual current value i.
If there is a certain degree of coincidence (if they are close to each other), the two comparators 7A and 7B do not operate together. That is, the switches 2A to 2D are likely to be in a state where both the left arm and the right arm are at rest. Conversely, when the dead zone δ is made negative, the two comparators 7A and 7B are both in the operating state, and the left and right switches 2A to 2D are always in the switching state, so that the dead zone in the control is eliminated and the control performance is improved, but the extra switching is performed. As a result, loss and noise will increase. It is considered that such a dead zone δ is taken for granted, but when the matter regarding the waveform of the tact signal and the hysteresis width Δi come to be related to each other with respect to the instantaneous value control in which the hysteresis width Δi is only set, the It is possible to say that this is a relatively important item in this sense, since it becomes possible to adjust and absorb the error of (5) in the dead zone δ.

FIG. 10A is a circuit diagram showing a modified embodiment of the main part in FIG. That is, a tact signal generator 9B shifted by a half cycle is provided in addition to 9A, and the tact signal is changed in electrical angle 18
This is an example in which the left arm and the right arm are allotted to each other by shifting them by 0 °.

In this way, by eliminating the dead zone or making it a little negative in the above sense to put the two comparators 7A and 7B in the operating state, the switching cycle T of each arm does not change and T is a correction operation in control. Since it is performed every / 2, there is an advantage that the response speed of the system is doubled. That is, by adding the second tact signal generator 9B to the embodiment shown in FIG. 10 as shown in FIG. 10A, smoother and faster control becomes possible.

FIG. 11 is a circuit diagram showing still another embodiment of the present invention. That is, it is an embodiment in which the present invention is applied to the instantaneous value control of the three-phase current of a three-phase load such as a three-phase AC motor. In the figure, 5A to 5C are three-phase back electromotive force loads, and 4A to 4C.
C is a three-phase reactor, 2A to 2F are three-phase switch element groups, 13A to 13F are three-phase drive circuit groups, 7A to 7C are three-phase comparator groups, and 19 to 22 are adders, respectively.
Is.

Conventional three-phase current instantaneous value control cannot be expected to be smooth because it is strongly influenced by the switching of other arms and the switching frequency is originally random. However, due to the advantageous condition that the switching frequency is constant due to the effect of the tact signal and one or two of the three arms is actively stopped due to the effect of the dead zone δ, relatively stable and smooth control can be performed.

FIG. 11A is a circuit diagram showing a modified embodiment of the main part in FIG. That is, as a tact signal generator, 1 /
Generators 9A and 9 for generating tact signals that are shifted by 3 cycles
B and 9C are provided and assigned to each phase current. This does not change the performance of the main circuit switching element,
The control performance of the system can be improved. That is, the control cycle is substantially shortened to T / 3, the response becomes faster, and the control becomes smoother.

As described above, according to the present invention, the relationship between the back electromotive force E due to the load and the power supply voltage V _DC is Even in the case of the relationship, there is an advantage that it is possible to provide a load current instantaneous value control method capable of easily pulling the switch frequency to that of the tact pulse and synchronizing it.

It goes without saying that the present invention is also applicable to a voltage integration type instantaneous value control method and the like.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing a conventional load current instantaneous value control method,
FIG. 1A is a timing chart showing an example of the changing situation of the load current in the circuit of FIG. 1, and FIGS. 2 to 4 show the relationship between the DC power supply voltage V _DC and the back electromotive force E by the switch frequency. Explanatory diagram showing the effect on
FIGS. 5 and 6 are explanatory diagrams of the operating principle of the already proposed instantaneous value control method, FIG. 7 is a circuit diagram showing the already proposed instantaneous value control method, and FIG. 8 is an explanatory diagram of the operating principle of the present invention. ,
FIG. 9 is a circuit diagram showing an embodiment of the present invention, FIG. 10 is a circuit diagram showing another embodiment of the present invention, FIG. 10A is a circuit diagram showing a modified embodiment of the main part of FIG. FIG. 11 is a circuit diagram showing still another embodiment of the present invention, and FIG.
FIG. 9 is a circuit diagram showing a modified example of the main part in FIG. 1. Symbol description 1 ... DC power supply, 2 ... Switch element, 3 ... Freewheeling diode, 4 ... Reactor, 5 ... Back electromotive force load, 6 ... Current detector, 7 ... Comparator, 8
... Current command value generator, 9 ... Tact pulse generator, 1
0 ... Integrator, 11 ... Smoothing capacitor, 12 ... Voltage command value generator, 13 ... Switch element drive circuit, 14 ...
... dead band delta setter, 15 ... sign inverter, 16-22
...... Adder

 ─────────────────────────────────────────────────── ─── Continuation of the front page (56) Reference JP-A-53-101650 (JP, A) JP-A-59-76195 (JP, A) Practical application Sho-49-150334 (JP, U)

Claims (1)

[Claims] 1. A switching means interposed between a power source and a load, a means for detecting an instantaneous value of a load current, and a detected load current instantaneous value and a current command value are compared, and a high value is obtained depending on a magnitude relation between the two. A comparator for outputting a level or low level signal; and a means for driving the switching means on / off by the high level or low level signal, wherein the load current is set to the current command value by turning on / off the switching means. On the other hand, in the instantaneous value control method of the load current that repeatedly rises and falls between the upper limit level and the lower limit level determined by the circuit constant of the comparator, in any one of the load current instantaneous value and the current command value, A means for superimposing a tact signal including at least a sawtooth wave repeated at a constant period and inputting it to the comparator is provided, Thereby, the method for controlling the instantaneous value of the load current is characterized in that the ON / OFF cycle of the switching means is pulled in and synchronized with the repetition cycle of the tact signal.