JP2006141192A - Inductive load drive device and inductive load drive method - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To smooth a current supplied from power supply, and to further reduce the frequency where the larger current is supplied from the power supply, even in case where two motors are driven independently such as in positioning control. <P>SOLUTION: A first PWM control signal generating portion 36, and a second PWM control signal generating portion 37 generate a first and a second PWM control signals, based on sawtooth wave carriers C, D whose phase difference is 1/2 wavelength. These two DC motors are made to operate with the first and the second PWM control signals, by which since the timing of the current inflow to the two DC motors from the power supply will not overlap, as long as one of duty ratios does not exceed 50%, even in case where the both duty ratios change at each moment, the current supplied from the power supply is smoothed, and the frequency where the larger current is supplied from the power supply is reduced significantly. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to an inductive load driving device and an inductive load driving method for controlling power supplied to two inductive loads.

Conventionally, a pulse width modulation control (hereinafter referred to as PWM control) using a so-called H bridge is generally used as a drive circuit for normal rotation inversion of an inductive load (eg, a motor with a DC brush). Depending on the system, it is necessary to control the two DC motors independently, in which case two pairs of H bridges are provided.
In the case of driving a plurality of loads by PWM control, several techniques have been proposed to reduce harness capacity, connector contact capacity, noise filter, shunt resistance rating, and the like.

JP 2002-43910 A assumes a resistive load, and when two loads are simultaneously driven by PWM control, the fall of one PWM control signal and the rise of the other PWM control signal By matching the timing and the timing of the rise of one PWM control signal with the fall of the other PWM control signal, the increase and decrease in load current flowing from the power supply to the load are offset and the rate of change is reduced. It is what.
Japanese Patent Application Laid-Open No. 2004-72592 discloses the fall start timing of one PWM control signal and the other PWM signal in order to solve the problem when the apparatus of Japanese Patent Application Laid-Open No. 2002-43910 is applied to an inductive load. There is described a configuration in which the rising end timing of the control signal is matched, and the rising end timing of one PWM control signal is matched with the falling start timing of the other PWM control signal.
JP 2002-43910 A JP 2004-72592 A

  However, in such a conventional load driving apparatus, when the duty ratio of the PWM control signal is fixed (for example, the duty ratio of two DC motors is fixed to 50%), the power source (for example, battery) is used. The current supplied to the drive circuit is smoothed to keep the upper limit of the current value low, but the two motor currents change from moment to moment, such as positioning control by the motor, that is, the two motors are controlled. There is a problem that it is not suitable when applied to a drive device in which the duty ratio of the PWM control signal changes every moment.

  Therefore, in view of such a problem, the present invention is a case where two motors are independently driven by positioning control or the like, that is, a case where the duty ratio of PWM control signals of both motors changes every moment. Another object of the present invention is to provide an inductive load driving device and an inductive load driving method capable of smoothing a current supplied from a power source and greatly reducing the frequency of supplying a large current from the power source. .

  In the present invention, since two inductive loads connected to a power source are independently driven by an H-bridge circuit using a pulse width modulation control method, a first pulse width modulation control signal generator and a second pulse width modulation Each of the control signal generators generates a modulation control signal by comparing the duty ratio in the pulse width modulation control and the sawtooth carrier wave, and the sawtooth carrier wave used by each has the same frequency, The phase of the sawtooth carrier wave and the other sawtooth carrier wave are shifted by ½ wavelength.

  According to the present invention, even when the two motors are independently driven by positioning control, for example, even when the duty ratio of both of them changes every moment, the current supplied from the power source is smoothed. And the frequency with which a large current is supplied from the power supply can be greatly reduced.

Next, embodiments of the present invention will be described by way of examples.
In each of the embodiments described below, the present invention is applied to an inductive load driving device that drives a DC motor that is independently positioned and controlled.
First, the first embodiment will be described.
FIG. 1 shows the overall configuration of the first embodiment.
The voltage supplied from the power source 1 to the control unit 2 is applied to the H bridge circuits 5 and 6 via the filter 3 and the current detection resistor 4.
The H bridge circuit 5 and the H bridge circuit 6 are configured by field effect transistors (FETs) 7 to 10 and field effect transistors (FETs) 11 to 14, respectively, and DC motors 15 and 16 are combined by switching combinations of the field effect transistors. In contrast, a current for rotating the motor forward or backward is supplied.

The first switching element drive circuit 17 that drives the H-bridge circuit 5 applies a drive voltage to the gate terminals of the field effect transistors 7 to 10 based on a command from the microcomputer 19. Similarly, the second switching element drive circuit 18 that drives the H-bridge circuit 6 applies a drive voltage to the gate terminals of the field effect transistors 11 to 14 based on a command from the microcomputer 19.
The microcomputer 19 detects motor rotation angle command values of the DC motors 15 and 16 specified by an external command unit (not shown) and motor rotation angle sensors 22 and 23 for detecting the rotation angles of the DC motors 15 and 16. And the motor current value signals output from the motor current sensors 24 and 25 for detecting the motor current flowing to the DC motors 15 and 16, and the field effect transistors 7-14. Is generated and applied to the first switching element driving circuit 17 and the second switching element driving circuit 18.

Next, an internal process of the microcomputer 19 that generates the PWM control signal will be described.
FIG. 2 is a diagram showing an internal configuration of the microcomputer, and FIG. 3 shows a PWM control signal obtained from the carrier wave and the duty ratio.
The microcomputer 19 reads the motor rotation angle command value for the DC motor 15 output from the command unit (not shown) by the communication unit 26 and reads the motor rotation angle command value for the DC motor 16 by the communication unit 27.
The motor rotation angle signal output from the motor rotation angle sensor 22 (see FIG. 1) for detecting the motor rotation angle of the DC motor 15 is read by the analog signal input unit 28, and the motor rotation for detecting the motor rotation angle of the DC motor 16 is read. The motor rotation angle signal output from the angle sensor 23 (see FIG. 1) is read by the analog signal input unit 30.

The motor current value signal output from the motor current sensor 24 is read by the analog signal input unit 29, and the motor current value signal output from the motor current sensor 25 is read by the analog signal input unit 31.
As described above, the communication unit 26 and the analog signal input units 28 and 29 read signals related to the DC motor 15, and the communication unit 27 and the analog signal input units 30 and 31 read signals related to the DC motor 16.

The motor rotation angle command value read by the communication unit 26 and the motor rotation angle signal read by the analog signal input unit 28 are input to the positioning servo unit 32.
The positioning servo unit 32 calculates the motor current command value so that the actual motor rotation angle of the DC motor 15 follows the motor rotation angle command value.
Similarly, the motor rotation angle command value read by the communication unit 27 and the motor rotation angle signal read by the analog signal input unit 30 are input to the positioning servo unit 33, and the positioning servo 33 performs the actual operation of the DC motor 16. The motor current command value is calculated so that the motor rotation angle follows the motor rotation angle command value.

The motor current command value calculated by the positioning servo unit 32 and the motor current value read by the analog signal input unit 29 are input to the current control unit 34.
The current control unit 34 performs duty control A (Duty ratio) for PWM control of the DC motor 15 so that the actual motor current value flowing through the DC motor 15 follows the motor current command value calculated by the positioning servo unit 32. A = 0 to 1) is calculated.
Similarly, the motor current command value calculated by the positioning servo unit 33 and the motor current value read by the analog signal input unit 31 are input to the current control unit 35, and the current control unit 35 actually flows to the DC motor 16. The duty ratio B (Duty ratio B = 0 to 1) for PWM control of the DC motor 16 is calculated such that the motor current value follows the motor current command value calculated by the positioning servo unit 33.
Thereby, the duty ratios A and B which fluctuate as shown in FIG. 3 are obtained.

As shown in FIG. 3, the carrier wave generation unit 40 generates a sawtooth carrier wave C having a frequency of 10 kHz and an amplitude of 1, for example, and outputs it to the first PWM control signal generation unit 36 and the phase adjuster 41.
The first PWM control signal generator 36 generates a first PWM control signal S1 for driving the DC motor 15.
Specifically, the value of the duty ratio A input from the current control unit 34 and the sawtooth wave carrier C input from the carrier generation unit 40 by the comparator 38 provided in the first PWM control signal generation unit 36. As shown in FIG. 3, when the duty ratio A is greater than or equal to the sawtooth wave carrier C, a high level is output as the first PWM control signal S1. To do.
The first PWM control signal S1 generated by the first PWM control signal generation unit 36 is output from the microcomputer 19 toward the first switching element drive circuit 17.

The phase adjuster 41 generates a sawtooth carrier wave D (see FIG. 3) in which the sawtooth carrier wave C input from the carrier wave generator 40 is shifted by a half wavelength, that is, by 50 μsec.
The phase adjuster 41 outputs the generated sawtooth carrier wave D to the second PWM control signal generator 37.
The second PWM control signal generation unit 37 generates a second PWM control signal S2 for driving the DC motor 16.
Specifically, the value of the duty ratio B input from the current control unit 35 and the sawtooth carrier wave D input from the phase adjuster 41 by the comparator 39 provided in the second PWM control signal generation unit 37. As shown in FIG. 3, when the duty ratio B is equal to or higher than the sawtooth wave carrier D, a comparator output that is at a high level and when the duty ratio B is less than the sawtooth wave carrier D is output as the second PWM control signal S2. To do.
The second PWM control signal S2 generated by the second PWM control signal generation unit 37 is output from the microcomputer 19 toward the second switching element drive circuit 18.

The first switching element driving circuit 17 and the second switching element driving circuit 18 operate the DC motors 15 and 16 by driving the H bridge circuits 5 and 6 based on the input first and second PWM control signals S1 and S2. Let
Here, when driving an inductive load by PWM control, when the switching element of the H bridge circuit is in the ON state, a current flows from the power source to the load, and the switching element of the H bridge circuit is in the OFF state. In the case (when one of the pair of ON-state switching elements is turned OFF), the current flows back through the H-bridge circuit by the reactance component of the inductive load, and the current does not flow from the power source.
Therefore, by setting the phase difference between the sawtooth carriers C and D for generating the PWM control signal to ½ wavelength, as long as one of the duty ratios does not exceed 50%, the power source 1 to the DC motor 15, It can be avoided that the timing of current inflow to 16 overlaps.

In this embodiment, the first and second PWM control signals S1 and S2 are generated based on the sawtooth carriers C and D having a phase difference of ½ wavelength, and the first and second PWM control signals are generated. By operating the DC motors 15 and 16 by S1 and S2, the current inflow timing from the power source 1 to the DC motors 15 and 16 does not overlap unless the duty ratio of either one exceeds 50%. Even when both duty ratios change every moment, the current supplied from the power source 1 can be smoothed and the frequency at which a large current is supplied from the power source 1 can be greatly reduced.
Further, the frequency of supplying a large current from the power source 1 is reduced, which is effective in designing a harness capacity, a connector contact capacity, a noise filter, a shunt resistance rating, and the like.

In the present embodiment, the sawtooth wave carrier C is used as a carrier for generating the PWM control signal, but a triangular wave carrier E can be used instead as shown in FIG.
In this case, the first PWM control signal generation unit 36 generates the first PWM control signal S3 from the triangular wave carrier E generated by the carrier generation unit 40 and the duty ratio A, and the triangular wave carrier E is The second PWM control signal generation unit 37 generates the second PWM control signal S4 from the triangular wave carrier wave F generated by shifting two wavelengths and the duty ratio B.
Also in this case, as in the embodiment, unless one of the duty ratios exceeds 50%, the current inflow timing from the power source 1 to the DC motors 15 and 16 does not overlap. Even when the voltage changes from moment to moment, the current supplied from the power source 1 can be smoothed and the frequency at which a large current is supplied from the power source 1 can be greatly reduced.

Next, a second embodiment will be described.
In this embodiment, a microcomputer 19A having a different internal configuration is used instead of the microcomputer 19 in the first embodiment, and the overall configuration is the same as that shown in FIG.
Next, an internal process of the microcomputer 19A that generates the PWM control signal will be described.
FIG. 5 is a diagram showing an internal configuration of the microcomputer, and FIG. 6 shows a PWM control signal obtained from the carrier wave and the duty ratio.
The microcomputer 19A uses a carrier wave processing unit 41A in place of the phase adjuster 41 inside the microcomputer 19 in the first embodiment.
The configuration other than the carrier wave processing unit 41A is the same as the configuration shown in FIG. 2 of the first embodiment, and the same numbers are assigned and the description is omitted.

The sawtooth carrier C is generated by the carrier generator 40, and the sawtooth carrier C is output to the first PWM control signal generator 36 and the carrier processing unit 41A.
The carrier wave processing unit 41A generates a sawtooth carrier wave G that repeats monotonic decrease from the sawtooth carrier wave C that repeats monotonic increase generated by the carrier wave generation unit 40.
Specifically, by the carrier wave processing unit 41A,
Sawtooth carrier G = 1−sawtooth carrier C
The operation is performed.
The second PWM control signal generation unit 37 generates a second PWM control signal S6 from the sawtooth carrier wave G and the duty ratio B, and outputs the second PWM control signal S6 to the second switching element drive circuit 18, and also the first PWM control signal generation unit 36. Generates a first PWM control signal S5 from the sawtooth carrier C and the duty ratio A, and outputs the first PWM control signal S5 to the first switching element drive circuit 17.

In this embodiment, the first and second PWM control signals S5 and S6 are generated based on the sawtooth carrier C that repeats monotonic increase and the sawtooth carrier G that repeats monotonic decrease. As long as the sum of the duty ratios does not exceed 100%, it is possible to avoid the timing of current inflow from the power source 1 to the DC motors 15 and 16 from overlapping.
Thereby, the frequency at which a large current is supplied from the power supply 1 can be further greatly reduced.

Next, a third embodiment will be described.
In this embodiment, a microcomputer 19B having a different internal configuration is used in place of the microcomputer 19 in the first embodiment. The overall configuration is the same as that shown in FIG.
Next, an internal process of the microcomputer 19B that generates the PWM control signal will be described.
FIG. 7 is a diagram showing an internal configuration of the microcomputer, and FIG. 8 shows a PWM control signal obtained from the carrier wave and the duty ratio.
The microcomputer 19B in this embodiment is obtained by deleting the phase adjuster 41 in the microcomputer 19 in the first embodiment and adding a duty ratio processing section 42 and a signal inverting section 43.
The other configuration is the same as the configuration shown in FIG. 2 of the first embodiment, and the same numbers are assigned and the description is omitted.

The sawtooth carrier C is generated by the carrier generator 40, and the sawtooth carrier C is output to the first PWM control signal generator 36 and the second PWM control signal generator 37B.
The first PWM control signal generation unit 36 generates a first PWM control signal S7 from the duty ratio A and the sawtooth carrier wave C.
The duty ratio B generated by the current control unit 35 is input to the duty ratio processing unit 42, and the duty ratio processing unit 42 generates the duty ratio B ′ from the duty ratio B.
Specifically, in order to take a complement of 1 (100%) for the duty ratio B, in the duty ratio processing unit 42,
Duty ratio B ′ = 1−Duty ratio B
The operation is performed.
The duty ratio B ′ generated by the duty ratio processing unit 42 is input to the second PWM control signal generation unit 37B.

The second PWM control signal generation unit 37 </ b> B includes a comparator 39 and a signal inversion unit 43.
The comparator 39 compares the sawtooth carrier wave C with the duty ratio B ′ (the comparison method is the same as that of the first embodiment), and the result is input to the signal inverter 43 as the comparator output shown in FIG.
The signal inverting unit 43 performs level conversion (high level to low level, or low level to high level conversion) of the input comparator output, and generates the second PWM control signal S8.
The second PWM control signal S8 is output from the microcomputer 19 to the second switching element drive circuit 18.

The present embodiment is configured as described above. Among the duty ratios A and B, the duty ratio B ′ which is a complement of 1 (100%) with respect to one duty ratio B is calculated, and the comparator 39 calculates the duty ratio. Since the comparator output is obtained by comparing B ′ with the sawtooth carrier C, the level of the comparator output is converted by the signal inverting unit 43 to generate the second PWM control signal S8. As long as the current does not exceed 100%, it is possible to avoid the overlapping of the current inflow timing from the power source 1 to the DC motors 15 and 16.
Thereby, the frequency at which a large current is supplied from the power supply 1 can be further greatly reduced.

Next, a fourth embodiment will be described.
FIG. 9 shows the overall configuration. The DC motors 15 and 16 are driven by a control unit 2C having a microcomputer 19C instead of the microcomputer 19 in the first embodiment.
The voltage supplied from the power source 1 to the control unit 2C is applied to the H bridge circuits 5 and 6 via the filter 3 and the input current sensor 44. The sensor output of the input current sensor 44 is input to the microcomputer 19C.
As in the first embodiment, the H bridge circuit 5 and the H bridge circuit 6 are composed of field effect transistors (FETs) 7 to 10 and field effect transistors (FETs) 11 to 14, respectively.

The first switching element drive circuit 17 that drives the H-bridge circuit 5 applies a drive voltage to the gate terminals of the field effect transistors 7 to 10 based on a command from the microcomputer 19C. Similarly, the second switching element drive circuit 18 that drives the H-bridge circuit 6 applies a drive voltage to the gate terminals of the field effect transistors 11 to 14 based on a command from the microcomputer 19C. The DC motors 15 and 16 are supplied with a current for rotating the motor forward or backward by a combination of switching of each field effect transistor.
The other overall configuration is the same as the configuration shown in FIG.

FIG. 10 is a diagram showing an internal configuration of the microcomputer 19C.
In the configuration of the microcomputer 19 in the first embodiment, the microcomputer 19C has a carrier processing unit 41A and a phase adjuster 41C instead of the phase adjuster 41 between the carrier generation unit 40 and the second PWM control signal generation unit 37. Is provided. The other configuration is the same as the configuration shown in FIG. 2 of the first embodiment, and the same numbers are given and the description is omitted.
The above configuration corresponds to a microcomputer 19A of the second embodiment with a phase adjuster 41C added.

As shown in FIG. 3, the carrier wave generation unit 40 generates a sawtooth carrier wave C having a frequency of 10 kHz and an amplitude of 1, for example, and increases to a first PWM control signal generation unit 36 and a carrier wave processing unit 41A.
The first PWM control signal generation unit 36 compares the value of the duty ratio A input from the current control unit 34 and the sawtooth carrier wave C input from the carrier wave generation unit 40 by a comparator 38 provided therein, When the duty ratio A is greater than or equal to the sawtooth wave carrier C, a comparator output that is high level and when the duty ratio A is less than the sawtooth carrier wave C is output as the first PWM control signal S9.
The carrier wave processing unit 41A is based on the sawtooth carrier wave C that repeats the monotonic increase generated by the carrier wave generation unit 40.
Sawtooth carrier G = 1−sawtooth carrier C
The sawtooth carrier wave G that repeats monotonic decrease is generated.

The phase adjuster 41C offsets the phase of the monotonously decreasing sawtooth wave carrier G output from the carrier wave processing unit 41A by a predetermined time, and outputs the offset to the second PWM control signal generation unit 37 as a sawtooth carrier wave H. The phase adjuster 41 is different from the phase adjuster 41 in the first embodiment in that the wavelength is shifted by ½ wavelength. For example, the phase adjuster 41 shifts the phase in units of several% within 10% of the carrier wave period.
The second PWM control signal generation unit 37 compares the duty ratio B input from the current control unit 35 with the sawtooth carrier wave H and outputs a second PWM control signal S10.
The sensor output (that is, the current value) detected by the input current sensor 44 corresponds to the first and second PWM control signals S5 and S7, and thus indicates the current flowing into the DC motors 15 and 16.

Next, the offset procedure of the phase of the sawtooth carrier wave G by the phase adjuster 41C will be described.
First, the first and second PWM control signals S9 and S10 for driving the DC motors 15 and 16 are output at a duty of 50%, respectively, as shown in (a) and (b) of FIG. At this time, as shown in FIG. 11C, a peak like P1 appears at the switching point of the first and second PWM control signals in the input current of the control unit 2C (sensor output of the input current sensor 44). .
This is due to the following reason.
That is, a switching element such as a field effect transistor has a response delay, and even if a pulsed drive signal is input, as shown in FIGS. 12A and 12B, the output current ( (Current to the DC motor) does not rise immediately, and the output current does not stop immediately even when the drive signal is turned off. As a result, the current waveform by the PWM control is almost first-order lag, and the currents flowing into the DC motors 15 and 16 overlap as shown in FIG. 12C, forming a peak P in the input current of the control unit 2C. It depends. In general, output rise and fall characteristics are also different.

Therefore, the phase adjuster 41C sequentially decreases the duty ratio of the second PWM control signal S10 and increases / decreases the phase of the sawtooth carrier wave, searches for a point where no peak of the input current occurs, and offset value of the sawtooth carrier wave H. And
For example, first, as shown in FIG. 13B, the duty ratio of the second PWM control signal S10 for driving the DC motor 16 is reduced to 45%, and the state of the input current of the control unit 2C is checked. Here, the first PWM control signal S9 is held at Duty 50% as shown in FIG.

Here, as shown in FIG. 13 (c), when the peak of the input current remains P1, the sawtooth wave of the sawtooth carrier wave H is shown in FIGS. 14 (a) and 14 (b). The phase for carrier C is advanced to 2.5% of the carrier period and offset. Thereby, as shown in (c) of Drawing 14, the peak of the input current of control unit 2C decreases like P2.
Subsequently, as shown in FIGS. 15A and 15B, the phase of the sawtooth carrier H with respect to the sawtooth carrier C is advanced to 5% of the carrier cycle. Here, when the peak of the input current increases again as shown in P3 as shown in FIG. 15C, the phase difference of 2.5% at which the peak decreased immediately before is set to an appropriate value and 2.2. The sawtooth carrier wave H is offset by a time corresponding to 5%.

  The setting of the phase offset of the sawtooth carrier wave can be performed, for example, when the control unit is activated. That is, when the control unit is activated, for example, when the initial position learning of the motor rotation angle sensors 22 and 23 for inputting the motor rotation angle signal to the positioning servo units 32 and 33 is performed, for example, the actuator connected to the motor is temporarily moved within the operating range. Abut against the limit stopper. Therefore, at this time, an arbitrary PWM control signal based on the sawtooth carrier wave whose phase is shifted can be output without affecting the other.

In the present embodiment, the first and second PWM control signals are generated on the basis of the sawtooth carrier C that repeats monotonic increase and the sawtooth carrier G that repeats monotonic decrease. Similarly, it is possible to avoid the overlapping of the current inflow timing from the power source 1 to the DC motors 15 and 16, and to reduce the frequency of supplying a large current from the power source 1.
Since the sawtooth carrier C and the sawtooth carrier H to be compared with the duty ratio B by the second PWM control signal generation unit 37 have an offset corresponding to a predetermined time, that is, a phase difference, the H bridge circuit Even when there is a response delay in the operation of the field effect transistors forming the, the control is performed at a timing at which no overlap occurs due to the response delay between the field effect transistors, and the upper limit of the input current can be reliably suppressed low. This is a great advantage especially when using a general mass-produced field effect transistor in which various characteristics including rise and fall vary from product to product.

As for the offset between the sawtooth carrier C and the sawtooth carrier H, the DC motors 15 and 16 are started to drive with the first and second PWM control signals set to predetermined fixed duty ratios, and the input current of the control unit 2C Is adjusted via the input current sensor 44, the duty ratio and the phase difference are adjusted, and the phase difference at which the peak of the input current is reduced is selected. Therefore, the setting is easy.
In the present embodiment, in setting the offset, the duty ratio of the second PWM control signal S10 is first reduced to 45% and the phase is advanced in units of 2.5% of the carrier wave period. The increase / decrease width of the phase can be arbitrarily set according to the period of the carrier wave, the characteristics of the field effect transistor used in the H-bridge circuit, and the like.

Next, a fifth embodiment will be described.
This is provided with a microcomputer 19D instead of the microcomputer 19C in the fourth embodiment, and the other overall configuration is the same as the configuration shown in FIG.
FIG. 16 is a diagram showing an internal configuration of the microcomputer 19D.
The microcomputer 19D uses the second PWM control signal generation unit 37B including the signal inversion unit 43 instead of the second PWM control signal generation unit 37 in the configuration of the microcomputer 19 in the first embodiment, and also uses the current control unit 35. And a duty ratio processing section 42 between the second PWM control signal generation section 37B. A phase adjuster 41C is provided between the carrier wave generation unit 40 and the second PWM control signal generation unit 37B. The other configuration is the same as the configuration shown in FIG. 2 of the first embodiment, and the same numbers are given and the description is omitted.
The above configuration corresponds to the microcomputer 19B of the third embodiment with a phase adjuster 41C added.

  The first PWM control signal generator 36 receives the sawtooth carrier C from the carrier generator 40 and the duty ratio A (Duty ratio A = 0 to 1) from the current controller 34. The first PWM control signal generator 36 compares the duty ratio A and the sawtooth carrier C with the internal comparator 38, and when the duty ratio A ≧ the sawtooth carrier C, the high level, the duty ratio A <sawtooth carrier. In the case of C, a first PWM control signal S11 that is at a low level is generated.

The phase adjuster 41C generates a sawtooth carrier wave J having a phase obtained by offsetting the sawtooth carrier C input from the carrier generator 40 by a predetermined time, and outputs the sawtooth carrier J to the second PWM control signal generator 37B.
In the duty ratio processing unit 42, the duty ratio B ′ is generated from the duty ratio B (duty ratio B = 0 to 1) generated by the current control unit 35 by the calculation of the following equation.
Duty ratio B ′ = 1−Duty ratio B
The duty ratio B ′ generated by the duty ratio processing unit 42 is input to the second PWM control signal generation unit 37B.
In the second PWM control signal generation unit 37B, the duty ratio B ′ and the sawtooth carrier wave J are compared by the internal comparator 39, and when the duty ratio B ′ ≧ the sawtooth carrier wave J, the high level and the duty ratio B ′ < In the case of the sawtooth wave carrier wave J, a low level signal is generated, and this signal is inverted by the signal inverter 43 and output as the second PWM control signal S12.

The present embodiment is configured as described above, and similarly to the third embodiment, the duty ratio B ′ that is a complement of 1 (100%) with respect to one duty ratio B out of the duty ratios A and B is set as follows. After calculating and comparing the duty ratio B ′ and the sawtooth carrier wave J by the comparator 39 to obtain the comparator output, the signal inversion unit 43 performs level conversion of the comparator output to generate the second PWM control signal S12. Therefore, as long as the sum of the duty ratios of both does not exceed 100%, it is possible in principle to avoid overlapping the timing of current inflow from the power source 1 to the DC motors 15 and 16.
Furthermore, since the sawtooth carrier C and the sawtooth carrier J are offset by a predetermined time, even if there is a response delay in the operation of the field effect transistor forming the H bridge circuit, the field effect transistor The upper limit of the input current can be surely kept low by controlling at the timing at which no overlap occurs due to the response delay.

It is a figure which shows the whole structure of a 1st Example. It is a figure which shows the internal structure of the microcomputer in a 1st Example. It is a figure which shows the PWM control signal obtained from the carrier wave and Duty ratio in a 1st Example. It is a figure which shows the PWM control signal obtained from the carrier wave and Duty ratio in a modification. It is a figure which shows the internal structure of the microcomputer in a 2nd Example. It is a figure which shows the PWM control signal obtained from the carrier wave and Duty ratio in a 2nd Example. It is a figure which shows the internal structure of the microcomputer in a 3rd Example. It is a figure which shows the PWM control signal obtained from the carrier wave and Duty ratio in a 3rd Example. It is a figure which shows the whole structure of a 4th Example. It is a figure which shows the internal structure of the microcomputer in a 4th Example. It is a figure which shows the point of the phase offset of a carrier wave. It is explanatory drawing which shows the response delay of a field effect transistor. It is a figure which shows the point of the phase offset of a carrier wave. It is a figure which shows the point of the phase offset of a carrier wave. It is a figure which shows the point of the phase offset of a carrier wave. It is a figure which shows the internal structure of the microcomputer in a 5th Example.

Explanation of symbols

1 Power supply 2, 2C Control unit 5, 6 H bridge circuit 7-14 Field effect transistor 15, 16 DC motor (inductive load)
17 First switching element driving circuit 18 Second switching element driving circuit 19, 19A, 19B, 19C, 19D Microcomputer 26, 27 Communication unit 28-31 Analog signal input unit 32, 33 Positioning servo unit 34, 35 Current control unit 36 First PWM control signal generation unit (first pulse width modulation control signal generation unit)
37, 37B Second PWM control signal generator (second pulse width modulation control signal generator)
38, 39 Comparator 40 Carrier wave generation unit 41 Phase adjuster 41A Carrier wave processing unit 41C Phase adjuster (carrier wave phase adjusting means)
42 Duty ratio processing unit 43 Signal inversion unit 44 Input current sensor

Claims (8)

A first pulse width modulation control signal generating section and a second pulse width modulation control signal generating section for generating a modulation control signal for independently driving two inductive loads connected to a power source by an H bridge circuit using a pulse width modulation control method; In an inductive load driving device including a pulse width modulation control signal generation unit,
The first pulse width modulation control signal generation unit and the second pulse width modulation control signal generation unit generate the modulation control signal by comparing the duty ratio in the pulse width modulation control and the sawtooth carrier wave, respectively.
The sawtooth carriers used by the first pulse width modulation control signal generator and the second pulse width modulation control signal generator have the same frequency, and the phase of one sawtooth carrier and the other sawtooth carrier is 1/2. An inductive load driving device characterized by being shifted in wavelength. A first pulse width modulation control signal generating section and a second pulse width modulation control signal generating section for generating a modulation control signal for independently driving two inductive loads connected to a power source by an H bridge circuit using a pulse width modulation control method; In an inductive load driving device including a pulse width modulation control signal generation unit,
The first pulse width modulation control signal generation unit and the second pulse width modulation control signal generation unit generate the modulation control signal by comparing the duty ratio in the pulse width modulation control and the triangular wave carrier wave, respectively.
The triangular wave carriers used by the first pulse width modulation control signal generation unit and the second pulse width modulation control signal generation unit have the same frequency, and the phases of one triangular wave carrier and the other triangular wave carrier are shifted by ½ wavelength. An inductive load driving device characterized by comprising: A first pulse width modulation control signal generating section and a second pulse width modulation control signal generating section for generating a modulation control signal for independently driving two inductive loads connected to a power source by an H bridge circuit using a pulse width modulation control method; In an inductive load driving device including a pulse width modulation control signal generation unit,
The first pulse width modulation control signal generation unit and the second pulse width modulation control signal generation unit generate the modulation control signal by comparing the duty ratio in the pulse width modulation control and the sawtooth carrier wave, respectively.
The sawtooth wave carriers used by the first pulse width modulation control signal generation unit and the second pulse width modulation control signal generation unit have the same frequency, and one sawtooth carrier wave is a monotonically increasing repetition, and the other sawtooth wave. An inductive load driving device characterized in that the carrier wave is a monotonically decreasing repetition. A first pulse width modulation control signal generating section and a second pulse width modulation control signal generating section for generating a modulation control signal for independently driving two inductive loads connected to a power source by an H bridge circuit using a pulse width modulation control method; In an inductive load driving device including a pulse width modulation control signal generation unit,
A duty ratio processing unit for calculating a complement of 1 with respect to the duty ratio in the pulse width modulation control;
The second pulse width modulation control signal generation unit includes a signal inversion unit that inverts a signal level from a high level to a low level or from a low level to a high level,
The first pulse width modulation control signal generation unit and the second pulse width modulation control signal generation unit use sawtooth carrier waves having the same frequency and the same phase,
The first pulse width modulation control signal generation unit generates the modulation control signal by comparing a duty ratio in the pulse width modulation control with the sawtooth carrier wave,
The second pulse width modulation control signal generation unit performs signal level inversion on the signal obtained by comparing the value calculated by the duty ratio processing unit with the sawtooth carrier wave by the signal inversion unit, An inductive load driving device characterized in that the level-inverted signal is generated as the modulation control signal. A first pulse width modulation control signal generating section and a second pulse width modulation control signal generating section for generating a modulation control signal for independently driving two inductive loads connected to a power source by an H bridge circuit using a pulse width modulation control method; In an inductive load driving device including a pulse width modulation control signal generation unit,
Carrier phase adjustment means for offsetting the sawtooth carrier used by the second pulse width modulation control signal generator by a phase corresponding to a predetermined time with respect to the sawtooth carrier used by the first pulse width modulation control signal generator;
The sawtooth wave carriers used by the first pulse width modulation control signal generation unit and the second pulse width modulation control signal generation unit have the same frequency, and one sawtooth carrier wave is a monotonically increasing repetition, and the other sawtooth wave. The carrier is a monotonically decreasing iteration,
The first pulse width modulation control signal generation unit generates the modulation control signal by comparing the duty ratio in the pulse width modulation control and the sawtooth carrier wave,
The second pulse width modulation control signal generation unit generates the modulation control signal by comparing a duty ratio in pulse width modulation control with a sawtooth carrier wave whose phase is offset. Drive device. A first pulse width modulation control signal generating section and a second pulse width modulation control signal generating section for generating a modulation control signal for independently driving two inductive loads connected to a power source by an H bridge circuit using a pulse width modulation control method; In an inductive load driving device including a pulse width modulation control signal generation unit,
A duty ratio processing unit for calculating a complement of 1 with respect to the duty ratio in the pulse width modulation control;
Carrier wave phase adjusting means for offsetting the sawtooth carrier wave used by the second pulse width modulation control signal generator by a phase corresponding to a predetermined time with respect to the sawtooth carrier wave used by the first pulse width modulation control signal generator. ,
The second pulse width modulation control signal generation unit includes a signal inversion unit that inverts a signal level from a high level to a low level or from a low level to a high level,
The sawtooth carrier waves used by the first pulse width modulation control signal generation unit and the second pulse width modulation control signal generation unit have the same frequency,
The first pulse width modulation control signal generation unit generates the modulation control signal by comparing the duty ratio in the pulse width modulation control and the sawtooth carrier wave,
The second pulse width modulation control signal generator generates a signal obtained by comparing the value calculated by the duty ratio processing unit with the sawtooth carrier wave having the phase offset by the signal inversion unit. An inductive load driving device characterized by performing level inversion and generating the level-inverted signal as the modulation control signal. The carrier wave phase adjusting means adjusts one duty ratio and the phase difference while monitoring the current value flowing into the inductive load from a state in which the inductive loads are each driven at a predetermined fixed duty ratio. An inductive load driving device characterized in that a phase difference for reducing a peak of the current value is obtained and the phase difference is used as an offset amount of the phase. The first pulse width modulation control signal generation unit and the second pulse width modulation control signal generation unit use the sawtooth carrier wave having the same frequency and the phase shifted by ½ wavelength, and the sawtooth carrier wave and the pulse width modulation control. Each of the modulation control signals is generated by comparing the duty ratio in the control circuit, and the two inductive loads connected to the power source are independently converted into a pulse width modulation control system by the H bridge circuit using the modulation control signal. An inductive load driving method characterized by driving using a power source.

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