JPS5911256B2 - Switching regulator - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a switching regulator used in a power source circuit such as a color television receiver.

For regulators used in constant voltage power supply circuits, switching type regulators are more efficient than continuous control type ones, and the losses of the semiconductor elements used are 35 less, making them suitable for devices with large power consumption such as color television receivers. It is suitable for power supply circuits and has become widely adopted in recent years.

An example of such a switching regulator is shown in FIG.

An alternating current voltage from an AC plug 1 connected to a commercial power source is rectified by bridge-connected diodes 2 to 5 to obtain a direct current between the terminals of a smoothing capacitor 7 to supply an input voltage Ei to the regulator.

At this time, the surge current is limited by the resistor 6. The input voltage Ei is applied to a chiyoke input type smoothing circuit consisting of a chiyoke coil 10 and a capacitor 11 via an output transistor 8 for switching, and is smoothed together with a commutating diode 9, resulting in an output that is regulated to a predetermined voltage. A voltage EO is supplied to the output terminal. An excitation pulse transformer 14 is connected between the base and emitter of the output transistor 8, and the switching state is controlled via the excitation transistor 16 according to fluctuations in the output voltage EO.

A capacitor 25 is connected to the base of the excitation transistor 16.
The flyback pulse FH is applied from
The error voltage signal is compared with the constant voltage of the Zener diode 26 by the transistor 17 for error amplification, and is detected at the collector of the transistor 17 as a variation of the voltage taken out by the resistors 12 and 13 in proportion to the output voltage EO. are added in a superimposed manner by resistors 20, 21, and 22.

Note that 18 is the collector resistance of the excitation transistor 16;
19 is the operating resistance of the Zener diode 26.

Further, a suitable power supply 1 is connected in series with the secondary winding of the excitation transformer 14 between the base and emitter of the output transistor 8.
5 is connected to apply a bias in the conduction direction to the output transistor 8.

Then, by setting the Zener voltage of the Zener diode 26 and the voltage taken out from the output voltage EO by the resistors 12 and 13 and applied to the base of the transistor 17 to appropriate values, the conduction period of the output transistor 8 is cut off. The output transistor 8 is switched to a point where the conduction period is almost equal to the switching period, so that stable constant voltage operation can be achieved even with an input voltage Ei slightly higher than the output voltage EO. I'm getting better at it.

Therefore, even when used as a power supply circuit for equipment that requires a voltage of 100 to 120V as a +B power source, such as a color television receiver, a power transformer or voltage doubler rectifier circuit must be used. Directly rectifies the 100V commercial AC power supply without boosting the input voltage E.
It can be used as an i, and can be configured extremely advantageously in terms of cost and equipment safety.

This operation will be explained using the waveform diagram in FIG.

The flyback pulse FH shown by waveform 27 is connected to resistor 23.
is integrated by the capacitor 24 to form a sawtooth wave as shown in the waveform 28, which is superimposed on the word difference voltage generated at the collector of the error amplification transistor 17 and applied to the base of the excitation transistor 16. As a result, the output voltage EO becomes When this changes, the average level of the sawtooth waveform 28 will rise or fall. If the conduction level of the transistor 16 is represented by a straight line 29, the transistor 16 is non-conductive only during the period when the waveform 28 is below the straight line 29. As a result, the collector waveform of the transistor 16 becomes as shown in 31, but a pulse with the opposite polarity of the waveform 31 is applied to the base of the output transistor 8 via the pulse transformer 14, and the transistor 8
is switched to the cut-off state by this pulse. As a result, the period TON during which the transistor 8 is in a conductive state
The output voltage E is the ratio between TOFF and the cutoff period TOFF.

When the output voltage EO increases beyond a predetermined value, the voltage applied to the base of the transistor 17 also increases, and its collector current decreases. As a result, waveform 2
8 decreases and the width of the portion below the straight line 29 becomes wider, the width of the pulse waveform 31 applied to the output transistor 8 also becomes wider, and its cutoff period T
Since the OFF period becomes longer and the conduction period TON becomes relatively shorter, the average value of the voltage supplied to the output side through the transistor 8 decreases, resulting in the output voltage E. The increased amount is canceled out, and control is applied to return the voltage to its original level. When the output voltage EO decreases below a predetermined value, the average value of the waveform 28 increases, the cutoff period TOFF becomes shorter, and the output voltage EO is controlled to increase, resulting in constant voltage operation.

As already explained, according to this switching regulator, since the bias from the appropriate bias source 15 is superimposed on the switching pulse from the pulse transformer 14, the switching of the output transistor 8 is controlled during the conduction period TON. The input voltage E
It has the advantage that the required output voltage EO can be obtained without making i too high, and is suitable for power supply circuits such as color television receivers. However, as is clear from FIG. 2, the input voltage Ei and the output voltage E.

When operating in a region where the difference between the output transistor 8 and The value of the portion 30 below the voltage level is reduced, and the excitation transistor 16
cannot be kept sufficiently shut off. Therefore, the pulse in the cutoff direction supplied to the output transistor 8 also has a waveform of 32.
This results in an uneven waveform as shown in FIG. 2, and it becomes impossible to stably switch the transistor 8. In such a state, abnormal oscillation may occur in the circuit or the output voltage E. As a result, the upper limit of the output voltage EO that can be stably obtained for a constant input voltage Ei will decrease, and the input voltage E
The disadvantage is that it is difficult to obtain a large output voltage EO with a value extremely close to i. Also, the portion 3 below the straight line 29 of the waveform 28
A smaller value of 0 means that the risk of malfunction due to noise or the like increases, and the operating range becomes even narrower. As already mentioned, color television receivers often require a B power source with a voltage of 100 to 120 V, and therefore, as in Japan and the United States, a voltage of 100 V to 120 V is required as an AC power source.
In areas where 120 power supplies are used, the input voltage El and the output voltage EO will be close values (
Normally, the AC input voltage is 9.
(The design standard is the value of the input voltage Ei obtained when the voltage drops to about 0%.) Therefore, with a regulator like the one shown in Figure 1, it may be difficult to obtain the required output voltage EO. Ta.

An object of the present invention is to eliminate the above-mentioned drawbacks of the switching regulator, to perform stable operation even when the input voltage Ei and the output voltage EO are close to each other, and to make the output voltage EO extremely low with respect to the input voltage Ei. An object of the present invention is to provide a switching regulator that can operate stably even in a wide range of conditions.

To achieve this object, the present invention is characterized in that another pulse unrelated to the average level change is superimposed on the sawtooth wave signal for controlling the switching pulse width. The present invention will now be described with reference to FIGS. 3 to 8.

FIG. 3 shows an embodiment of the present invention, which basically has almost the same configuration and operation as the circuit shown in FIG. 1, except that the signal applied to the base of the excitation transistor 16 is , a resistor 23, and a capacitor 24, and an error voltage signal from the transistor 17, but also a pulse from a pulse generating circuit 33 is superimposed on these signals. It's only on.

Next, the operation will be explained using the waveform diagram in Fig. 4.
A sawtooth waveform 28 is formed by the resistor 23 and the capacitor 24 from the flyback pulse shown by the waveform 27, and the average level thereof changes depending on the error voltage signal from the transistor 17, resulting in a straight line 29 indicating the conduction level of the excitation transistor 16.
The width of the pulse waveform 31 that controls the cutoff of the output transistor 8 by changing its position vertically relative to the output voltage E is the width of the pulse waveform 31 that controls the cutoff of the output transistor 8.

The point that the output voltage EO is stabilized by changing it according to fluctuations in the output voltage is the same as in the case of FIG. The difference is that a pulse 34 is added.
That is, the pulse generating circuit 33 is a circuit having a function of generating a pulse with an extremely small duty ratio that is synchronized with the sawtooth waveform 28 and phase-locked with the vicinity of the lowest level of the waveform.

Now, as this pulse 34 is applied to the base of the excitation transistor 16, the average level of the sawtooth waveform 28 becomes high, that is, the output voltage E.

If the control state is set so that Ei operates in a region very close to the input voltage Ei, it is natural that the waveform 28 will become a straight line 29.
The part that goes further downward becomes so small that it becomes impossible to sufficiently shut off the transistor 16, but there is a pulse 34 in this part, and its voltage value has a sufficient amplitude in the negative direction. Therefore, the transistor 16 is sufficiently driven in the cutoff direction by this pulse 34, and the cutoff period TOFF is extremely short. Furthermore, a pulse 31 with a correct waveform can be supplied to the output transistor 8, and stable operation can be achieved even when the output voltage EO is increased to a voltage extremely close to the input voltage Ei. Therefore, according to this embodiment, the first
Compared to the switching regulator shown in the figure, the cut-off period TOFF of the output transistor 8 can be made shorter than the conduction period TON, and the output voltage EO, which is extremely close to the input voltage Ei, can be stably supplied to the load. It will be possible.

At this time, the width of the pulse 34 determines how close the output voltage EO can be to the input voltage Ei, so the width of the pulse 34 may be appropriately determined as necessary.

In reality, due to the carrier accumulation effect of the output transistor 8, the conduction period of the output transistor 8 becomes longer than the conduction period of the excitation transistor 16. In other words, the cut-off period of the output transistor 8 is shorter than the cut-off period of the excitation transistor 16 by the carrier accumulation time.
If the width of the pulse 34 supplied from the pulse generating circuit 33 is made to be approximately the same as the carrier accumulation time of the output transistor 8, the conduction period T will cover almost the entire switching period. N, and the input voltage Ei
It is possible to stably supply an output voltage EO approximately equal to . FIG. 5 shows an example of a pulse width control signal generating circuit used in the present invention, and the entire circuit is shown as a block diagram. In this example, a flyback pulse FH is applied to the integrating circuit 36 (which corresponds to the resistor 23 and capacitor 24 in the embodiment of FIG. 3) and the differentiating circuit 37, and the output of the integrating circuit 36 is transferred to the phase inverting circuit 37.
8 and is combined with the output of the differentiating circuit 37 to obtain a signal to be applied to the base of the excitation transistor 16.

FIG. 7 is a waveform diagram thereof, in which a waveform 27 indicating the flyback pulse FH becomes a sawtooth waveform 28 in an integrating circuit 36, and is inverted into a waveform 39 in a phase inverting circuit 38.

Further, the pulse waveform 27 applied to the differentiating circuit 37 is converted into a differentiated pulse waveform 40, which is synthesized to form a waveform 41.
will be formed. The signal shown by this waveform 41 is
Similar to the signal obtained by combining the waveforms 28 and 34 in FIG. 4, the output transistor 8 may be switched in addition to the excitation transistor 16 together with the error voltage signal from the transistor 17. This waveform 4
Since the negative part of the part consisting of the pulse waveform 40 of 1 has a value 43 that is sufficiently lower than the straight line 42 indicating the conduction level of the excitation transistor 16, this circuit can be constructed from the resistor 23 and capacitor 24 shown in FIG. A similar effect can be obtained by configuring a switching regulator in place of the integrating circuit and pulse generating circuit 33. At this time, a part of the differential pulse waveform 40 is added above the synthesized waveform 41, but since this part becomes the conduction region of the excitation transistor 16, it is not affected in any way. Further, although the operating pulses are obtained from flyback pulses, they may also be obtained from pulses from other parts of the horizontal deflection circuit, such as an excitation transformer, or a completely independent oscillator may be used. good.

FIG. 6 shows another example of the pulse width control signal generation circuit used in the present invention. What is different from the case of FIG. 5 is that a phase inversion circuit 38 is provided after the differentiating circuit 37, The phase of the pulse is inverted and superimposed on the sawtooth wave from the integrating circuit 36, and its operation is explained in the fifth section.
This is the same as the case shown in the figure. FIG. 8 shows an embodiment of the present invention specifically showing the configuration of a pulse width control signal generating circuit, and is an example of the configuration corresponding to the circuit of FIG. 5.

The integrating circuit 36 in FIG. 5 consists of a resistor 23 and a capacitor 2.
4, the phase inversion circuit 38 is made up of a transistor 44, and the differentiating circuit 38 is made up of a capacitor 45 and a resistor 46.

Note that 48 and 49 are bias resistors for the transistor 44, 50 is a load resistor, 47 is a gain setting resistor for the transistor 44, and 55 is a coupling capacitor. The flyback pulse FH applied via the coupling capacitor 25 is
Waveform 4 in Figure 7 is obtained by differentiating the capacitor 45 and resistor 46.
It is shaped into a differential pulse of 0, integrated by a resistor 23 and a capacitor 24, shaped into a sawtooth waveform 28, inverted by a truss register 47, and synthesized into a waveform 39.
The error voltage signal from the transistor 17 is superimposed as a waveform 41 and applied to the base of the excitation transistor 16.

The stabilizing operation of the output voltage EO due to this is the same as described with reference to FIG. 3, and therefore will not be described here.

In the above embodiment, the output transistor 8 and the choke coil 10, which are switching elements, are shown as being inserted on the negative line side, but it goes without saying that they may be inserted on the positive line side. . Further, although the output transistor 8 is used as a switching element, it may be constructed from a general switching element such as a thyristor or a gate turn-off thyristor.

As explained above, according to the present invention, the switching element can be made conductive over almost the entire switching period by using a simple configuration in which a pulse generation circuit is provided and a pulse is superimposed on a sawtooth wave signal. Since it is possible to stably supply an output voltage EO that is almost close to the input voltage Ei, there is no need to obtain a high input voltage Ei using a voltage doubler rectifier or the like. It is possible to reduce costs and also obtain products with high safety.

Furthermore, even if the switching element is controlled so that its cut-off period covers almost the entire switching period, it operates stably, so the output voltage E is extremely low compared to the input voltage Ei. However, it can be supplied stably, and can be used to great effect in power supply circuits for television receivers that share a common battery. In this case, the positive direction pulse portion of the waveform 41 in FIG. 7 is utilized, and the excitation transistor 16 is sufficiently driven in the conduction direction even during an extremely short conduction period, so that stable operation can be obtained. become.

[Brief explanation of drawings]

FIG. 1 is an electric circuit diagram of a switching regulator that applies bias to a switching element, FIG. 2 is a waveform diagram for explaining its operation, and FIG. 3 is an electric circuit diagram showing a basic embodiment of the present invention. Figure 4 is a waveform diagram 0 for explaining its operation.
5 and 6 are block diagrams showing examples of pulse width control signal generation circuits used in the switching regulator of the present invention, FIG. 7 is a waveform diagram for explaining its operation, and FIG. 8 is a specific example of the present invention. FIG. 2 is an electrical circuit diagram showing a typical embodiment. 8...Output transistor for switching, 9.
... Commutation diode, 10 ... Smoothing coil, 14 ... Pulse transformer for excitation, 15 ... Bias power supply, 16 ...
...Excitation transistor, 17...Error amplification transistor, 23, 24...Resistor and capacitor for integration, 33...Pulse generation circuit, 36.
... Integrating circuit, 37... Differentiating circuit, 38
... Phase inversion circuit, 44 ... Phase inversion transistor, 45, 46 ... Capacitor and resistor of differentiating circuit.

Claims (1)

[Scope of Claims] 1. An unstable DC power source, a switching element that intermittents the output DC voltage of the DC power source, a DC converter that converts the intermittent output DC voltage into a DC voltage, and a sawtooth-shaped device with a predetermined period. a signal source that generates a wave signal, and a first superimposing means that is connected to the signal source and the output of the DC converting means and that superimposes a sawtooth wave signal on the DC component that fluctuates in accordance with fluctuations in the output voltage of the DC converting means. and a control means for controlling the ratio of conduction and non-conduction periods of the switching element so that the fluctuation of the DC component is suppressed depending on whether the output of the superimposition means exceeds a predetermined level;
a second superimposing means for superimposing a pulse with a small duty ratio synchronized with a sawtooth wave signal having a polarity that makes the switching element non-conductive on the cutting edge value of the superimposing means during the non-conductive period of the switching element; A switching regulator characterized in that it is operable by extending a conduction period or a non-conduction period of a switching element to nearly the entire area. 2. The switching regulator according to claim 1, wherein the width of the pulse superimposed on the sawtooth signal is set to a duration approximately equal to the carrier accumulation time of the output transistor, which is a switching element. 3. The switching regulator according to claim 1, wherein a pulse shaped from a flyback pulse is used as the pulse superimposed on the sawtooth signal. 4. The switching regulator according to claim 1, wherein a pulse obtained from a horizontal excitation circuit is used as the pulse superimposed on the sawtooth wave signal. 5 As a signal for controlling the pulse width of the switching element,
The use of a signal obtained by inverting the phase of either the sawtooth wave signal obtained by integrating the pulse from the horizontal deflection circuit or the pulse obtained by differentiating the pulse and then combining the two is considered to be equivalent. The switching system according to claim 1
regulator. 6. The switching regulator according to claim 5, wherein a flyback pulse is used as the pulse from the horizontal deflection circuit. 7. The switching regulator according to claim 5, wherein a pulse from a horizontal excitation circuit is used as the pulse from the horizontal deflection circuit.