JPH0440956B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION [Field of Application of the Invention] The present invention relates to a control device for an induction motor, and particularly to a control device for a vector-controlled induction motor capable of high-response torque control.

[Background of the invention]

As a method of controlling an induction motor with high response, there is vector control in which the primary current is decomposed into a component in the same direction as the magnetic flux axis (excitation current) and a component orthogonal to it (torque current), and each component is controlled separately.

In order to perform vector control with high precision, the applicant has developed a method in which the excitation current and torque current of the primary current are detected as DC amounts, and the inverter is controlled from the deviation between the command value and actual value of each component. 1977-
It is proposed as No. 199489.

However, when high-response torque control is performed using this method, the following problem arises when high-precision torque control is performed particularly in a high-speed region.
When the excitation current fluctuates, the torque voltage component, which is a voltage component in the same direction as the torque current, fluctuates, which causes the torque current to fluctuate. The product of the excitation current and the primary frequency gives a torque voltage component that almost matches the torque voltage component, so the higher the speed, the more the influence of the excitation current on the torque voltage component cannot be ignored. This is because when the torque changes, the torque voltage component changes, and as a result, the torque current also changes. Conversely, when fluctuations occur in the torque current, the excitation current also changes. As a result, high-response torque control becomes impossible.

[Purpose of the invention]

The present invention has been made in response to the above-mentioned problems, and its purpose is to provide an induction motor control device that can perform highly responsive torque control even in high-speed ranges.

[Summary of the invention]

The feature of the present invention is that the primary current of the induction motor is decomposed into an excitation current and a torque current, and a DC signal corresponding to each is obtained, and the electric power is adjusted so that the deviation between the DC signal of both components and the command signal of both components becomes zero. When controlling the converter, a signal corresponding to the product of the excitation current and the primary frequency is added to a torque voltage component command signal that commands the magnitude of the torque voltage component orthogonal to the magnetic flux.

[Embodiments of the invention]

FIG. 1 shows an embodiment of the present invention.

In FIG. 1, an induction motor 3 is driven by a PWM inverter 2 supplied with voltage from an AC power source 1. The PWM inverter 2 has a configuration in which switching elements such as transistors are connected in a grates manner, and a flywheel diode is connected in antiparallel to each switching element. PWM inverter 2
The output current of is detected by a current detector 18. A speed detector 4 is mechanically directly connected to the induction motor 3. The speed command signal N ^* of the speed command circuit 5 and the speed detection signal N of the speed detector 4 are added by an adder 6 with the polarity shown, and the deviation ΔN is calculated by the speed control circuit 7.
is input. The speed control circuit 7 outputs a torque current command signal I _t ^* that commands the magnitude of the torque current of the electric motor 3 in proportion to the speed deviation ΔN. The slip frequency calculation circuit 8 inputs the torque current command signal I _t ^* and outputs a slip frequency command signal ω _s ^* according to the signal I _t ^* . The slip frequency command signal ω _s ^* is sent to the adder 9
is inputted into the adder 9 and added to the speed detection signal N, and a primary frequency command signal ω ₁ ^* is output. The primary frequency command signal ω ₁ ^* of the adder 9 is input to the oscillator 17 . The oscillator 17 generates a two-phase sine wave signal sinω ₁ ^* t corresponding to the primary frequency command signal ω ₁ ^* ,
Output cosω ₁ ^* t. The two-phase sine wave signal is input to a current component detection circuit 19 and a Bertol calculation circuit 20. The current detection signal i detected by the current detector 18 is input to the current component detection circuit 19, and the current component detection circuit 19 generates an exciting current detection signal I _n and a torque current detection signal based on the output signal of the oscillator 17.
Find I _t . The excitation current command signal I _n ^* of the excitation current command circuit 10 and the excitation current detection signal I _n are added by an adder 11 with the polarity shown, and the deviation ΔI _n is input to the first current control circuit 12 . First current control circuit 1
2 is an excitation voltage component command signal that commands the magnitude of the voltage component in the axial direction of the excitation current according to the current deviation ΔI _n
Output V _n ^* . On the other hand, the torque current command signal I _t ^* of the speed control circuit 7 and the (torque current) detection signal I _t are added by the adder 13 with the polarity shown, and the deviation ΔI _t
is input to the second current control circuit 14. The second current control circuit 14 outputs a torque voltage component command signal V _t ^* that commands the magnitude of the voltage component in the torque current axis direction (orthogonal direction to the magnetic flux axis) according to the current deviation ΔI _t . Torque voltage component command signal V _t ^* is output to adder 16 . Further, the primary frequency command signal ω ₁ ^* and the excitation current detection signal _In are multiplied by a multiplier 15 , and the multiplied value is also input to an adder 16 . The adder 16 calculates the sum of the torque voltage component command signal V _t ^* and the signal V ₂ (corresponding to the secondary induced voltage) proportional to the product of the primary frequency command signal ω ₁ ^* and the exciting current detection signal I _n . Signal V _t0
Output ^* . The signal V _t0 ^* becomes the modified torque voltage component command signal V _t0 ^* . The excitation voltage component command signal V _n ^* and the modified _torque ^voltage component _command signal V _t0 ^* are input to the vector calculation circuit ²⁰ . This is performed based on the two-phase sine wave signal, and a three-phase AC voltage command signal (instantaneous value) v ^* is output.
The AC voltage command signal v ^* of the vector calculation circuit 20 is
Input to PWM inverter 2.

Next, its operation will be explained.

The speed control circuit 7 outputs a torque current command signal I _t ^* that commands the magnitude of the torque current of the induction motor 3 according to the speed deviation ΔN. The slip frequency command signal ω _s ^* and the torque current command signal I _t ^* have the following relationship.

ω _s ^* = 1/T ₂ I _n ^* I _t ^* ...(1) Here, T ₂ is the second-order time constant of the induction motor 3 (=
l _n +l ₂ /r ₂ ), l _n : excitation inductance, l ₂ : secondary leakage inductance, r ₂ : secondary circuit.

As is clear from equation (1), the excitation current command signal I _n
If ^* is constant, the slip frequency command signal ω _s ^* is proportional to the torque current command signal I _t ^* . The slip frequency calculation circuit 8 calculates equation (1) and outputs a slip frequency command signal ω _s ^* . The adder 9 calculates the equation (2): ω ₁ ^* =ω _s ^* +N (2), and outputs the primary frequency command signal ω ₁ ^* . In equation (2), the signals ω ₁ ^* and ω _s ^* are the primary angular frequency and slip angular frequency of the induction motor 3, and the speed detection signal N represents the rotational angular frequency. The oscillator 17 outputs two-phase sine wave signals sinω _{1 *} ^t , csω ₁ ^* t having frequencies corresponding to ^the primary frequency command signal ω 1 *.

FIG. 2 shows an example configuration diagram of the oscillator 17.

The primary frequency command signal ω ₁ ^* is input to the absolute value circuit 101 . The V/F converter 102 receives the output signal of the absolute value circuit 101 and outputs a pulse example signal proportional to the input signal. The primary frequency command signal ω ₁ ^* is also input to the polarity determining circuit 103 to determine its sign. The up-down counter 104 determines whether to up-count or down-count the pulse example of the V/F converter 102 in accordance with the output signal of the polarity determination circuit 103.
The count value of the counter 104 corresponds to the phase ω ₁ ^* t, and is input to predetermined addresses of the ROMs 105 and 106. The ROM 105 outputs a digital signal S _s (=sinω ₁ ^* t) corresponding to the sine wave signal,
Further, the ROM 106 outputs a digital signal S _e (=cosω ₁ ^* t) corresponding to the cosine wave signal. The digital signals S _s and S _e are input to a current component detection circuit 19 and a vector calculation circuit 20, respectively.

FIG. 3 shows an example configuration diagram of the current component detection circuit 19. U phase of current detection signal i of current detector 18,
When the V-phase currents are i _u and i _v , it is expressed as i _u =Isin(ω ₁ t+θ) i _v =Isin(ω ₁ t−2/3π+θ) (3). In equation (3), I is the magnitude of the current, and θ is the phase difference between the output signal S _s of the oscillator 17 and the current detection signal i _u . The current detection signals i _u and i _v are converted into two-phase currents by coefficient multipliers 107, 108, 109 and an adder 110 having weighting coefficients of 1, 1/√2, 2/√3, as shown in FIG. Detection signal i〓,
It is converted to i〓. The calculation is as shown in equation (4).

i = i _u = Isin (ω ₁ t + θ) i = 1/√3i _u +2/√3i _v = −Icos (ω ₁ t + θ) ...(4) Current detection signal i =, i = and sine wave signal S _s , S _e is D/
After being multiplied by A converters 111, 112, 113, 114 as shown in the figure, adders 115, 1
16, the signals are added with the polarity shown. The current component detection circuit 19 calculates equation (5).

I _n =i〓S _s −i〓S _e I _t =i〓S _s −i〓S _e ……(5) Substituting formula (4) into formula (5) and rearranging, we get the following formula .

I _n = Icom θ I _t = Isin θ (6) As is clear from equation (6), the excitation current component I _n and the torque current component I _t of the current detection signal i are detected as DC amounts.

Returning to FIG. 1, the first current control circuit 12 outputs an excitation voltage component command signal V _n ^* according to the current deviation ΔI _n between the excitation current command signal I _n ^* and the excitation current detection signal I _n ,
Further, the second current control circuit 14 outputs the torque voltage component command signal V _t ^* according to the current deviation ΔI _t between the ^torque current command signal I t * and _the torque current detection signal I _t .

By the way, the relationship between the primary voltage V, excitation voltage V _n and torque voltage V _t applied to the induction motor 3, and the relationship between the primary current I, excitation current I _n and torque current I _t is shown in a vector diagram as shown in Fig. 4. become. Figure a shows when the speed is high, and b shows when the speed is low. Fourth
As can be seen from the figure, the excitation current I _n and the torque current I _t
The voltage varies depending on the speed even _if the _relationship between
It can be seen that the speed varies greatly depending on the speed. Torque voltage V _t is expressed as in equation (7).

V _t ∝I _n ×ω ₁ ...(7) In Fig. 1, the torque voltage component command signal V _t ^*
A signal V ₂ obtained by multiplying the excitation current detection signal _In and the primary frequency command signal ω ₁ ^* is added to the corrected torque voltage component command signal V _t0 ^* . In this way, since the signal V _t0 ^* is obtained using the signal V ₂ which changes greatly depending on the speed, the exciting current detection signal I _n and the primary frequency can be adjusted without affecting the torque current I _t in any way. A modified torque voltage command signal V _t0 ^* having a value corresponding to the magnitude of the command signal ω ₁ ^* can be obtained. Note that the command signal I _n ^* is used instead of the excitation current detection signal I _n
Alternatively, a signal proportional to the magnitude of the magnetic flux φ shown in FIG. 4 may be used, and the speed detection signal N may be used instead of the primary frequency command signal ω ₁ ^* . Furthermore, if the excitation current detection signal I _n is constant, a coefficient multiplier may be used instead of the multiplier 15 and a signal proportional to the signal ω ₁ ^* or N may be input to the adder 16.

When the excitation voltage component command signal V _n ^* and the corrected torque voltage component command signal V _t0 ^* are obtained in this way, the vector calculation circuit 20 outputs a three-phase AC voltage command signal v ^* in the following manner.

FIG. 5 shows an example configuration diagram of the vector calculation circuit 20. Signals V _n ^* , V _t ^* , S _s , S _e are supplied to D/A converters 117, 118, 119, 1 which operate as multipliers.
20, and their outputs are input to adders 121,
At 122, the signals are added as shown in the figure to obtain two-phase AC voltage command signals v〓 ^* , v〓 ^* . Signal v〓 ^* , v〓 ^* is D/A
Converters 117-120 and adders 121, 12
2, the calculation of equation (8) is executed.

v〓 ^* =V _n ^* S _s +V _t ^* S _e v〓 ^* = −V _n ^* S _e +V _t ^* S _s ……(8) When formula (8) is rearranged, it becomes formula (9).

v〓 ^* =V ^* sin (ω ₁ ^* t+) v〓 ^* = −V ^* cos (ω ₁ ^* t+) ...(9) In this way, the two-phase AC voltage command signal v〓 ^* ,
v〓 ^* is obtained.

Note that V _* = √ ( _n ^* ) ² + ( _t ^* ) ² ... (10) = tan ^-1 (V _t ^* /V _n ^* ) ... (11).

The AC voltage command signals v〓 ^* , v〓 ^* are 1,
Coefficient unit 12 with weighting coefficients of 1/2, 2/√3
3,124,125. Furthermore, the outputs of the coefficient multipliers 124 and 125 are outputted to the adder 1 with the polarity shown in the figure.
26,127. Therefore, the three-phase AC voltage command signal v ^* (v _u ^* , v _v ^* , v _w ^* ) is expressed by the following equation.

v _u ^* =v〓 ^* =V ^* sin(ω ₁ ^* t+) v _v ^* =−1/2v〓 ^* +2/√3v〓 ^* =V ^* sin(ω ₁ ^* t−2/3π+) v _w ^* =-1/2v〓 ^* -2/√3v〓 ^* =V ^* sin(ω ₁ ^* t-4/3π+) ...(12) AC voltage command signal v ^* is proportional to the output voltage of PWM inverter 2, The switching element of the PWM inverter 2 is turned on and off according to the signal v ^* . The output voltage (phase voltage) of inverter 2 is the AC voltage command
It becomes proportional to v ^* .

By controlling as described above, the rotational speed of the induction motor 3 is controlled to be proportional to the speed command signal N ^* . At this time, the corrected torque voltage component command signal V _t0 ^* , which varies greatly depending on the rotational speed N, is approximately determined by the output signal V ₂ of the multiplier 15 that performs the calculation of equation (7). In the current control circuits 12 and 14, it is only necessary to perform an operation to correct the deviation of the current components I _n and I _t, respectively, so that fluctuations in the torque current I _t do not affect the torque voltage component V _t , and the excitation current I _n fluctuations in V t will affect the torque voltage component V _t that changes accordingly. Therefore, stable speed control can be performed over a wide speed range.

FIG. 6 shows the operating characteristics of the conventional method and the present invention. FIG. 6a shows a response waveform diagram when the torque current command signal I _t ^* changes according to the conventional method, and FIG. 6b shows a response waveform diagram when the torque current command signal I _t ^* changes according to the present invention. As you can see, it's the same picture.
According to the conventional method, the excitation current I _n oscillates when the torque current command signal I _t ^* changes, but according to the present invention, the excitation current I _n does not oscillate, and the torque current I _t also does not oscillate. It can be seen that the target value is followed in a non-oscillatory manner.

FIG. 7 shows another embodiment of the present invention.

In FIG. 7, the same symbols as in FIG. 1 indicate equivalents. The embodiment shown in Fig. 7 is an AC voltage command signal.
The means for obtaining V ^* is different from the embodiment shown in FIG. The excitation voltage component command signal V _n ^* of the current control circuit 12 and the torque voltage component command signal V _t ^* of the adder 16 are input to the vector addition circuit 24 . Vector addition circuit 2
4 performs the square root calculation of equation (10) and outputs a voltage amplitude command signal V ^* that commands the magnitude of the AC voltage. The voltage amplitude command signal V ^* of the vector addition circuit 24 is input to the voltage command calculation circuit 25. Further, the excitation voltage component command signal V _n ^* and the modified torque voltage component command signal V _t0 ^* are input to the divider 21 , and the output thereof is applied to the function generator 22 . The divider 21 and the function generator 22 perform arctangent calculation of equation (11) and output a signal according to the phase shown in FIG. The primary frequency command signal ω ₁ ^* and the phase signal are input to the oscillator 23 .

FIG. 8 shows a configuration diagram of an example of the oscillator 23.

In FIG. 8, part numbers 101 to 106 indicate parts equivalent to those in FIG. The phase signal is input to an A/D converter 128 and converted to a corresponding digital signal. The output signals of A/D converter 128 and counter 104 are input to digital adder 129. The adder 129 adds the digital signals corresponding to the phase and ω ₁ ^* t. Adder 12
The output signal of 9 is input to ROMs 130 and 131. Digital signals corresponding to two-phase sine wave signals S _s ′=sin (ω ₁ ^* +) and S _c ′=cos (ω ₁ ^* t+) are obtained from the ROMs 130 and 131 . Signal S _s ′,
S _c ′ is input to the voltage command calculation circuit 25, and the signal
S _s ′ and S _c ′ are input to the current component detection circuit 19 .

Returning to FIG. 7, the voltage amplitude command signal V ^* and the sine wave signals S _s ′, S _c ′ are input to the voltage command calculation circuit 25 . The voltage command calculation circuit 25 generates a voltage amplitude command signal.
Multiply V ^* by the sinusoidal signals S _s ′ and S _c ′ to obtain two-phase AC voltage command signals v〓 ^* , v〓 ^* , and then perform 2-phase to 3-phase conversion to obtain 3-phase AC voltage command signals v ^* (v _u ^* , v _v ^* ,
v _w ^* ). AC voltage command signal v ^* is PWM
Added to inverter 2.

FIG. 9 shows an example configuration diagram of the voltage command calculation circuit 25.

The voltage amplitude command signal V ^* and the sine wave signals S _s ′, S _c ′ are input to the D/A converters 132 and 133, and the voltage amplitude command signal V ^* and
S _s ′ and S _c ′ are multiplied. D/A converter 132,1
The output signal of 33 is a two-phase AC voltage command signal v〓 ^* ,
v〓 ^* . The AC voltage command signals v〓 ^* , v〓 ^* are processed by a coefficient unit 13 with weighting coefficients of 1, 1/2, and 2/√3, respectively.
4,135,136 are input as shown.
The outputs of the coefficient multipliers 135 and 136 are input to adders 137 and 138 with the polarities shown. In this way, three-phase AC voltage command signals v ^* (v _u ^* , v _v ^* , v _w ^* ) are obtained.

In the embodiment shown in FIG. 7, the same operation as that shown in FIG. 1 is performed, and the effect shown in FIG. 6 is obtained.
Furthermore, since the embodiment shown in FIG. 7 calculates the magnitude and phase of the voltage separately when obtaining the voltage command signal v ^* , it is suitable for program control using a digital configuration using a microcomputer.

〔Effect of the invention〕

As explained above, according to the present invention, the torque voltage component command signal on the torque current axis, which changes significantly depending on the rotational speed, is almost established by the product of the rotational speed and the excitation speed, so it can be used over a wide speed range. Torque control can be performed stably and with high response.

In addition, in the above embodiment, as a power converter
Although the case where a PWM inverter is used is shown, it can also be applied when a current source inverter or a cycloconverter is used. Further, although the above-described embodiment shows an analog configuration, it goes without saying that the present invention can also be applied to digital control using a microcomputer.

[Brief explanation of drawings]

FIG. 1 is a configuration diagram showing one embodiment of the present invention, and FIG.
3 and 3 are detailed configuration diagrams of examples of the oscillator or current component detection circuit in FIG. 1, FIG. 4 is a vector diagram for explaining the operation of FIG. 1, and FIG. 5 is an example of the vector calculation circuit in FIG. 1. A detailed configuration diagram, FIG. 6 is a characteristic diagram for explaining the effects of the present invention, FIG. 7 is a configuration diagram showing another embodiment of the present invention, and FIG. 8 and FIG. FIG. 2 is a configuration diagram showing an example of a voltage command calculation circuit. 2...PWM inverter, 3...Induction motor,
7...Speed control circuit, 8...Slip frequency calculation circuit, 9...Adder, 10...Exciting current command circuit,
12, 14... Current control circuit, 15... Multiplier,
16... Adder, 17... Oscillator, 19... Current component detection circuit, 20... Vector calculation circuit.

Claims (1)

[Claims] 1. An induction motor driven by a power converter;
A speed control means for commanding the magnitude of a torque current of the induction motor by comparing a speed command value and an actual speed value; a slip frequency calculation means for calculating a slip frequency corresponding to the torque current; frequency command means for determining a primary frequency command value from the sum of a slip frequency and a rotational frequency; excitation current command means for commanding the magnitude of an excitation current of the induction motor; a current component detection means for outputting a DC signal proportional to the magnitude of the torque current and the excitation current; and an excitation voltage component command value that is output by comparing the actual excitation current value calculated by the current component detection means and the excitation current command value. a first current control means for comparing the actual torque current value and the torque current command value and outputting a voltage component command value in terms of torque; and a voltage corresponding to the secondary induced voltage of the induction motor. calculation means, adds the torque voltage component command value and the output of the voltage calculation means to obtain a corrected torque voltage component command value, and calculates the voltage component of the torque of the primary voltage based on the corrected torque voltage component command value. 1. A control device for an induction motor, characterized in that the size of the induction motor is controlled. 2. The induction according to claim 1, wherein the voltage calculating means corresponding to the secondary induced voltage calculates it from the product of the actual excitation current value or the excitation current command value and the calculated primary frequency value. Electric motor control device. 3. The control device for an induction motor according to claim 1, wherein the voltage calculating means corresponding to the secondary induced voltage calculates the voltage from the product of the main magnetic flux of the induction motor and the calculated primary frequency value.