JPH08191600A - Device for controlling inverter current - Google Patents

Abstract

PURPOSE: To improve the responsiveness of a current control without saturating the output of an inverter. CONSTITUTION: The controlling device 100 of the current of an inverter which controls a magnetic flux component current 104 and the torque component current 103 in the output current of the inverter 2 driving an induction motor 1, on the basis of a magnetic flux component current command value 101 and a torque component current command value 102, is provided with a limiting circuit which limits the voltage command value 105 of a magnetic flux component and the voltage command value 106 of a torque component so that the output voltage of the inverter 2 may not be saturated, in accordance with a state of computation corresponding to the deviation of the magnetic flux component current from the magnetic flux component current command value and a state of computation corresponding to the deviation of the torque component current from the torque component current command value.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a current controller for an inverter suitable for use in a controller for an AC motor such as an induction motor or a permanent magnet type synchronous motor (DC brushless).

[0002]

2. Description of the Related Art In controlling an induction motor which is one of AC motors, in order to perform speed control with high accuracy in a wide operating frequency range, a primary three-phase AC current of the induction motor is detected, and a magnetic flux current component and a torque current are detected. There is a control device for decomposing into components and controlling both components as independent DC amounts (JP-A-57 / 57).
-199489 gazette "induction motor control device").

FIG. 7 is a block diagram showing a configuration relating to the control of the magnetic flux component current and the torque component current in the conventional induction motor controller as described above. In this figure, 1 is a three-phase induction motor, and 2 is a voltage type PWM (pulse width modulation) control inverter. The voltage-type PWM control inverter 2 is composed of a three-phase transistor / inverter, etc., and controls the pulse width of a DC voltage supplied from a DC power supply unit (not shown) to generate a three-phase AC voltage having an arbitrary voltage, frequency and phase. Converted to and output. Then, the three-phase AC voltage output from the voltage-type PWM control inverter 2 is applied to the three-phase induction motor 1 to drive the three-phase induction motor 1.

Reference numerals 3 and 4 denote current detectors, which detect the primary u-phase current and the w-phase current of the three-phase induction motor 1,
Output to the 2-phase to 3-phase converter 5. 2-phase to 3-phase converter 5
Is based on the primary current of the three-phase induction motor 1 input from the current detectors 3 and 4 and the output of the speed detector of the three-phase induction motor 1 (not shown) and the like by a known conversion method. The primary current of 1 is decomposed into a magnetic flux component and a torque component and output as a magnetic flux component current Id and a torque component current Iq. However, in the case shown in this figure, among the three-phase primary currents of the three-phase induction motor 1, the v-phase current not directly detected is the other two detected by the current detectors 3 and 4.
Calculated based on the phase current.

Numerals 12 and 13 are subtracters, which obtain deviation signals between the magnetic flux component current Id and the torque component current Iq and their respective command values Id ^* and Iq ^*, and output them to the current control circuits 6 and 7. Then, the current control circuits 6 and 7 obtain the magnetic flux component voltage command Vd ^* and the torque component voltage command Vq ^* by multiplying the input deviation signals by, for example, a predetermined proportional gain, and the two-phase-three-phase Output to the converter 8. The current control circuit 6 and 7, with respect to the magnetic flux component voltage command Vd ^* and torque component voltage command Vq ^*, multiplied by a predetermined clamping the output to a magnetic flux component voltage command Vd ^* and torque component voltage command Vq ^* is It is limited to a value within a predetermined range. This clamp will be described later.

The two-phase to three-phase converter 8 calculates the magnetic flux component voltage command Vd ^* and the torque component voltage command Vq ^* by a predetermined calculation, and the primary three-phase voltage commands (instantaneous values) Vu ^* , Vv ^* , Vw.
Convert to ^* and output. Then, the primary three-phase voltage command V
u ^* , Vv ^* , Vw ^* are limit circuits 9, 10 and 1
After being limited to a predetermined range by 1, voltage-type PWM
It is input to the control inverter 2 as a command value of the three-phase output voltage. The limit circuits 9, 10 and 11 apply upper and lower limits to each voltage command value so that each voltage command value does not exceed the maximum amplitude of the voltage-type PWM control inverter 2. Here, it is assumed that the maximum outputtable amplitude of the voltage-type PWM control inverter 2 is 1, and the limit values in the limit circuits 9, 10 and 11 are set to ± 1.

Now, here, the current control circuits 6 and 7, the two-phase to three-phase converter 8 and the limit circuits 9, 1 are used.
Details of the operations of 0 and 11 will be described. As described above, in the current control circuits 6 and 7, the magnetic flux component current Id
And a magnetic flux component voltage command Vd ^* and a torque component voltage command V based on a deviation signal between the torque component current Iq and the respective command values Id ^* and Iq ^*.
Although q ^* is obtained, the case where proportional and integral operations are used will be described here as an example.

In this case, in the current control circuits 6 and 7,
First, the internal voltage command value Vd of the magnetic flux component and the torque component
^* _a and Vq ^* _a are calculated by the following equations. _{^{Vd * a = K 1 (Id}} * -Id) + K 2 ∫ (Id * -Id) dt ...... (1) Vq * a = K 3 (Iq * -Iq) + K 4 ∫ (Iq * -Iq) dt ... (2) Here, K ₁ and K ₃ represent proportional gains, and K ₂ and K ₄ represent integral gains. However, the internal voltage command values Vd ^* _a and Vq ^* _a are values before being clamped. next,
With respect to the respective command values Vd ^* _a and Vq ^* _a , the magnitudes of the magnetic flux component voltage command Vd ^* and the torque component voltage command Vq ^* actually output to the two-phase to three-phase converter 8 are ± 1 at maximum, respectively. Clamp is applied so that. The value of this clamp is
This corresponds to the maximum outputtable amplitude value (= 1) of the voltage type PWM control inverter 2 described above.

On the other hand, two-phase in the two-phase to three-phase converter 8
The three-phase conversion can be performed, for example, by the following formula (3) according to the polar form component.
~ (5) is performed. _{^{Vu * = V 1 * sin (}} θ * + α) ......... (3) Vv * = V 1 * sin (θ * + α-2π / 3) ......... (4) Vw * = V 1 * sin (θ * + Α-4π / 3) (5) where θ ^* = ∫ (ωr / p + ωs ^* ) dt (6) V ₁ ^* = √ ((Vd ^* ) ² + (Vq ^* ) ² ) ... …… (7) α = tan ⁻¹ (Vq ^* / Vd ^* ) ………… (8)

Where θ ^* is the predicted position of the secondary magnetic flux of the three-phase induction motor 1, ωr is the rotational speed, p is the number of pole pairs, ωs ^* is the slip frequency command, and V ₁ ^* is the amplitude command of the primary voltage vector. , Α is the angle (phase difference angle) of the primary voltage vector with respect to the secondary magnetic flux. In this case, the amplitude command V ₁ ^* of the primary voltage vector matches the amplitude voltage in the voltage-type PWM control inverter 2. Therefore, when the amplitude command V ₁ ^* of the primary voltage vector is larger than 1, each instantaneous voltage Vu ^* , Vv ^* ,
Vw ^* changes the limit circuits 9 and 1 in accordance with the change of the phase θ ^*.
Occurs when the clamp value at 0 and 11 exceeds ± 1. At this time, the limit circuits 9, 10 and 11 are clamped, the output of the voltage-type PWM control inverter 2 is saturated, and the three-phase output voltage waveform is distorted.

[0011]

In the current control circuits 6 and 7, as described above, the magnetic flux component voltage command Vd ^*.
And torque component voltage command Vq ^* are ± 1 at maximum.
, The composite vector v ^* of each component on the dq axes is within the range of the square shown in FIG.
(Within the range of 1). On the other hand, the instantaneous value of the primary three-phase voltage obtained in a two-phase three-phase converter 8, as shown in equation (7), the resultant vector v ^* size (= V ₁
Since it is obtained as a periodic function whose amplitude is ^* ), the relationship of V ₁ ^* ≤ 1 is established only when the composite vector v ^{* of} each component is within the circle of radius 1 centered on the origin O shown in FIG. To establish.

Therefore, the vector v ^* synthesized from the magnetic flux component voltage command Vd ^* and the torque component voltage command Vq ^{* .}
8 is outside the circle with radius 1 centered on the origin O indicated by the diagonal lines in FIG. 8 and within the square of side 2 the amplitude command V ₁ ^* of the primary voltage vector becomes larger than 1. As a result, the output of the voltage type PWM control inverter 2 will be saturated. That is, although the current control circuits 6 and 7 clamp the magnetic flux component voltage command Vd ^* and the torque component voltage command Vq ^* within the range of ± 1, the voltage type PWM control is performed. There is a problem that the output of the inverter 2 is controlled in the saturation region.

Further, as described above, the current control circuit 6
In 7 and 7, when the voltage command values of the magnetic flux component and the torque component are obtained by a calculation including an integral element such as a proportional + integral calculation, there are the following problems. Figure 9
FIG. 6 is a diagram in which the horizontal axis represents time and the vertical axis represents the voltage of the torque component. In this case, it is assumed that the magnetic flux component voltage command Vd ^{* (} not shown ⁾ is set to be constant at 1 / √2. It is also assumed that at time t1, the sign of the deviation signal between the torque command current command value Id and the current command value Id ^* has changed. Therefore, in this case, at time t1,
Equation (2) proportional portion K ₃ of the voltage command value Vq ^* _a torque components shown in (Iq ^* -Iq) is changed to the P1 P2, also integral portion K ₄ ∫ (Iq ^* -Iq) After time t1, dt changes so as to gradually decrease from I1.

Under the above conditions, the internal voltage command value Vq ^* _a (= P1 + I) of the torque component is obtained from time 0 to t1.
1) is larger than 1, the torque component voltage command Vq ^* output from the current control circuit 7 is limited to 1. Then, at time t1, when the deviation signal changes as described above, the torque component voltage command value Vq ^* _a (= P2 + I
1) is 1 or less, the torque component voltage command Vq
^* Coincides with the voltage command value Vq ^* _a, and gradually decreases after time t1.

However, in this case, assuming that the magnetic flux component voltage command Vd ^* is constant at 1 / √2 as described above, the range in which the torque component voltage command Vq ^* shaded in the figure exceeds 1 / √2 ( In the range of (Vq ^* ) ² > 1- (Vd ^* ) ² , the output of the voltage-type PWM control inverter 2 becomes saturated. Therefore, in the case shown in this figure, if the torque component voltage command Vq ^* drops to 1 / √2 at time t2, the voltage-type PWM control inverter 2 outputs 1
The next three-phase voltages Vu, Vv, and Vw are from time 0 to t2,
It is in a saturated state. That is, although the deviation signal changes at time t1, the actual primary three-phase voltages Vu, Vv, Vw output from the voltage-type PWM control inverter 2 are saturated from time t1 to t2. You will not be able to escape.

Although the delay of the control relating to the voltage command value of the torque component has been described with reference to FIG. 9, the same can be said for the voltage command value of the magnetic flux component. As described above, when the voltage command values of the magnetic flux component and the torque component are obtained by the proportional + integral calculation, the limit value of the voltage command output from the current control circuits 6 and 7 corresponds to the saturation value in the voltage type PWM control inverter 2. Therefore, there is a problem that the voltage control in the voltage-type PWM control inverter 2 is delayed, and as a result, the responsiveness of the current control is deteriorated.

The present invention has been made under such a background, and an inverter current control device capable of improving the responsiveness of the current control without saturating the output of the inverter used for the current control. The purpose is to provide.

[0018]

According to the first aspect of the present invention,
In the current control device of the inverter that controls the magnetic flux component current and the torque component current in the output current of the inverter that drives the AC motor, based on the magnetic flux component current command value and the torque component current command value, the magnetic flux component current command value According to a deviation between the magnetic flux component current and the magnetic flux component current, first computing means for computing a voltage command value of the magnetic flux component of the output voltage of the inverter, and a deviation between the torque component current command value and the torque component current The inverter according to the second computing means for computing the voltage command value of the torque component of the output voltage of the inverter, the computing state in the first computing means, and the computing state in the second computing means. Of the magnetic flux component and the voltage command value of the torque component so as to prevent the output voltage from being saturated. That.

According to a second aspect of the present invention, in the first aspect of the invention, at least one of the first arithmetic means and the second arithmetic means performs at least a proportional arithmetic operation and an integral arithmetic operation. By a calculation including, to calculate the voltage command value of the magnetic flux component or the voltage command value of the torque component,
The limiting means limits the voltage command value of the magnetic flux component or the voltage command value of the torque component so that the proportional calculation has priority over the integral calculation.

[0020]

According to the above-mentioned structure, the limiting means sets the voltage command of the magnetic flux component so that the output voltage of the inverter is not saturated depending on the calculation state of the first calculation means and the calculation state of the second calculation means. The value and the voltage command value of the torque component are limited. Then, a voltage having a magnetic flux component and a torque component according to the voltage command value of each component is output from the inverter. Therefore, the output voltage of the inverter does not saturate, and compared with the case where it saturates, the magnetic flux component current and the torque component current of the alternating current flowing through the AC motor are based on the magnetic flux component current command value and the torque component current command value. It is possible to improve controllability with.

Further, according to the above configuration of the present invention, at least one of the first calculating means and the second calculating means calculates the magnetic flux component by the calculation including the proportional calculation and the integral calculation. When calculating the voltage command value or the voltage command value of the torque component, the limiting means first limits the voltage command value of the magnetic flux component or the voltage command value of the torque component based on the proportional calculation, and then performs the integral calculation. Based on this, the voltage command value of the magnetic flux component or the voltage command value of the torque component is limited. Therefore, the voltage command value of each component can be changed with good responsiveness in accordance with the deviation between the magnetic flux component current command value and the magnetic flux component current or the deviation between the torque component current command value and the torque component current.

[0022]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An embodiment of the present invention will be described below with reference to the drawings. 1 is a block diagram showing the configuration of an induction motor controller using an inverter current controller according to an embodiment of the present invention. In this figure, a current control circuit 100 is the inverter current controller of the present invention. It is the corresponding part. In FIG. 1, parts corresponding to those in FIG. 7 are designated by the same reference numerals, and the description thereof will be omitted. In the control device for the induction motor shown in FIG. 1, a current control circuit 100 is provided instead of the current control circuits 6 and 7 shown in FIG. 7, and the limit circuits 9, 10 and 11 are omitted, and the two-phase control circuit is provided. -Primary three-phase voltage command Vu ^* , Vv ^* , Vw output from the three-phase converter 8
^* Is directly input to the voltage type PWM control inverter 2.

The current control circuit 100 outputs the magnetic flux component current Id and the torque component current Iq output from the two-phase to three-phase converter 5 and the respective command values Id ^* and Iq ^* ,
The magnetic flux component voltage command Vd ^* and the torque component voltage command Vq ^* are calculated by inputting from the input ends 104, 103, 101 and 102, and are output from the output ends 105 and 106.

FIG. 2 is a block diagram showing a detailed structure of the current control circuit 100 shown in FIG. The current control circuit 100 is composed of a one-chip microprocessor (DSP; Digital Signal Processor) dedicated to digital signal processing. In this case, the current control circuit 100 shown in FIG. 2 is configured to calculate the magnetic flux component voltage command Vd ^* and the torque component voltage command Vq ^{* based} on the equations (1) and (2). 2, the input terminals 101 to 104 and the output terminals 105 and 106
Corresponds to that shown in FIG.

In FIG. 2, the magnetic flux component current command value Id ^* input from the input terminal 101 and the magnetic flux component current Id input from the input terminal 104 are input to the subtractor 130 and calculated by (Id ^* -Id). The generated deviation signal is output from the subtractor 130. The output of the subtractor 130 is input from the input terminal X to the sample hold 131. The sample hold 131 samples a signal input from the input terminal X based on a clock signal of a predetermined cycle supplied from an external clock oscillation circuit (not shown) to the terminal b via the clock input terminal 195,
Hold the input value for one cycle of the clock signal,
Output from the output terminal Y.

Reference numerals 132 and 133 denote multipliers. The multiplier 132 obtains the result of multiplying the proportional gain (K ₁ ) input from the input terminal 191 by the output of the sample hold 131, and the multiplier 133 inputs the input terminal. Integral gain (K ₂ ) input from 192 and sample hold 13
The result of multiplication with the output of 1 is obtained. The magnetic flux component current command value Id ^* , the torque component current command value Iq ^* , the proportional gain and the integral gain are respectively input from a central control unit (CPU), a register and the like (not shown). The multiplication result of the multiplier 132 is output from the output terminal Y and input to the terminal X of the limit circuit 134. The limit circuit 134 determines the value output from the output terminal Y according to the following conditions according to the values input from the terminal X, the terminal MAX, and the terminal MIN. However, the value input from the terminal MAX is always larger than the value input from the terminal MIN. 1) When the value input from the terminal X is greater than or equal to the value input from the terminal MAX: The value input from the terminal MAX is output from the output terminal Y. 2) When the value input from the terminal X is less than or equal to the value input from the terminal MIN: The value input from the terminal MIN is output from the output terminal Y. 3) When the value input from the terminal X is smaller than the value input from the terminal MAX and is larger than the value input from the terminal MIN: The value input from the terminal X is directly output from the output terminal Y. That is, the limit circuit 134 limits the maximum value and the minimum value of the signal input from the terminal X to the values input from the terminals MAX and MIN, and outputs from the terminal Y.

The output of the limit circuit 134 is input to one of the two input terminals of the adder 135. On the other hand, the multiplication result of the multiplier 133 is output from the output terminal Y and input to one input terminal of the adder 136, and the output of the adder 136 is input to the input terminal X of the limit circuit 137. The limit circuit 137 outputs from the terminal Y an output obtained in the same manner as the limit circuit 134 according to the signals input from the terminal X, the terminal MAX and the terminal MIN. Then, the signal output from the terminal Y is input to the other input terminal of the adder 135. Therefore, the adder 135 outputs the result of adding the output of the limit circuit 134 and the output of the limit circuit 137. Also, the limit circuit 137
The signal output from the output terminal Y of is also input to the X terminal of the unit delay 138.

This unit delay 138 has a terminal b.
The signal input from the terminal X is delayed by one cycle of the clock signal input from the terminal Y and output from the terminal Y. The output of the unit delay 138 is input to one input terminal of the adder 136. Therefore, the unit delay 138 and the adder 136 constitute an integrator, and the integrated value of this integrator is the limit circuit 13.
It is always limited to the upper and lower limits of 7. According to the above configuration, when the limit circuits 134 and 137 output the values input from the respective input terminals X from the output terminal Y as they are, the voltage command of the magnetic flux component represented by the equation (1) is given. A digital signal corresponding to the value Vd ^* _a is output from the terminal 105 as the voltage command value Vd ^* .

On the other hand, the signal processing of the torque component is performed by the circuits denoted by the reference numerals 150 to 158 configured in the same manner as the circuits denoted by the reference numerals 130 to 138 described above. Then, when the limit circuits 154 and 157 output the values input from the respective input terminals X as they are from the output terminal Y, the voltage command value Vq ^* _a of the torque component represented by the equation (2) is set. The corresponding digital signal is output from the terminal 106 as the voltage command value Vq ^* . In addition,
The proportional gain (K ₃ ) and the integral gain (K ₄ ) in the signal processing of the torque component are input from the input terminal 193 and the input terminal 194, respectively.

Next, the limit circuits 134, 137 and 15
The input signals of the terminals MAX and MIN of 4 and 157 will be described. Reference numeral 139 denotes an adder, which obtains the result of adding the proportional component and the integral component of the internal voltage command value of the magnetic flux component by adding the output of the multiplier 132 and the output of the adder 136 to obtain the limit value generating circuit 200. Output to the input terminal X1. Reference numeral 159 denotes an adder similar to 139, which limits the result obtained by adding the proportional component and the integral component of the internal voltage command value of the torque component obtained by adding the output of the multiplier 132 and the output of the adder 136. Output to the input terminal X2 of the value generation circuit 200.

The limit value generating circuit 200 is a signal processing circuit composed of a plurality of arithmetic circuits, and sets the maximum value of the voltage command value of the magnetic flux component to the terminal Y1 and the minimum value in accordance with the values input from the input terminals X1 and X2. The value is output from the terminal Y2, the maximum value of the voltage command value of the torque component is output from the terminal Y3, and the minimum value is output from the terminal Y4. However, the limit value generation circuit 20
The value output from each terminal of 0 is a value within the range of ± 1. The details of the limit value generating circuit 200 will be described later.

Each output of the limit value generating circuit 200 is input to the circuit blocks 170 and 180.
The circuit block 170 includes subtractors 170 and 173 and clamp circuits 172 and 174 that clamp the upper and lower limit values of the output signal at ± 1. Then, the outputs of the terminals Y1 and Y2 of the limit value generation circuit 200 are directly connected to the terminals MAX and M of the limit circuit 134.
The result of subtracting the output of the limit circuit 134 from the output of the terminal Y1 and the terminal Y2 is output to IN and is output to the clamp circuits 172 and 174, respectively. Then, the outputs of the clamp circuits 172 and 174 are input to the terminal MAX and the terminal MIN of the limit circuit 137, respectively. With this configuration, the voltage command value Vd of the magnetic flux component
^In the limit circuit 134, the proportional part in ^* is limited within the range of the maximum value and the minimum value of the voltage command value of the magnetic flux component output from the limit value generation circuit 200, while the integral part is limited in the limit circuit 137. It is limited to a value obtained by subtracting the proportional amount from the maximum value and the minimum value of the voltage command value of the magnetic flux component output from the limit value generation circuit. Therefore, the terminal 105, which is a value obtained by adding the proportional component and the integral component,
The upper and lower limit values of the voltage command value Vd ^* output from the maximum value output from the limit value generation circuit 200 (terminal Y1)
And the range of the minimum value (terminal Y2).

On the other hand, the circuit block 180 is similar to the circuit block 170 in that the clamp circuits 18 for clamping the subtracters 180 and 183 and the upper and lower limit values of the output signal to ± 1.
It is composed of four circuits 2,184. Then, each limit value is output to each terminal MAX and terminal MIN of the limit circuits 154 and 157. Therefore, similarly to the magnetic flux component, the upper and lower limit values of the voltage command value Vq ^* of the torque component output from the terminal 106 are also the maximum value output from the terminal Y3 and the minimum value output from the terminal Y4 of the limit value generating circuit 200. Is limited to the range of.

Next, the details of the limit value generating circuit 200 will be described with reference to FIGS. 3 and 4. FIG. 3 and FIG. 4 are diagrams showing the dq plane similarly to FIG. 8, and are explanatory diagrams showing limit value generation methods by two different methods in the limit value generation circuit 200, respectively.

In FIG. 3, Vd ^* _a0 is a vector corresponding to the internal voltage command value of the magnetic flux component input from the terminal X1 of the limit value generating circuit 200, and Vq ^* _a0 is the terminal X.
2 shows a vector corresponding to the internal voltage command value of the torque component input from 2. In this case, Vd ^* _a0 is (0.
8, 0) and Vq ^* _a0 are ( _0, 0.8 × √3). Therefore, the size of the combined vector Va (0.8, 0.8 × √3) of the vector Vd ^* _a0 and the vector Vq ^* _a0 is 1.6, and the phase difference angle θa (corresponding to α in equation (8)) is π
/ 3. In such a case, in order not to saturate the output of the voltage-type PWM control inverter 2 (see FIG. 1), the vector Vd ^* _a0 and the vector Vq ^* _a0 are limited so that the combined vector Va has a magnitude of 1 or less. , That is, within the circle of radius 1 shown in this figure.

Therefore, the limit value generating circuit 200 controls each component V so that the magnitude of the composite vector Va becomes 1 or less.
The maximum and minimum limits for d ^* _a0 and Vq ^* _a0 are generated as follows. FIG. 3 shows a limit value generation circuit 200.
In the figure, the case where a limit value for limiting each of the two components is generated with the phase difference θ _a held is shown. That is, in this case, the limit value generation circuit 200
Then, the vector Vd ^* ₁₀ corresponding to the maximum value of the voltage command value Vd ^* of the magnetic flux component becomes (cos θ _a , 0), and the vector Vq ^* corresponding to the maximum value of the voltage command value Vq ^* of the torque component ^.
The limit value of each component is generated so that ₁₀ becomes (0, sin θ _a ). In this case, the size of the combined vector v ^* ₁₀ of the vector Vd ^* ₁₀ and the vector Vq ^* ₂₀ is 1.

Next, another method of generating a limit value in the limit value generating circuit 200 will be described with reference to FIG. The internal voltage command value Vd ^* _{a0 of the} magnetic flux component and the internal voltage command value Vq ^* _a0 of the torque component shown in FIG. 4 are the same vectors as those shown in FIG. In this case, the voltage command value of the magnetic flux component is prioritized, and when the internal voltage command value Vd ^* _a0 is ± 1 or less, the voltage command value of the magnetic flux component is not limited. Therefore, in this case, the voltage command value Vd of the magnetic flux component
^The vector Vd ^* ₂₀ corresponding to the maximum value of ^* matches the internal voltage command value Vd ^* _a0 . For the voltage command value of the torque component, the size of the combined vector v ^* ₂₀ of the vector Vd ^* ₂₀ and the vector Vq ^* ₂₀ corresponding to the maximum value of the voltage command value Vq ^* of the torque component is set to 1. Then, the magnitude of the vector Vq ^* ₂₀ is determined based on the expression ± √ (1- (Vd ^* _a0 ) ² ). Then, the vector Vq ^* determined as described above
Each limit value corresponding to ₂₀ is output from the limit value generation circuit 200.

Next, the phase difference angle θ explained with reference to FIG.
An example of the configuration of the limit value generating circuit 200 in the case of generating each limit value so that _a becomes constant will be described with reference to FIG. The input / output terminals X1 and X shown in FIG.
2, Y1, Y2, Y3 and Y4 correspond to the terminals shown in FIG. In this figure, 210 is an arctangent function circuit, which is used for inputting the signal input from the terminal X1 and the terminal X2.
Based on the signal input from tan ^-1 (X2 / X
The phase signal (corresponding to θ _a shown in FIG. 3) corresponding to 1) is output. When X1 = 0, the arctangent function circuit 21
0 outputs π / 4 or −π / 4. Reference numeral 211 is a cosine function circuit, which outputs a cosine function value (corresponding to cos θ _a ) based on the phase signal input from the arctangent function circuit 210. Reference numeral 212 denotes a comparator circuit which outputs "1" when the value input from the input terminal l is equal to or more than the value input from the input terminal r, and outputs "0" otherwise. Is the output of the cosine function circuit 211,
On the other hand, the 0 signal output from the constant signal generation circuit 213 is input to the terminal r.

Reference numerals 214 and 215 denote selectors,
When "1" is input to the input terminal b, the value input from the terminal t is selected and output, and when "0" is input to the input terminal b, the value input from the terminal f is selected. Select and output. Note that one signal output from the constant signal generation circuit 216 is input to the terminal f of the selector 214, and the output of the cosine function circuit 211 is input to the terminal t, while the selector 21 is input.
The -1 signal output from the constant signal generation circuit 217 is input to the terminal t of 5, and the output of the cosine function circuit 211 is input to the terminal f.

On the other hand, the output of the arctangent function circuit 210 is also supplied to the sine function circuit 221, and the sine function value (corresponding to sin θ _a ) obtained by the sine function circuit 221 corresponds to the terminal l of the comparison circuit 222. It is supplied to the terminal t of the selector 224 and the terminal f of the selector 225. The comparison circuit 2
22, the selectors 224 and 225, and the constant signal generation circuits 223, 226, and 227 are configured similarly to the above-described comparison circuit 212, selectors 214 and 215, and constant signal generation circuits 213, 216, and 217.

With the above configuration, limit value generating circuit 200 shown in FIG. 5 outputs the following signals from output terminals Y1, Y2, Y3 and Y3 based on the signals input from input terminals X1 and X2. 1) In the case of X1 ≧ 0 Y1 = cos (tan ⁻¹ (X2 / X1)) Y2 = −1 2) In the case of X1 <0 Y1 = 1 Y2 = cos (tan ⁻¹ (X2 / X1)) 3) X2 When ≧ 0 Y3 = sin (tan ⁻¹ (X2 / X1)) Y4 = −14 4) When X2 <0 Y3 = 1 Y4 = sin (tan ⁻¹ (X2 / X1))

As described above, in the limit value generating circuit 200 shown in FIG. 5, the phase difference angle θ _a is the same and has a magnitude according to the internal voltage command values Vd ^* _a and Vq ^* _a of the magnetic flux component and the torque component. The limit values of the magnetic flux component and the torque component corresponding to the combined vector are generated such that

Next, the operation of the current control circuit 100 shown in FIG. 1 configured as described above will be described with reference to FIG. Similar to FIG. 9, FIG. 6 is a diagram in which the horizontal axis represents time and the vertical axis represents the voltage of the torque component. In the case shown in this figure, the limit value generation circuit 20 in the current control circuit 100 is
It is assumed that 0 is configured as shown in FIG. Further, in this case, the phase difference angle θ _a of the combined vector of the magnetic flux component and the torque component is π / 3. Where time t _a
At, the torque command current command value Id and the current command value I
It is assumed that the sign of the deviation signal of d ^* has changed.

[0044] In this case, during the period from time 0 to t _a, the voltage command value of the torque component Vq ^* (= P11 + I11) is, sin
Limited to (π / 3). Here, in the period from time 0 to t _a, for example, the proportional content of P11 (K ₃ (Iq ^* -
Iq)) is 0.5, the integral I11 (K ₄
∫ (Iq ^* −Iq) dt) is sin (π / 3) −0.
It is limited to 5 (about 0.366). Then, at time t _a, the code changes of the deviation signal, proportional amount from P11 P12
Altered and when to, at time t _a, since the integral component is I11, the voltage command value Vq ^* of a torque component, changes, for example, as shown in FIG. That is, from the point of time t _a the sign of the deviation signal changes, without delay, the voltage command value of the torque component Vq ^* starts to change. Incidentally, during the period from time 0 to _{t a, cos (π / 3} ) so as not to exceed the limit is applied at the same time also to the voltage command value Vd of the magnetic flux components ^*.

As described above, in the current control circuit 100, the magnetic flux component voltage command Vd ^* and the torque component voltage command Vq ^* are limited according to the relationship between the two components, so that the two-phase to three-phase converter 8 is used. Primary-phase three-phase voltage commands Vu ^* , V input to the voltage-type PWM control inverter 2 via
v ^* and Vw ^* never become command values within the saturation region. Further, in the current control circuit 100, the proportional component of the voltage command value of each component is prioritized, and the integral component of each component is clamped corresponding to the maximum output amplitude of the voltage-type PWM control inverter 2. Therefore, for example, as shown in FIG. 6, the command value can be changed without delay with respect to the change of the deviation signal. Therefore, compared to the conventional
It is possible to reduce overshoot or undershoot in current control.

Although the embodiments of the present invention have been described above with respect to the induction motor, for example, in a permanent magnet type synchronous motor (DC brushless) used in an AC servo, a similar inverter is used with an exciting current command set to 0. Since the current control is performed, the present invention can be applied to a permanent magnet type synchronous machine based on the same principle as above.

[0047]

As described above, according to the invention described in claim 1, the limiting means outputs the output of the inverter in accordance with the calculation state in the first calculation means and the calculation state in the second calculation means. Since the voltage command value of the magnetic flux component and the voltage command value of the torque component are limited so that the voltage is not saturated,
The controllability of the magnetic flux component current and the torque component current of the AC current flowing through the AC motor can be improved as compared with the case where the output voltage of the inverter is not saturated and is saturated. That is, there is an effect that the response of the current control can be improved without saturating the output of the inverter used for the current control.

According to the second aspect of the invention, the proportional calculation has priority over the integral calculation in the limiting means. Therefore, the voltage command value of each component can be changed with good responsiveness according to the change of the current deviation of each component, so that the overshoot or undershoot in the current control can be reduced.

[Brief description of drawings]

FIG. 1 is a block diagram showing the configuration of an induction motor control device using an inverter current control device according to an embodiment of the present invention.

FIG. 2 is a block diagram showing a configuration example of a current control circuit 100 shown in FIG.

FIG. 3 is a plan view of d (magnetic flux component) -q (torque component) for explaining a method of limiting each command value in the current control circuit 100 shown in FIG.

4 is a dq plan view for explaining another method of limiting each command value in the current control circuit 100 shown in FIG. 2. FIG.

5 is a block diagram showing an example of an internal configuration of a limit value generating circuit 200 shown in FIG.

FIG. 6 is a current control circuit 100 according to an embodiment of the present invention.
3 is a time chart for explaining the operation of FIG.

FIG. 7 is a block diagram showing a configuration of a control device for an induction motor using a conventional inverter current control device.

FIG. 8 is an explanatory diagram showing a relationship between an AC voltage and a biaxial (magnetic flux component-torque component) voltage.

FIG. 9 is a time chart for explaining the operation of a conventional inverter current control device.

[Explanation of symbols]

 1 3 Phase Induction Motor 2 Voltage Type PWM Control Inverter 100 Current Control Circuit 170 Circuit Block 180 Circuit Block 200 Limit Value Generation Circuit

Claims (2)

[Claims] 1. A current control device for an inverter, which controls a magnetic flux component current and a torque component current in an output current of an inverter that drives an AC motor, based on a magnetic flux component current command value and a torque component current command value. First computing means for computing a voltage command value of a magnetic flux component of the output voltage of the inverter according to a deviation between the magnetic flux component current command value and the magnetic flux component current; the torque component current command value and the torque component current A second calculation means for calculating a voltage command value of a torque component of the output voltage of the inverter according to a deviation between the second calculation means, a calculation state in the first calculation means and a calculation state in the second calculation means. Accordingly, a limiting unit is provided for limiting the voltage command value of the magnetic flux component and the voltage command value of the torque component so that the output voltage of the inverter is not saturated. A current control device for an inverter, characterized in that: 2. The at least one arithmetic means of the first arithmetic means and the second arithmetic means is a voltage command value of the magnetic flux component or a voltage command of the torque component by an operation including a proportional operation and an integral operation. 2. A value is calculated, and the limiting means limits the voltage command value of the magnetic flux component or the voltage command value of the torque component so that the proportional calculation has priority over the integral calculation. Inverter current controller.