JPH0247126B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION [Technical Field of the Invention] The present invention relates to a current mirror circuit, and is intended to stabilize its output current.

[Technical background of the invention]

A conventional current mirror circuit is constructed as shown in FIG. PNP transistor Q1,
The base of Q2 is common and connected to the constant voltage source 1 together with the collector of the PNP transistor Q1. The emitters of transistors Q1 and Q2 are connected to constant voltage source 2 via resistors _R1 and _R2 , respectively. A load 3 is connected to the collector of the transistor Q2.

In the current mirror circuit described above, the values of the resistors R ₁ and R ₂ are generally equal, and the transistors Q1 and Q
2 is ideally treated as having a good set of characteristics. That is, the base-emitter voltage V _BE of a transistor is a function of the collector current,
It is assumed that each transistor Q1 and Q2 operates with the same base-emitter voltage V _BE and, as a result, operates with the same collector current. transistor Q
If the base currents of transistors Q1 and Q2 are ignored, the collector current of transistor Q1 is equal to the constant current Iref, and therefore the output current Iout is also considered to be equal to the constant current Iref.

[Problems with background technology]

Incidentally, the collector current Ic of a transistor operating in the forward region is generally expressed by the following equation.

Ic = (1+V _CE /V _A ) Is exp (V _BE /V _T ) ...(1) However, Is: Reverse saturation current V _CE ; Collector-base voltage V _T ; Thermoelectromotive force V _A ; Early voltage As can be seen from equation (1), transistors Q1, Q
Even if 2 are operating at the same V _BE , for example,
If V _CE changes, Ic will change.

The above change in V _CE is particularly important for transistor Q2
This occurs when the power supply voltage fluctuates when the collector potential of is determined from the load 3 side. This will be explained below using numerical examples.

V _CE of transistors Q1 and Q2 respectively
Let V _CE1 , V _CE2 and V _BE be V _BE1 and V _BE2 . Since the bases of transistors Q1 and Q2 are common,
Ignoring the base current, assuming that Is is equal, and assuming that the characteristics of transistors Q1 and Q2 are ideally matched from now on.

R ₁ Iref + V _BE1 = R ₂ Iout + V _BE2 ...(2) From equation (1), V _BE = V _T {Ic/(1+VCE/V _A )Is} ...(3) Therefore, R ₁ Iref+V _Tlo {Iref/( 1+V _CE1 /V _A )Is}=
R ₂ Iout + V _Tlo {Iout/(1+V _CE2 /V _A ) Is}...(4). R ₁ and R ₂ are the values of the resistors R ₁ and R ₂ . (Four)
Although Iout cannot be immediately determined from the formula, Iref=100 [μA], V _CE1 =0.7 [V], V _CE2 =6
[V], R ₁ = R ₂ = 300 [Ω], V _A = 20 [V], V _T = 26
[mV], Is ₁ = Is ₂ = 4.2×10 ^-16 [A], and when Iout is calculated using a computer, Iout = 110 [μA] is obtained. Under this condition, the output current of transistor Q2 is 10 compared to the input current of transistor Q1
〔%〕It has increased. In other words, the power supply voltage fluctuates,
When V _CE2 changes, the output current Iout also changes, and the accuracy of the current mirror circuit cannot be maintained.

[Purpose of the invention]

The present invention has been made in view of the above circumstances, and an object of the present invention is to provide a current mirror circuit that can accurately and stably maintain an output current against the above-mentioned early effect of transistors with a simple configuration. shall be.

[Summary of the invention]

In this invention, in order to achieve the above object,
For example, as shown in Figure 1, the emitter of transistor Q12 on the current output side is connected to a first constant voltage source (power supply) via resistors R ₁₂ and R ₁₃ , and
A current path formed by a resistor R ₁₄ is connected between the connection point of R ₁₂ and R ₁₃ and the second constant voltage source (ground potential section). As a result, even if the power supply voltage rises more than necessary, the _VBE of the transistor Q12 is lowered by the voltage drop of the resistor _R13 , thereby suppressing fluctuations in the output current Iout due to the Early effect.

[Embodiments of the invention]

Embodiments of the present invention will be described below with reference to the drawings.

FIG. 1 shows an embodiment of the present invention, in which the bases of transistors Q11 and Q12 and the transistor Q
The collector of the transistor Q11 is connected to the ground potential section via the constant current source 11, and the collector of the transistor Q12 is connected to the ground potential section via the load 12.
Also, the emitter of transistor Q11 is a resistor.
It is connected to the power supply line 13 via R ₁₁ , and the emitter of the transistor Q12 is connected to the power supply line 13 via resistors R ₁₂ and R ₁₃ in series. next,
In the circuit of the present invention, a resistor _R15 is connected between the connection point 14 between the resistors _R12 and _R13 and the ground potential section. The circuit with resistor R ₁₅ is the output transistor Q
A compensation circuit is formed to suppress an increase in the emitter current of 12.

One embodiment of the present invention is constructed as described above, and its characteristic operation will be explained. At a certain value of the power supply voltage V _CC , a certain current I _B flows through the resistor R ₁₄ . When V _CC goes above that value, the resistor
The voltage applied to R ₁₄ also increases, the current flowing increases, and the voltage drop across resistor R ₁₃ also increases. As a result, the base-emitter potential of the transistor 12
Output current Iout tries to increase as V _BE decreases
suppresses the effect and cancels the early effect.

Determining the voltage balance equation for the above circuit, V _BE1 + R ₁₁ Iref = V _BE2 + R ₁₂ Iout + R ₁₃ (Iout + I _B )
...(5) V _BE1 ; Base-emitter potential of transistor Q11 V _BE2 ; Base-emitter potential of transistor Q12 R ₁₁ ; Value of resistor R ₁₁ , R ₁₂ ; Value of resistor R ₁₂ , R ₁₃ ; Resistor R ₁₃ , from formula (3), V _BE1 = V _Tlo {Iref/(1+V _CE1 /V _A )Is ₁ } ...(6) V _BE2 = V _Tlo {Iouf/(1+V _CE2 /V _A )Is ₂ } ...(7), I _B = V _CC −R ₁₃ Iout／R ₁₃ + R ₁₄ ...(8) is obtained, and substituting equations (6), (7), and (8) into equation (5) Then, V _Tlo = {Iref/(1+V _CE1 /V _A )Is ₁ }+R ₁₁ Iref=
V _Tlo {Iout/(1+V _CE2 /V _A )Is ₂ } R ₁₂ Iout+R ₁₃ /R ₁₃ +R ₁₄ (V _CC +R ₁₄ Iout) ...(9).

Using equation (9), the output current when the power supply changes
When Iout was determined, as shown in the current characteristic of 4A shown in FIG. 4, there was almost no change, and it was confirmed that the output current value was highly accurate. Also, output current characteristic 4C is that of a conventional current mirror circuit, but when the power supply voltage changes from 1.5 [V] to 6.0 [V],
There was a difference of 10 [μA].

The present invention is not limited to the above embodiment, but as shown in FIG. 2, diodes D1 and D2 are connected in series with the resistor _R14 to set a threshold value, and the compensation circuit is An operation start voltage point may be determined. In the previous embodiment, the resistor R ₁₄ has a high resistance, and when integrated, it may be difficult to improve the accuracy of the resistance value. Further, since a current always flows through the resistor R ₁₄ for any value of the power supply voltage V _CC , even if current fluctuations due to the Early effect are canceled, the output current Iout is always smaller than the constant current Iref. In the circuit shown in Figure 2,
Since two diodes D1 and D2 are used, V _CC
When is 1.5 [V] or more, current I _B flows and the output current
Iout is always maintained at the same value as when V _CC =1.5 [V]. Since the other parts have the same configuration as the embodiment shown in FIG. 1, they are given the same reference numerals as in FIG. 1 and their explanation will be omitted.

Next, the embodiment shown in FIG. 2 will be further explained. When V _CC =1.5 [V], the potential at the connection point between diode D1 and resistor _R14 is also approximately 1.5 [V], so
Diodes D1 and D2 are off, and current I _B does not flow. However, when V _CC becomes 1.5 [V] or more,
Diodes D1 and D2 are turned on and current I _B flows. This increases the output current
For Iout, the current I _B flows, so the resistance R ₁₃
The voltage drop across transistor Q12 increases and V _BE of transistor Q12 increases.
As a result, the increase in the output current Iout is suppressed. That is, in the case of this circuit, the output current Iout is maintained at the value when V _CC =1.5 [V] even if V _CC becomes 1.5 [V] or more.

Now, since the on-voltages of diodes D1 and D2 are equal, let this be V _D , I _B = V _CC -2V _D -R ₁₃ Iout/R ₁₃ + R ₁₄ ...(10) This can be substituted for equation (8). Substituting into equation (5), V _Tlo {Iref/(1+V _CE1 /V _A )Is ₁ }+R ₁₁ Iref=V _T
lo {Iout／(1+V _CE2 /V _A )Is ₂ } +R ₁₃ Iout＋R ₁₃ /R ₁₃ +R ₁₄ (V _CC +R ₁₄ Iout−2V _D )
...(11) becomes.

Iout cannot be determined directly from equations (9) and (11). Therefore, R ₁₁ = R ₁₂ = 300 [Ω], R ₁₃ = R ₁₄
= 150 [Ω], Iref = 100 [μA], Is ₁ = Is ₂ = 4.2×10 ^-1
6
[A], V _T =26 [mV], V _A =20 [V], V _CE =0.7
[V], V _D = 0.75 [V] and constants are determined. Also, _R14
is 129 obtained from equation (11) with Iout = 100 [μA]
[KΩ], load 12 is assumed to be a diode whose potential is determined from the ground potential side, and resistors (R ₁₂ and R ₁₃ )
Assuming that the voltage drop at is sufficiently small compared to V _CE2 , substitute V _CE2 =V _CC -0.7 [V] into equations (9) and (11).

From the above, when Iout is determined by extrapolation using a computer, as shown in output current characteristic 4B in Figure 4,
It has stable characteristics that are not affected by fluctuations in power supply voltage.

As mentioned above, up to V _CC =6 [V], Iout97.5 [μA] in the first embodiment, and Iout97.5 [μA] in the second embodiment.
The expected value of Iout100 [μA] was obtained.

Furthermore, in the case of the above embodiment, when the ambient temperature changes, for example at low temperatures, the transistor β
(forward current amplification factor) decreases and the output current Iout decreases, but as the diode on-voltage increases, I _B also decreases and acts in the direction of increasing Iout, so the change due to temperature change is eventually canceled out. Ru. When the temperature is high, β increases and the on-state voltage of the diode decreases, contrary to when the temperature is low, so that the output current is also insensitive to temperature changes. Therefore, temperature compensation can also be obtained at the same time.

FIG. 3 shows yet another embodiment of the invention,
This is an example in which a diode D3 is further connected in series with the diodes D1 and D2. In this case, the diodes D1 to D3 turn on around V _CC =3.0 [V], so in this circuit, it is possible to suppress fluctuations in the output current Iout due to the Early effect when V _CC =3.0 [V] or higher. can. Further, in the present invention, compensation circuits having different threshold values may be connected in parallel and provided between the connection point of the resistors R ₁₂ and R ₁₃ and the ground potential.

In the above explanation, the current supplied to the current input node, which is the collector of transistor Q11, was explained as being from a constant current source, but this is for convenience of explanation, and it goes without saying that it may be an operating current. . Furthermore, a similar effect can be obtained with a circuit using an NPN transistor.

〔Effect of the invention〕

As explained above, the present invention has a current mirror circuit that eliminates the influence of the Early effect of the current mirror transistor on the output current and makes the output current stable and accurate over a wide range of power supply voltages with a simple configuration. can be provided.

[Brief explanation of the drawing]

FIG. 1 is a circuit diagram showing one embodiment of the present invention, FIGS. 2 and 3 are circuit diagrams showing other embodiments of the invention, and FIG. 4 is a circuit diagram showing a circuit according to the present invention and a conventional circuit. Characteristic diagram showing a comparison of output current characteristics,
FIG. 5 is a circuit diagram showing a conventional current mirror circuit. Q11, Q12...transistor, _R11 to _R14
...Resistor, D1, D2, D3...Diode.

Claims (1)

[Claims] 1. The bases of the first and second transistors and the collector of the first transistor are connected to a current input terminal, and the emitter of the first transistor is connected to the first transistor through a first resistor. to the constant voltage source, and the second
In a current mirror circuit in which a collector of a transistor is connected to a second constant voltage source via a load, an emitter of the second transistor is connected to the first constant voltage source.
is connected to a constant voltage source via second and third resistors, and the first constant voltage source fluctuates between the connection point of the second and third resistors and the second constant voltage source. A current mirror circuit characterized in that it forms a current path for suppressing changes in the collector-emitter potential of the second transistor. 2. Claim 1, wherein the current path connected between the connection point of the second and third resistors and the second constant voltage source has a fourth resistor connected in series. The current mirror circuit described. 3. A claim characterized in that the current path connected between the connection point of the second and third resistors and the second constant voltage source is a series circuit of a fourth resistor and a plurality of diodes. The current mirror circuit according to item 1.