JPH0131807B2 - - Google Patents

Description

Detailed Description of the Invention The present invention receives a plurality of orthogonally multiplexed amplitude modulation (hereinafter abbreviated as QAM) signals via a transmission line, samples them at a predetermined sampling clock, and converts them from analog to digital. (hereinafter abbreviated as AD) conversion,
Wave operations and discrete Fourier transform operations (below
This invention relates to a carrier timing combination control method in a receiving system for orthogonal multiplexed signals that demodulates a plurality of original baseband signals using DFT (abbreviated as DFT) or the like.

The orthogonal multiplex transmission method, in which multiple QAM signals are orthogonally multiplexed and transmitted, and each baseband signal is demodulated on the receiving side, enables highly efficient data transmission, and if DFT is introduced into the digital signal processing process, transmission and reception will be possible. It is already known that the device can be significantly simplified. (For example, as a reference, the publication "IEEE TRANSACTIONS ON
COMMUNICATIONS, Vol.COM−29, No.7”
The paper “AN” on pages 982-989 of
ORTHOGONALLY MULTIPLEXED QAM
SYSTEM USING THE DISCRETE
FOURIER TRANSFORM” and “TRANSMITTER AND RECEIVER FOR” described in US Patent No. 4300229, registered in 1981 as Document 2
AN ORTHOGONALLT MULTIPLEXED
QAM SIGNAL OF A SAMPLING RATE
N TIMES THAT OF PAM SIGNALS,
COMPRISING, AN TRANSFORM
(Please refer to “PROCESSOR”).

Additionally, a two-dimensional automatic equalizer for demodulated complex baseband signals has already been proposed, and if this two-dimensional automatic equalizer is used, it is possible to completely equalize the line distortion experienced in the transmission path. Furthermore, it is known that static carrier phase deviations and timing phase deviations are also suppressed at the same time (for example, in the publication "IEEE TRANSACTIONS ON
COMMUNICATIONS, Vol.COM−28, No.1”
The paper “AN ANALYSIS” on pages 73-83 of
OF AUTOMATIC EQUALIZERS FOR
ORTHOGONALLY MULTIPLEXED QAM
SYSTEMS).

However, the automatic equalizer alone cannot follow so-called dynamic carrier phase fluctuations and timing phase fluctuations with sufficient accuracy. As a method for suppressing such dynamic phase fluctuations, a method is already known in which a pilot is sent along with the transmission signal, and the receiving side detects the phase fluctuation of this pilot and applies appropriate compensation phase rotation to the complex baseband signal. (For example, please refer to Patent Application No. 4382 filed in 1982 as Document 4).

However, the above method requires an extra band for pilot transmission, which may be a drawback, especially when high transmission efficiency is important.

An object of the present invention is to provide a carrier timing coupling control system that can suppress dynamic phase fluctuations without using an extra pilot, based on the above-mentioned viewpoints.

That is, the present invention is characterized in that all orthogonally multiplexed QAM signals are staggered QAM (QAM with a delay difference of exactly half the symbol period between in-phase and orthogonal signals), and that the symbol timings of all QAM signals are synchronized. By focusing on these points and configuring a maximum likelihood receiving system that skillfully combines these properties, we provide a carrier timing coupling control system that can suppress dynamic phase fluctuations with high accuracy.

Hereinafter, the principle of the present invention and its configuration will be explained using the drawings.

FIG. 1 is a block diagram showing a general process of generating a two-dimensional modulation signal, in which 1 and 2 are input terminals into which in-phase data and quadrature data are input, respectively;
are an in-phase side baseband shaping filter and an orthogonal side baseband shaping filter, respectively, 5 and 6 represent multipliers, and 7 represents an adder. 8 is an output end, and a two-dimensional modulated signal is sent out to the transmission path via this output end. In FIG. 1, in-phase data {a _k } and orthogonal data {b _k } are input to input terminals 1 and 2.
The clock period of is T seconds, and the unit response of the in-phase baseband shaping filter shown by 3 is p(t),
Let q(t) be the unit response of the orthogonal side baseband shaping filter shown by 4. Further, if the modulation carrier angular frequency is ω _c (radian/second), the two-dimensional modulation signal s(t) obtained at the output end 8 is expressed as follows.

s(t)=〓 ^k {a _k p(t-kT) cosω _c t +b _k q(t-kT) sin-ω _c t} Here, regarding the reception operation for the above two-dimensional modulated signal s(t) I'll think about it. For the sake of simplicity, it is assumed that line distortion in the transmission path can be ignored, and that only transmission delay t' and carrier offset Δω are suffered. At this time, the received signal r(t) is s
(t) is expressed as the sum of a signal ro(t) obtained by receiving a transmission delay t' and a carrier offset Δω, and a white noise n(t). Here, ro(t) is ω′ _c
As =ω _c +△ω, ro(t)= 〓 ^k {a _k p(t−t′−kT) cosω _c ′(t−t′) +b _k q(t−t′−kT) sinω _c ′ (t-t')}.

If t'=mT+τ (where |τ|T/2) Θ=Δωt−ω' _c t', ro(t) can be expressed as follows.

ro(t)= 〓 ^k {a _k p(t-τ-kT) cos(ω _c t+Θ) +b _k p(t-τ-kT) cos(ω _c t+Θ} In other words, ro(t) is This means that the timing shift τ and carrier phase shift Θ are unknown quantities.The actual received signal r(t) is ro
(t) is buried in noise n(t), that is,
r(t)=ro(t)+n(t), but on the receiving side, from this signal r(t), the original data sequence {a _k }{b _k }
must be restored. When performing this restoration, it is first necessary to compensate for the timing deviation τ and the carrier phase deviation Θ.
Maximum likelihood estimation method is known. The basic idea of maximum likelihood estimation method is shown below.

Now, let the estimated sides of {a _k }, {b _k }, τ, and Θ be {a^ _k },
Let {b^ _k }, τ^, Θ^. Using these estimates,
Set the next reference signal u(t).

u(t)= 〓 ^k {a^ _k p(t-τ-kT) cos(ω _c t+Θ^) +b^ _k q(t-τ^-kT) sin(ω _c t+Θ^)} Maximum likelihood estimation From our point of view, the most probable estimate is the one that maximizes the a posteriori probability of obtaining the received signal r(t) when it is assumed that u(t) is sent on the transmitting side. In this estimation process, the method in which estimation is performed on the assumption that the data {a _k }, _{ b _k } are accurately known is called the decision feedback type (hereinafter abbreviated as DA type). , {b _k }, a method that performs estimation on the premise that only its statistical properties are known is called a decision-free feedback method (hereinafter abbreviated as NDA method). Generally τ, Θ
The DA type is better in steady-state performance of the control system against small fluctuations in , but when the fluctuations in τ and Θ are large, errors tend to occur in the data judgment results, and as a result, the performance of the control system deteriorates significantly. In addition, since correct judgment results are not obtained during the initial pull-in of the control system, the pull-in characteristics deteriorate in the DA type. That is, if high tracking ability for τ and Θ is expected, it is desirable to use the NDA type maximum likelihood estimation method.

The NDA type likelihood function L(τ^, Θ^) for the two-dimensional modulated signal is defined by the following equation, where n(t) is white Gaussian noise.

However, σ ² is the variance of n(t), and E{・} is
It represents an averaging operation on {a _k }{b _k }.

L(τ^, Θ^) is a probability density function, always takes a positive value, and has a unique maximum value at the optimal point of τ^, Θ^, so L(τ^, Θ^) The logarithm, ie, the log-likelihood function Ω(τ^, Θ^), also has a unique maximum value. Therefore, the control system for τ^ and Θ^ is Ω
It turns out that it is sufficient to control τ^ and Θ^ in order to maximize (τ^, Θ^). Here, if {a _k } and {b _k } are regarded as a type of noise having a Gaussian distribution, Ω(τ^, Θ^) is expressed by the following equation, as is well known.

Ω (τ^, Θ^) = constant × 〓 ^m v ² ₁ (mT + τ^) ・ 〓 ^m v ² ₂ (nT + τ^) However, v ₁ (t) = ∫ _T0 w ₁ (s)・p(s- t)ds v ₂ (t)=∫ _T0 w ₁ (s)q(s-t)ds w ₁ (t)=r(t)・cos(ω _c t+Θ^) w ₂ (t)=r(t ) × sin (ω _c t + Θ^) In other words, v ₁ (t) is the in-phase side demodulated signal w ₁ (t),
v ₂ (t) is an output obtained by passing the orthogonal side demodulated signal w ₂ (t) through a matching filter for the orthogonal side shaping filter.

Control over τ^ and Θ^ is performed using the following algorithm. That is, to obtain the next estimated values τ^ _{k +1} and Θ^ _k+1 from the current estimated values τ^ _k and Θ^ k, set K〓 and K〓 as loop gains, shall be. Here, when calculating ∂Ω/∂τ^ and ∂Ω/∂Θ^, ∂Ω/∂τ constant × 〓 ^m {v ₁ (mT+τ^)v〓 (mT+τ^) +v ₂ (mT+τ^)v〓 ₂ (mT+τ^)} ∂Ω/∂Θ^ constant× 〓 ^m {−v ₁ (mT+τ^) +y ₂ (mT+τ^)+v ₂ (mT+τ^)y ₁ (mT+τ^)} {carry, y ₁ (t )=∫ _T0 w ₁ (s)q(st)ds y ₂ (t)=∫ _T0 w ₂ (s)p(st)ds}.

In a normal QAM signal, p(t)=q(t), so y ₁ (t)=v ₁ (t), y ₂ (t)=v ₂ (t),
∂Ω/∂Θ^ becomes equal to 0.

That is, as is well known, NDA type carrier phase control cannot be applied to normal QAM signals. In contrast, the so-called staga
In QAM, q(t)=p(t-t ₀ ) (however,
t ₀ is the time difference between in-phase and quadrature), and NDA type phase control can be applied to both τ^ and Θ^, and especially when t ₀ = T/2, the control ability is maximum. becomes.

FIG. 2 is a block diagram showing the configuration of a timing/carrier combination control system for a staggered QAM signal with t ₀ =T/2. That is, in FIG. 2, the received signal input to the input terminal 11 is transmitted to the multiplier 12,
At 14, the output signal is inversely modulated by the output of the voltage controlled oscillator 17 and an in-phase carrier and a quadrature carrier obtained by phase-shifting the output by 90 degrees.

The in-phase side output and the quadrature side output are further supplied to matching filters 15 and 18, respectively. The outputs of the matching filters 15 and 18 are correlated with each other in a multiplier 19, and the oscillation phase of the voltage controlled oscillator 17 is controlled by this correlation value. Regarding timing control, for the voltage controlled oscillator 25 that determines the timing phase, the output of the matched filter 15 is delayed by T/2 seconds and its differential signal correlates with the output signal of the matched filter 18 and its differential signal. This is executed by adding the signal and the correlation in an adder 24 and supplying the obtained control signal. The samplers 26 and 27 sample the in-phase signal and the quadrature signal, respectively, using the timing clocks obtained in this way, and their outputs are sent to a subsequent automatic equalizer, etc. via output terminals 28 and 29. Supplied.

FIG. 3 is a block diagram showing a configuration for realizing the control system shown in FIG. 2 entirely through digital processing. The signal is sampled by the sampling clock obtained as the output of the signal, and is supplied to multipliers 33 and 34. The multipliers 33 and 34 are
The input signal is multiplied by a cosine sample value series and a sine sample value series obtained by driving the read-only memory 35 with the sampling clock to obtain an in-phase signal and a quadrature signal. The in-phase signal and quadrature signal thus obtained are sent to the phase rotation circuit 3.
8, the amount of phase rotation is controlled by the in-phase-quadrature correlation value obtained as the output of the multiplier 40. The in-phase signal among the outputs of the phase rotation circuit 38 is delayed by T/2 seconds in the delay circuit 39 and then reaches the output terminal 47 . On the other hand, the orthogonal signal reaches the output terminal 48 as it is. A control signal for the voltage controlled oscillator 46 is supplied from an adder 45, and this adder 45 calculates the correlation value between the in-phase signal delayed by T/2 seconds and its differential signal, and the correlation between the quadrature signal and its differential signal. Performs addition with the value.

The carrier timing coupling control method according to the present invention is applied to an orthogonal QAM transmission system, but this orthogonal QAM transmission system is basically the staggered
This is a method in which multiple QAM signals are frequency-multiplexed and transmitted while allowing spectrum overlap.

FIG. 4 shows the basic configuration of the orthogonal QAM transmission system, and FIG. 5 shows an example of the spectrum of the orthogonal QAM signal sent to the transmission path.

In FIG. 4, terminals 51 and 52 have first channel in-phase data {a ⁽¹⁾ _k } and quadrature data {b ⁽¹⁾ _k }, respectively.
is input with a clock cycle of T seconds. The in-phase data {a ⁽¹⁾ _k } is delayed by T/2 seconds in the delay circuit 59 and then passes through the shaping filter 63 to the multiplier 71 . On the other hand, the orthogonal data {b ⁽¹⁾ _k } passes through the shaping filter 64 and reaches the multiplier 72 . Molded filter 6
3 and 64 have the same frequency transfer characteristic G(ω), and this G(ω) is set to a so-called root Nyquist shape. If the impulse response of G(ω) is g(t), the output signal u ⁽¹⁾ _R (t) of the shaping filter 63 and the output signal u ⁽¹⁾ _I (t) of the shaping filter 64 are respectively u ^{( 1)} _R (t) = 〓 ^k a ⁽¹⁾ _k・g (t-T/2-kT) ...(1) u ⁽¹⁾ _I (t)=Σb ⁽¹⁾ _k・g (t-kT ) ...(2) The multiplier 71, the multiplier 72, and the adder 79 constitute a quadrature amplitude modulator with a carrier angular frequency ω ₁ , and the output of the adder 79 has the first QAM
The signal q ⁽¹⁾ (t) is q ⁽¹⁾ (t) = u ⁽¹⁾ _R (t) cosω ₁ t + u ( ^{1 )} _I (t) sinω ₁ t = _Re )＋ju ⁽¹⁾ _I (t)〕e ^-j 〓 ^1t }
...(3) It is expressed as. However, Re{·} represents an operation that takes only the real part. Adder 8 in exactly the same way
The outputs of 0, 81, and 82 each have the second QAM signal q ⁽²⁾
(t), 3rd QAM signal q ⁽³⁾ (t), 4th QAM signal q ⁽⁴⁾
(t) is obtained. These signals are expressed by the following equations.

q ⁽²⁾ (t) = Re {[u ⁽²⁾ _R (t) + ju ⁽²⁾ _I (t)] e ^-j
〓 ^2t
...(4) q ⁽³⁾ (t) = Re {[u ⁽³⁾ _R (t) + ju ⁽³⁾ _I (t)] e ^-j
〓 ^3t
...(5) q ⁽⁴⁾ (t) = Re {[u ⁽⁴⁾ _R (t) + ju ⁽⁴⁾ _I (t)] e ^-j
〓 ^4t
……(6) However, u ⁽²⁾ _R (t)= 〓 ^k a ⁽²⁾ _k g(t−kT) ……(7) u ⁽²⁾ _I (t)= 〓 ^k b ⁽²⁾ _k g(t-T/2-kT) ……(8) u ⁽³⁾ _R (t)= 〓 ^k a ⁽³⁾ _k g(t-T/2-kT) ……(9) u ⁽³⁾ _I (t)= 〓 ^k b ⁽³⁾ _k g (t-kT) ……(10) u ⁽⁴⁾ _R (t)= 〓 ^k a ⁽⁴⁾ _k g (t-kT) ……(11) u ⁽⁴⁾ _I (t) = 〓 ^k b ⁽⁴⁾ _k g (t-T/2-kT) ...(12), and {a ⁽²⁾ _k } and {b ⁽²⁾ _k } are , terminals 53 and 54, respectively.
{a ⁽³⁾ _k } and {b ⁽³⁾ _k } represent the second channel in-phase data and quadrature data input to the terminals 55 and 56, respectively. ⁽⁴⁾ _k } and {b ⁽⁴⁾ _k } respectively indicate fourth channel in-phase data and quadrature data input to terminals 57 and 58, respectively. Here, the relationship ω _k −ω _k-1 = 2π/T is established between each carrier angular frequency ω ₁ , ω ₂ , ω ₃ , ω ₄ , and furthermore, the T/2 second delay circuit 5
9, 60, 61, and 62 are inserted alternately on the in-phase side, the orthogonal side, the same side, and the orthogonal side, respectively. As a result, the first to fourth QAM signals are orthogonally multiplexed on the frequency axis. Letting this orthogonal multiplexed signal be s(t), s(t) is obtained as the output of the adder circuit 83, and in the example of 4-channel multiplexing as shown in the figure, s(t)= ₄ 〓 ⁿ⁼¹ g ^{(n )} (t) ...(13) Further, the spectrum of the orthogonal multiplexed signal s(t) is as shown in FIG. That is, in FIG. 5, 110, 111, 112, 113
represent the spectra of the first, second, third, and fourth QAM signals, respectively, and spectrum overlap always occurs between adjacent channels. A feature of the orthogonal multiplex transmission system is that even if such adjacent spectrums overlap, the original baseband data can be restored without error on the receiving side, and thus a highly efficient data transmission system can be realized.

Now, in FIG. 4, on the receiving side, the inverse calculation of the calculation performed on the transmitting side is performed. For example, multipliers 84, 85 and shaping filters 92, 93
is the in-phase baseband signal v ⁽¹⁾ _R from the first QAM signal
(t), is used to restore the orthogonal baseband signal v ⁽¹⁾ _I (t). Here, the molded filters 92, 9
The frequency transfer characteristic of No. 3 is based on the shaping filter 6 on the reverse signal side.
It has exactly the same root Nyquist characteristics as 3, 64, etc. Therefore, v ⁽¹⁾ _R (t) and v ⁽¹⁾ _I (t) are each given by the following equations when there is no noise in the transmission path. (However, ω ₁ is assumed to be larger than 2π/T.) v ⁽¹⁾ _R (t) = (u ⁽¹⁾ _R (τ) cosΘ ₁ −u ⁽¹⁾ _I (t) sin
Θ ₁ ) * g (t) + (u ⁽²⁾ _R (t) cos (Θ ₁ −2π/Tt
) −u ⁽²⁾ _I (t) sin(Θ ₁ −2π/Tt) * g(t
)...(14) v ⁽¹⁾ _I (t) = (u ⁽¹⁾ _R (t) sinΘ ₁ +u ⁽¹⁾ _I (t) cos
Θ ₁ ) * g (t) + (u ⁽²⁾ _R (τ) sin (Θ ₁ −2π/Tt
) +u ⁽²⁾ _I (t) cos (Θ ₁ −2π/Tt)) * g (
t)...(15) Here, Θ ₁ is the phase shift of the first carrier on the receiving side, and * represents the convolution operation.
Substituting equations (1), (2), (7), and (8) into equations (14) and (15), v ⁽¹⁾ _R (t)= 〓 ^k a ⁽¹⁾ _k h ₀ (t− T/2−kT) cosΘ ₁ − 〓 ^k b ⁽¹⁾ _k h ₀ (t−kT) sinΘ ₁ + 〓 ^k a ⁽²⁾ _k (h _c (t−kT) cosΘ ₁ h _s (t−kT) sinΘ ₁ )+ 〓 〓 ^k b ⁽²⁾ _k (hc (t-T/2-kT) sinΘ ₁ -h _s (t-T/2
−kT) cosΘ ₁ …(16) v ⁽¹⁾ _I (t)= 〓 ^k a ⁽¹⁾ _k h ₀ (t−T/2−kT) sinΘ ₁ − 〓 ^k b ⁽¹⁾ _k h ₀ ( t-kT) cosΘ ₁ + 〓 ^k a ⁽²⁾ _k (h _c (t-kT) sinΘ ₁ -h _s (t-kT) cosΘ ₁ )- 〓 〓 ^k b ⁽²⁾ _k (h _c (t- T/2-kT) _cosΘ1 - _hs (t-T/2
−kT) sinΘ ₁ )...(17) is obtained. However, h ₀ (t) and h _s (t) are functions defined below.

h _p (t)=g(t)*g(t) h _c (t)=g(t) cos2π/Tt*g(t) h _s (t)=g(t) sin2π/Tt*g(t ) Since g(t) is the impulse response of a filter having root Nyquist characteristics, for an integer value m, h _p (mT) = 1 when m = 0, 0 when m≠0, and for any arbitrary It is already known that h _s (mT)=0 and h _c (mT+T/2)=0 hold for an integer m. Therefore,
In equations (16) and (17), Θ ₁ =0 and v ⁽¹⁾ _R (t)
,
If v ⁽²⁾ _R (t) is sampled at the timings of t=mT+T/2 and t=mT, then v ⁽¹⁾ _R (mT+T/2)=a ⁽¹⁾ _n v ⁽¹⁾ _I (mT)=b ⁽¹⁾ _n , and it can be seen that the in-phase data and orthogonal data of the first channel can be correctly restored. 2nd to 4th
Channel in-phase data and quadrature data can also be correctly restored in exactly the same manner. However,
Generally, on the receiving side, there is a carrier phase shift and a timing shift, so it is necessary to control these shifts. In this orthogonal QAM transmission system, the present invention focuses on the fact that each channel is a staggered QAM and that all channels operate with a synchronous clock, and precisely controls the carrier phase shift and timing shift described above. A carrier timing coupling control method is provided.

Now, in FIG. 4, the true timing shift on the receiving side is τ ^* , and the true carrier phase shifts for the first to fourth QAM channels are respectively Θ ^* ₁ , Θ ^* ₂ ,
Let Θ ^* ₃ and Θ ^* ₄ . The transmission complex envelope and reception complex envelope of the k-th channel are β _k (t) and γ _{k ,} respectively.
(t), then γ _k (t)=β _k (t+τ ^* )e ^j 〓 ^*k +β _k-1 (t+τ ^*
)e ^j 〓 ^0(t+ 〓 ^*)+j 〓 ^k* +β _k+1 (t+τ ^* )e ^j 〓 ^0(t+ 〓 ^*)
+j 〓 ^*k +n _k (t) k=1, 2, 3, 4 (19). However, β ₀ (t)=β ₅ (t)=0.

Also, ω ₀ =2π/T, and n _k (t) is the complex baseband noise of the k-th channel. Since the second and third terms in equation (19) are inter-channel interference independent of the desired signal in the first term, they can be approximately considered Gaussian noise, and γ _k (t) can be expressed as follows: Write down.

γ _k (t)=β _k (t+τ ^* )e ^j 〓 ^*k +N _k (t)……(20
) However, N _k (t) is the sum of all terms other than the first term in equation (19). If the likelihood function in this system is L (τ, Θ ₁ , Θ ₂ , Θ ₃ , Θ ₄ ), then L
(τ, Θ ₁ , Θ ₂ , Θ ₃ , Θ ₄ ) = {transmission complex envelope set β ₁ (t+τ)e ^j 〓 ¹ ,
When β ₂ (t+τ)e ^j 〓 ³ , β ₄ (t+τ)e ^j 〓 ⁴ are sent, the received complex envelope set γ ₁ (t), γ ₂
(t), γ ₃ (t), γ ₄ (t)}. Here, the noise N ₁ (t), N ₂ (t), N ₃
(t) and N ₄ (t) can be regarded as mutually uncorrelated Gaussian noise, so L(τ, Θ ₁ , Θ ₂ , Θ ₃ , Θ ₄ ) = ₄ 〓 ^k=1 {β _k (t+τ)e ^j 〓 Posterior probability of receiving γ _k (t) when sending ^k }
The log likelihood function obtained by taking the logarithm is Ω
(τ, Θ ₁ , Θ ₂ , Θ ₃ , Θ ₄ ) is Ω(τ, Θ ₁ , Θ ₂ ,
Θ ₃ , Θ ₄ )=Σ log {β _k (t+τ) e ^j 〓 posterior probability of receiving γ _k (t) when sending ^k }.

If Ω (τ, Θ ₁ , Θ ₂ , Θ ₃ , Θ ₄ ) is averaged over the data, then = ₄ 〓 ^k=1 _k (τ, Θ _k ) ...(21) However, _k is When k is an odd number, (τ, Θ _k ) = constant × Σv ² ₁ , _k (mT + T / 2 + τ) Σv ² ₂ , _k (nT + τ) ... (22) When k is an even number, (τ, Θ _k ) = constant × 〓 ^m v ² ₁ , _k (mT+τ)〓 _o v ² ₂ , _k (nT+T/2+τ)...(23) where v ₁ , _k (t) , v ₂ , _k (t) is
respectively v ₁ , _k (t)∫Re{γ _k (s)e ^-j 〓 ^k }g(s-t)ds v ₂ , _k (t)∫lm{γ _k (s)e ^-j 〓 ^k } g(s-t)ds, and signals corresponding to this are obtained at output terminals 101-108 in FIG. That is, the output end 10
1,102, the in-phase component v _1,1 (t) and quadrature component v _2,1 (t) of the first channel demodulated baseband signal are obtained, and the output terminals 103, 104 have v _1,2 (t ), v _2,2
(t), but the output terminals 105 and 106 have v _1,3 (t),
v _2,3 , output terminals 107, 108 have v _1,4 (t), v _2,4
(t) are obtained respectively.

From the above equation (21), the following control method for the timing deviation τ and the carrier phase deviations Θ ₁ , Θ ₂ , Θ ₃ , and Θ ₄ can be obtained.

That is, the current estimates τ, Θ ⁽ⁱ⁾ ₁ , Θ ⁽ⁱ⁾ ₂ , Θ ⁽ⁱ⁾ ₃ , Θ
⁽ⁱ⁾ Estimates from ₄ to τ ⁽ⁱ⁺¹⁾ , Θ ₁ ⁽ⁱ⁺¹⁾ , Θ ₂ ⁽ⁱ⁺¹⁾ , Θ ₃ ⁽ⁱ⁺¹⁾
,
To obtain Θ ₄ ⁽ⁱ⁺¹⁾ , let K〓, K〓 be the loop gains, And it is sufficient. In the orthogonal QAM transmission system, each channel is staggered QAM, so as mentioned above, NDA control is possible with respect to timing and carrier phase. Furthermore, regarding timing, since all channels are in a synchronous relationship, it is understood that common control can be applied by adding up all the timing control signals obtained from each channel.

FIG. 6 is a block diagram showing an embodiment of the present invention based on the above principle. That is, Fig. 6 is the 4th
The figure shows an example in which the carrier timing coupling control method according to the present invention is applied to a system in which all of the receiving section processing has been digitized, and 121, 122, and 123 are demultiplexing circuits, polyphase circuits, and inverse offset Fourier transform circuits, respectively. The four outputs of the inverse offset Fourier transform circuit 123 are T/2 seconds sampling of the first channel complex baseband signal to the fourth channel complex base hand signal obtained at the output terminals 101 to 108 in FIG. It has become a series. For details of the demultiplexing circuit, the polyphase circuit, and the inverse offset Fourier transform circuit, please refer to Patent Application No. 28740 filed in 1987 as Document 5.

In FIG. 6, the sampling circuit 138 is connected to the input terminal 1.
The received signal obtained at 20 is sampled by a sampling clock having a frequency _fs . Here, f _s and the baud rate 1/T of each channel usually maintain a simple integer ratio relationship, and therefore, the so-called timing time is determined based on this sampling clock. Therefore, in this system, the timing control signal is
It will control the VCO 137 which generates the sampling clock. 124 to 127 are phase rotation circuits for the first to fourth channels, and 128 to 131 are phase difference detection circuits for each channel. The phase rotation circuits 124 to 127 are configured as shown in FIG. 9, for example. That is, in FIG. 9, 160 and 161 are input terminals into which the in-phase and quadrature components of the demodulated baseband signal are input, respectively, and 162, 163, 164,
165 is a multiplier, 166 is an adder,
167 is a subtracter, 168 is a ROM in which sine wave series and cosine wave series are written, 169 is a 1 sample delay circuit, 170 is an adder, and 172 and 173 are bases subjected to phase rotation. This is the output terminal where the in-phase and quadrature components of the band signal are output. Now, if a phase control signal φ is input to the control terminal 171, the ROM 168
Addresses are accessed using binary codes corresponding to . In response to the input of φ, the ROM168 has cosφ
Internal data is written to output sinφ. Obtained as the first output of ROM168
cosφ is an in-phase signal at multipliers 162 and 165, respectively;
Multiplied by orthogonal signal. Similarly, the second output sinφ is multiplied by the in-phase signal and the quadrature signal by multipliers 163 and 164, respectively. Furthermore, the multiplier 162
The output of the adder 166 and the output of the multiplier 163 are
The sum signal is summed at the output terminal 172, and this sum signal is outputted to the output terminal 172 as a new in-phase signal. Multiplier 1
64 and the output of the multiplier 165 are the subtracter 16
7, the difference between them is taken, and this difference signal is outputted to the output terminal 173 as a new orthogonal signal. By this operation, the input complex signal undergoes a phase rotation of (-φ) and is output.

The phase control signal is output from the phase difference detection circuit 12 in FIG.
These phase difference detection circuits are configured as shown in FIG. 7, for example. That is, in FIG. 7, 14
0,141 are input terminals into which the in-phase and quadrature components of the complex baseband signal are input, and 142,143
is a T/2 second delay circuit, 144 and 145 are multipliers, 146 is a subtracter, and 147 is an output terminal from which a phase control signal is output. Input end 1
40 or 141 to which the in-phase component is input depends on how the T/2 second delay is applied on the transmitting side. That is, for channels such as the first and third channels in which the in-phase component is delayed by T/2 seconds on the transmitting side, the in-phase component is allocated to the input end 140 and the orthogonal component is allocated to the input end 141. For channels such as the second and fourth channels where the orthogonal portion is delayed by T/2 seconds on the transmitting side, the opposite allocation is performed. In this way, the difference component between the correlation values of the two inputs and the correlation values of the two T/2 seconds delayed inputs is outputted to the output terminal 147 as a phase control signal.

As described above, the carrier phase deviations Θ ₁ , Θ ₂ , Θ ₃ , Θ ₄
are phase rotation circuits 124, 125, 126, 1, respectively.
Corrected in step 27. This control loop only includes a delay of about 1 sample, so
Even if there is an offset in the carrier frequency,
The tracking characteristics for this are good.

Next, the timing deviation τ is 132,1 in FIG.
Timing deviation detection circuits 33, 134, and 135 detect each independently, and all of these detection outputs are added together in an adder circuit 136, and the VCO
137 will be controlled. Each timing deviation detection circuit is configured as shown in FIG. 8, for example.
That is, in FIG. 8, 150 and 151 are input terminals into which the in-phase and quadrature components of the phase-corrected complex baseband signal are input, and 152 and 153 are as follows.
It is a differential circuit realized as a discrete value system, and 15
4, 155 is a multiplier, 156 is an adder, 158 is a T/2 second delay circuit, and 157 is an output terminal. In the timing shift detection circuit for odd-numbered channels such as 132 and 134 in FIG. Orthogonal signals are input to the ends 151, respectively.

Conversely, in a timing shift detection circuit for even-numbered channels such as 133 and 135 in FIG. In-phase signals are input to input terminals 151, respectively.

In FIG. 8, multiplier 154 is connected to input terminal 150.
The multiplier 153 calculates the correlation between the signal input to the input terminal 151 and its differential signal and the signal obtained by delaying the signal input to the input terminal 151 by T/2 seconds. The correlation value of
After being added in step 6, the signals are outputted to a terminal 157 as a timing shift signal. This timing shift signal is common to all channels except for noise that mixes independently between each channel, so
A timing control signal with a high signal-to-noise ratio is obtained at the output of the adder circuit of FIG.

In one embodiment of the carrier timing coupling control method according to the present invention shown in FIG. 6, when the group delay distortion amplitude within the transmission band is sufficiently small compared to T seconds,
It can be applied as is. However, in a normal carrier band, extremely large group delay distortion may occur, especially near the band edges. For such lines, there is a sixth method that makes the timing control signal common to all channels.
The illustrated embodiment cannot be used. However, even for such lines, tracking of carrier timing can be made possible by sharing timing control signals among only a few channels whose spectrum is arranged near the center of the band.

FIG. 10 is a block diagram showing a general embodiment of the carrier timing coupling control method according to the present invention derived from the above viewpoint, and includes 181,
182, 183, and 184 correspond to the sampling circuit 138, demultiplexing circuit 121, polyphase circuit 122, and inverse offset Fourier transform circuit 123 in FIG. 6, respectively. Further, 185 to 188 are phase rotation circuits for the first to fourth channels corresponding to 124 to 127 in FIG. Furthermore,
189 and 190 are phase difference detection circuits corresponding to 129 and 130 in FIG. 6, and 191 and 192 are timing shift detection circuits corresponding to 133 and 134 in FIG. In a general embodiment of the present invention shown in FIG. 10, only the timing deviation signals obtained from the second channel and the third channel are shared in an adder circuit 193 to obtain a control signal for the VCO 194. . As a result, both the timing shift and the carrier shift of the second channel and the third channel are controlled. On the other hand, the first channel and the fourth channel are not equipped with a phase difference detection circuit or a timing shift detection circuit, however, the amount of time variation of the timing shift is common to all channels regardless of line distortion. This common deviation is compensated by a control loop made up of the VCO 194, sampling circuit 181, and the like. Further, phase rotation circuits 185 and 188 are provided in the first channel and the fourth channel,
Since each phase rotation amount is controlled by the phase difference detection circuit 189, the time variation of the carrier phase shift is also compensated for. That is, the addition circuit 19
Regarding the first and fourth channels that are not shared in the third channel, there is no temporal variation in timing/carrier phase in the output complex signal, and only a static phase shift remains. These static timing deviations and carrier phase deviations are
It is known that this can be absorbed by the two-dimensional automatic equalizer shown in .

As described above, according to the present invention, a carrier timing coupling control system with excellent tracking characteristics can be obtained.

It is clear that the embodiment shown in FIG. 6 can be regarded as a special case of the general embodiment shown in FIG.

[Brief explanation of drawings]

FIG. 1 is a block diagram showing a general process of generating a two-dimensional modulated signal, and 3 and 4 are an in-phase baseband shaping filter and an orthogonal baseband shaping filter, respectively. Figure 2 shows T/2 between in-phase and orthogonal
FIG. 2 is a block diagram showing the configuration of a carrier timing coupling control system for staggered QAM signals having a delay difference of seconds, in which 12, 14, 19, 21, and 22 are multipliers, 13 is a 90° phase shifter, 15, 1
8 is a matching filter, 16 is a T/2 second delay circuit, 1
7, 25 are VCOs, 20, 23 are differential circuits, 24
is an adder. Figure 3 shows the timing and timing of Figure 2.
32 is a block diagram showing a configuration for realizing a carrier coupling control system entirely through digital processing;
35 is a ROM, and 38 is a phase rotation circuit. FIG. 4 is a diagram showing the basic configuration of an orthogonal QAM transmission system, and FIG. 5 is a diagram showing an example of spectrum arrangement of orthogonal QAM signals sent to a transmission line in this transmission system. FIG. 6 is a block diagram showing a specific embodiment of the carrier timing coupling control system according to the present invention, and 121, 122, 123 are
Each includes a demultiplexing circuit, a polyphase circuit, an inverse offset Fourier transform circuit, 124 to 127 are phase rotation circuits, 128 to 131 are phase difference detection circuits,
132 to 135 are timing deviation detection circuits;
36 is an adder circuit, 137 is a VCO, 138
is the sampling circuit. FIG. 7 is a block diagram showing a specific example of the configuration of the phase difference detection circuit.
2,143 is T/2 second delay circuit, 144,145
is a multiplier, and 146 is a subtracter. FIG. 8 is a block diagram showing a specific example of the configuration of the timing deviation detection circuit, in which 152 and 153 are differentiating circuits;
54, 155 are multipliers, 156 are adders, 158
is a T/2 second delay circuit. FIG. 9 is a block diagram showing a specific configuration example of the phase rotation circuit, in which 162 to 165 are multipliers, 166 is an adder, 167 is a subtracter, 168 is a ROM, 169 is a one-sample delay circuit, 170 is an adder. FIG. 10 is a block diagram showing a general embodiment of the carrier timing coupling control system according to the present invention, in which 185 to 188 are phase rotation circuits;
189, 190 are phase difference detection circuits, 191, 192
193 is a timing shift detection circuit, 193 is an addition circuit,
194 is a VCO.

Claims (1)

[Claims] 1 The first to N-th orthogonal amplitude modulation signals receive orthogonally multiplexed signals in the frequency domain via a transmission line, are sampled by a sampling clock, and are used for analog-to-digital conversion, wave manipulation, and discrete In the receiving system of the orthogonal multiplexed signal which is demodulated into the first to Nth complex baseband signals by Fourier transform, a phase rotation operation is performed to independently give a phase rotation to the plurality of complex baseband signals, and a predetermined is obtained from the phase-rotated outputs of the k ₁ -km th complex baseband signals and their differential signals. The _phase of the sampling clock is controlled by the signal obtained by adding all m timing control signals, and
The amount of phase rotation for the kmth complex baseband signal is controlled by the correlation value between the real part and the imaginary part of each complex _baseband signal, and
A carrier timing coupling control method, wherein an amount of phase rotation for a complex baseband signal other than the km-th signal is controlled by at least one of the correlation values.