JPH0131808B2 - - Google Patents

Description

[Detailed description of the invention]

The present invention receives a plurality of orthogonally multiplexed amplitude modulation (hereinafter abbreviated as QAM) signals via a transmission path, samples them at a predetermined sampling clock,
Analog-to-digital (hereinafter abbreviated as AD) conversion, wave manipulation and discrete Fourier transform operation (hereinafter referred to as AD)
This invention relates to an offset suppressing carrier timing combination control method in a receiving system for orthogonal multiplexed signals that demodulates a plurality of original baseband signals using a method such as DFT (abbreviated as DFT). The orthogonal multiplex transmission method, in which multiple QAM signals are orthogonally multiplexed and transmitted, and each baseband signal is demodulated on the receiving side, enables highly efficient data transmission, and if DFT is introduced into the digital signal processing process, transmission and reception will be possible. It is already known that the device can be significantly simplified (for example, in the literature, ``IEEE TRANSACTIONS ON
COMMUNICATIONS, Vol.COM−29, No.7”
The article “AN
ORTHOGONALLY MULTIPLEXED QAM
SYSTEM USING THE DISCRETE
FOURIER TRANSFORM” and “TRANSMITTER AND RECEIVER FOR
AN ORTHOGONALLY MULTIPLEXED
QAM SIGNAL OF A SAMPLING RATE
N TIMES THAT OF PAM SIGNALS,
COMPRISNG, AN TRANSFORM
PROCESSOR). In addition, a two-dimensional automatic equalizer for demodulated complex baseband signals has already been proposed, and by using this two-dimensional automatic equalizer, it is possible to completely equalize the line distortion experienced in the transmission path. Furthermore, it is known that static carrier phase deviations and timing phase deviations are also suppressed at the same time (for example, in Document 3, 1981
Publication “IEEE TRANSACTIONS” published in January
ON COMMUNICATIONS, Vol.COM−28,
No. 1”, pages 73-83 of the paper “AN
ANALYSIS OF AUTOMATIC
EQUALIZERS FOR ORTHOGONALLY
(Please refer to "MULTIPLEXED QAM SYSTEMS"). However, the automatic equalizer described above cannot track so-called dynamic carrier phase fluctuations and timing phase fluctuations with sufficient precision. As a method, a method is already known in which a pilot is sent out along with the transmission signal, and the receiving side detects the phase fluctuation of this pilot and applies an appropriate compensation phase rotation to the complex baseband signal (for example, as described in Reference 4 in 1982). (Please refer to Patent Application No. 4382).However, the above method requires an extra band for pilot transmission, which may be a drawback, especially when high transmission efficiency is important. Based on the above-mentioned viewpoint, it is an object of the present invention to provide an offset suppressing carrier timing coupling control system capable of suppressing dynamic phase fluctuations without using an extra pilot. , the present invention is characterized in that all orthogonally multiplexed QAM signals are star QAM (QAM with a delay difference of half the symbol period between in-phase and orthogonal signals), and that the symbol timings of all QAM signals are synchronized. By focusing on these points and configuring a maximum likelihood receiving system that skillfully combines these properties, we provide an offset suppressing carrier-timing coupling control system that can suppress dynamic phase fluctuations with high accuracy. The principle of the present invention and its configuration will be explained below with reference to the drawings. Fig. 1 is a block diagram showing the general generation process of a two-dimensional modulation signal, and 1 and 2 are in-phase data and quadrature data, respectively. This is the input terminal for input, and 3, 4
are respective in-phase side baseband shaping filters and orthogonal side baseband shaping filters, 5 and 6 represent multipliers, and 7 represents an adder. 8 is an output end, and a two-dimensional modulated signal is sent out to the transmission path via this output end. In FIG. 1, in-phase data {a <sub>k</sub> } and orthogonal data {b <sub>k</sub> } are input to input terminals 1 and 2.
The clock period of is T seconds, and the unit response of the in-phase baseband shaping filter shown by 3 is p(t),
Let g(t) be the unit response of the orthogonal side baseband shaping filter shown by 4. Also, if the modulation carrier angular frequency is w <sub>c</sub> (radian/second), then the output terminal 8
The two-dimensional modulated signal s(t) obtained is expressed as follows. s(t) = 〓 <sup>k</sup> {a <sub>k</sub> p(t-kT) cosw <sub>c</sub> +b <sub>k</sub> g(t-kT) sinw <sub>c</sub> t} Now, let's consider the reception operation for the above secondary modulation signal s(t). . For simplicity, it is assumed that line distortion in the transmission path can be ignored and that only transmission delay t' and carrier offset Δw are suffered. At this time, the received signal r(t) is s(t)
is expressed as the sum of the signal r <sub>0</sub> (t) obtained by receiving the transmission delay t' and the carrier offset Δw and the white noise n(t). Here, r <sub>0</sub> (τ) is expressed as <sub>r 0 ( τ</sub> ) = Σ{a <sub>k</sub> p (t-t'-kT) cosw' <sub>c</sub> (t-t'
) +b <sub>k</sub> g (t-t'-kT) sinw' <sub>c</sub> (t-t')}. If t′=mT+τ (where |τ|T/2) θ=Δwt−w′ <sub>c</sub> t′, r <sub>0</sub> (τ) can be expressed as follows. r <sub>0</sub> (τ) = 〓 <sup>k</sup> {a <sub>k</sub> p(t-τ-kT) cos(w <sub>c</sub> t+θ) +b <sub>k</sub> g(t-τ-kT) sin(w <sub>c</sub> t+θ)} That is, r <sub>0</sub> (t) is The receiving side receives unknown amounts of timing deviation τ and carrier phase deviation θ. The actual received signal r(t) is r <sub>0</sub>
(t) is buried in noise n(t), that is, r(t)
= r <sub>0</sub> (t) + n(t), but the receiving side must restore the original data sequences {a <sub>k</sub> } and {b <sub>k</sub> } from this signal (r)(t). In this restoration, it is first necessary to compensate for the timing deviation τ and the carrier phase deviation θ, and the maximum likelihood estimation method is conventionally known as a method for this. The basic idea of maximum likelihood estimation method is shown below. Now, the estimated sides of {a <sub>k</sub> }, {b <sub>k</sub> }, τ, and θ are {a^ <sub>k</sub> },
Let {b^ <sub>k</sub> }, τ^, θ^. The next reference signal u(t) is set using these estimated values. u(t)= 〓 <sup>k</sup> {a^ <sub>k</sub> p(t−τ^−kT) cos(w <sub>c</sub> t+θ^) +b^ <sub>k</sub> g(t−τ^−kT) sin(w <sub>c</sub> t+θ^)} Maximum likelihood estimation From the standpoint of ``the most probable estimate is the one that maximizes the a posteriori probability of obtaining the received signal r(t) when it is assumed that u(t) is sent on the transmitting side''. In this estimation process, the method that performs estimation on the premise _that the data {a _k }, {b _k } are accurately known is called the decision feedback type (hereinafter abbreviated as DA type). The estimation method is based on the premise that {b _k } is known only from its statistical properties.
(abbreviated as NDA type). In general, the steady performance of a control system with respect to minute fluctuations in τ and θ is
The DA type is superior, but when the fluctuations in τ and θ are large, errors tend to occur in the data judgment results, which significantly degrades the performance of the control system. In addition, since correct judgment results are not obtained during the initial pull-in of the control system, the pull-in characteristics of the DA type deteriorate.
Particularly when the carrier offset is large, retraction is often impossible with the DA type. That is, τ,
If high tracking ability for θ is expected, it is desirable to use the NDA type maximum likelihood estimation method. The NDA type likelihood function L(τ^, θ^) for the two-dimensional modulated signal is defined by the following equation, where n(t) is white Gaussian noise. However, σ ² is the variance of n(t), and E{・} is
It represents an averaging operation on {a _k }{b _k }. L(τ^, θ^) is a probability density function and always takes a positive value, and has a unique maximum value at the optimal point of τ^, θ^, so the logarithm of L(τ^, θ^) The log-likelihood function Ω(τ^, θ^) also has a unique maximum value. Therefore, it can be seen that the control system for τ^ and θ^ should control τ^ and θ^ in order to maximize Ω(τ^, θ^). Here, if we consider {a _k } and {b _k } as a kind of noise with Gaussian distribution, then Ω
(τ^, θ^) is expressed by the following formula. Ω (τ^, θ^) = Constant × 〓 ^m v ² ₁ (mT + τ^)・Σv ² ₂ (nT + τ^) However, v ₁ (t) = ∫ _T0 w ₁ (s)・p (s−t) ds v ₂ (t)=∫ _T0 w ₂ (s)・g(s−t)ds w ₁ (t)=r(t)・cos(w _c t+θ^) w ₂ (t)=r(t)・sin(w _c t + θ^) In other words, v ₁ (t) is the output obtained by passing the in-phase demodulated signal w ₁ (t) through a matching filter for the in-phase shaping filter, and v ₂ (t) is the output of the quadrature side demodulated signal w 1 (t). demodulated signal
This is the output obtained by passing w ₂ (t) through a matching filter for the orthogonal side shaping filter. Control over τ^ and θ^ is performed using the following algorithm. That is, to obtain the next specified values τ^ _k+1 and θ^ _k+1 from the current estimated values τ^ and θ^, use K〓 and K〓 as loop gains.

[Formula] Calculating ∂Ω/∂τ^ and ∂Ω/∂θ^ here, ∂Ω/∂τ^ constant× 〓 ^m {v ₁ (mT+τ^)v〓 ₁ (mT+τ^) +v ₂ (mT+τ^)v 〓 ₂ (mT+τ^)} ∂Ω/∂θ^ constant× 〓 ^m {−v ₁ (mT+τ^)y ₂ (mT +τ^)+v ₂ (mT+τ^)y ₁ (mT+τ^)} However, y ₁ ( t)=∫ _T0 w ₁ (s)g(s-t)ds y ₂ (t)=∫ _T0 w ₂ (s)p(s-t)ds. In a normal QAM signal, p(t) = g(t), so y ₁ (t) = v ₁ (t), y ₂ (t) = v ₂ (t), and ∂Ω/∂θ^ is constant. Equally, it becomes 0. That is,
As is well known, NDA type carrier phase control cannot be applied to normal QAM signals. On the other hand, in so-called staggered QAM, g(t) = p(t-t ₀ ) (where t ₀ is the time difference between in-phase and quadrature), and the phase of NDA type for both τ^θ^ control can be applied, especially t ₀ = T/2
When , the control ability is at its maximum. FIG. 2 is a block diagram showing the configuration of a timing/carrier combination control system for a staggered QAM signal with t ₀ =T/2. That is, in FIG. 2, the received signal input to the input terminal 11
4, the output of the voltage controlled oscillator 17 and its
The signal is inversely modulated by an in-phase carrier and a quadrature carrier, which are obtained with a 90° phase shift.
The in-phase side output and the quadrature side output are further supplied to matching filters 15 and 18, respectively. The outputs of the matching filters 15 and 18 are correlated with each other in a multiplier 19, and based on this correlation value, the voltage controlled oscillator 1
7 oscillation phase is controlled. Regarding timing control, for the voltage controlled oscillator 25 that determines the timing phase, there is a correlation between a signal obtained by delaying the output of the matched filter 15 by T/2 seconds and its differential signal, and a correlation between the output signal of the matched filter 18 and its differential signal. This is executed by supplying a control signal obtained by adding the correlations of . The samplers 26 and 27 receive the in-phase signal using the timing clock thus obtained.
The orthogonal signals are each sampled, and their outputs are supplied to a subsequent automatic equalizer or the like via output terminals 28 and 29. FIG. 3 is a block diagram showing a configuration for realizing the control system shown in FIG. 2 entirely through digital processing. sampled by the sampling clock obtained as the output, multiplier 3
3 and 34, the input signal is multiplied by a cosine sample value series and a sine sample value series obtained by driving the read-only memory 35 with the sampling clock to obtain an in-phase signal and a quadrature signal. The in-phase signal and quadrature signal thus obtained undergo phase rotation in the phase rotation circuit 38, and the amount of phase rotation is controlled by the in-phase-quadrature correlation value obtained as the output of the multiplier 40. Among the outputs of the phase rotation circuit 38, the in-phase signal is delayed by T/2 seconds in the delay circuit 39 and then reaches the output terminal 47.
On the other hand, the orthogonal signal reaches the output terminal 48 as it is. A control signal for the voltage controlled oscillation circuit 46 is supplied from an adder 45, and this adder 45 calculates the correlation value between the in-phase signal delayed by T/2 seconds and its differential signal, and the orthogonal signal and its differential signal. Addition is performed with the correlation value of . The offset suppressing carrier timing coupling control method according to the present invention is applied to an orthogonal QAM transmission system, which basically uses a plurality of the above-mentioned staggered QAM signals while allowing spectrum overlap with each other. This is a method of multiplexing and transmitting. FIG. 4 shows the basic configuration of the orthogonal QAM transmission system, and FIG. 5 shows an example of the spectrum of the orthogonal QAM signal sent to the transmission path. In FIG. 4, terminals 51 and 52 each have a first
Channel in-phase data {a ⁽¹⁾ _k }, orthogonal data {b ⁽¹⁾ _k }
is input with a clock cycle of T seconds. The in-phase data {a ⁽¹⁾ _k } is delayed by T/2 seconds in the delay circuit 59 and then passes through the shaping filter 63 to the multiplier 71 . On the other hand, the orthogonal data {b ⁽¹⁾ _k } is the shape filter 6
4 to the multiplier 72. Molded filter 63
and 64 have the same frequency transfer characteristic G(w), and this G(w) is set to a so-called root Nyquist shape. If the impulse response of G(w) is g(t), the output signal U ⁽¹⁾ _R (t) of the shaping filter 63 and the output signal U ⁽¹⁾ _I (t) of the shaping filter 64 are each U ^{(1 )} _R (t)= 〓 ^k a ⁽¹⁾ _k・g(t-T/2-kT) ……(1) U ⁽¹⁾ _I (t)= 〓 ^k b ⁽¹⁾ _k・g(t- kT) ...(2) The multiplier 71, the multiplier 72, and the adder 79 constitute a quadrature amplitude modulator with a carrier angular frequency _w1 , and the output of the adder 79 has the first QAM
A signal g ⁽¹⁾ (t) is obtained. That is, g ⁽¹⁾ (t) is g (t) = U ⁽¹⁾ _R (t) cosw ₁ t + U ⁽¹⁾ _I (t) sinw ₁ t = Re {[U ⁽¹⁾ _R (t) + jU ⁽¹⁾ _I (t)〕e ^-jw ₁ ^t }
...(3) It is expressed as. However, Re{·} represents an operation that takes only the real part. In exactly the same way, the outputs of adders 80, 81, and 82 each receive the second QAM signal.
g ⁽²⁾ (t), 3rd QAM signal g ⁽³⁾ (t), 4th QAM signal
g ⁽⁴⁾ (t) is obtained. These signals are expressed by the following equations. g ⁽²⁾ (t) = Re {[U ⁽²⁾ _R (t) + jU ⁽²⁾ _I (t)] e ^-jw ₂
^t }
...(4) g ⁽³⁾ (t) = Re {[U ⁽³⁾ _R (t) + jU ⁽³⁾ _I (t)] e ^-jw ₃
^t }
...(5) g ⁽⁴⁾ (t) = Re {[U ⁽⁴⁾ _R (t) + jU ⁽⁴⁾ _I (t)] e ^-jw ₄
^t }
……(6) However, U ⁽²⁾ _R (t)= 〓 ^k a ⁽²⁾ _k g(t−kT) ……(7) U ⁽²⁾ _I (t)= 〓 ^k b ⁽²⁾ _k g(t-T/2-kT) ……(8) U ⁽³⁾ _R (t)= 〓 ^k a ⁽³⁾ _k g(t-T/2-kT) ……(9) U ⁽³⁾ _I (t) = 〓 ^k b ⁽³⁾ _k g (t-kT) ……(10) U ⁽⁴⁾ _R (t) = 〓 ^k a ⁽⁴⁾ _k g (t-kT) ……(11) U ⁽⁴⁾ _I (t) = 〓 ^k b ⁽⁴⁾ _k g (t-T/2-kT) ...(12), and {a ⁽²⁾ _k } and {b ⁽²⁾ _k } are {a ⁽³⁾ _k } and {b ⁽³⁾ _k } represent the second channel in-phase data and quadrature data input to terminals 53 and 54, respectively, and the third channel in-phase data and quadrature data input to terminals 55 and 56, respectively. In the orthogonal data, {a ⁽⁴⁾ _k } and {b ⁽⁴⁾ _k } respectively indicate the fourth channel in-phase data and the orthogonal data inputted to the terminals 57 and 58, respectively. Here, the relationship w _k −w _k-1 = 2π/T holds between each carrier angular frequency w ₁ , w ₂ , w ₃ , w ₄ , and the T/2 second delay circuit 59, 60,61,6
2 are inserted alternately on the in-phase side, the orthogonal side, the in-phase side, and the orthogonal side, respectively. This means that the first to fourth QAM signals are orthogonally multiplexed on the frequency axis. Letting this orthogonal multiplexed signal be s(t), s(t) is obtained as the output of the adder circuit 83, and in the example of 4-channel multiplexing as shown in the figure, s(t)= ₄ 〓 ⁿ⁼¹ g ^{(n )} (t) ...(13) Further, the spectrum of the orthogonal multiplexed signal s(t) is as shown in FIG. That is, in FIG. 5, 110, 111, 11
2,113 are the 1st, 2nd, 3rd, and 4th QAM respectively
It represents the spectrum of a signal, and spectrum overlap always occurs between adjacent channels. A feature of the orthogonal multiplex transmission system is that even when such adjacent spectrums overlap, the original baseband data can be restored without error on the receiving side, making it possible to realize a highly efficient data transmission system. Now, in FIG. 4, on the receiving side, the inverse calculation of the calculation performed on the transmitting side is performed. For example, the multipliers 84, 85 and shaping filters 92, 93 are
1QAM signal to in-phase baseband signal v ⁽¹⁾ _R (t),
It is used to restore the orthogonal baseband signal v ⁽¹⁾ _I (t). Here, the frequency transmission characteristics of the shaping filters 92 and 93 are the same as those of the shaping filters 63 and 64 on the transmitting side.
It has exactly the same root Nyquist characteristic as . Therefore, v ⁽¹⁾ _R (t) and v ⁽¹⁾ _I (t) are given by the following equations when there is no noise in the transmission path. (However, w ₁ is
It is assumed to be larger than 2π/T. ) V ⁽¹⁾ _R (t) = (U ⁽¹⁾ _R (t) cosθ ₁ −U ⁽¹⁾ _I (t) sinθ
₁ ) *g
(t) + (U ⁽²⁾ _R (t) cos (θ ₁ −2π/Tt) − U ⁽²⁾ _I (t
) sin (θ ₁ −2π/Tt)) * g (t) ... (14) V ⁽¹⁾ _I (t) = (U ⁽¹⁾ _R (t) sin θ ₁ + U ⁽¹⁾ _I (t) cos θ
₁ ) *g
(t) + (U ⁽²⁾ _R (t) sin (θ ₁ −2π/Tt) + U ⁽²⁾ _I (t
)cos(θ ₁ −2π/Tt))*g(t) (15) Here, θ ₁ is the phase shift of the first carrier on the receiving side, and * represents a convolution operation.
Substituting equations (1), (2), (7), and (8) into equations (14) and (15), we get V ⁽¹⁾ _R (t)= 〓 ^k a ⁽²⁾ _k h ₀ (t− T/2−kT) cosθ ₁ − 〓 ^k b ⁽¹⁾ _k h ₀ (t−kT) sinθ ₁ + 〓 ^k a ⁽²⁾ _k (hc(t−kT) cosθ ₁ + h・(t−kT) sinθ ₁ ) + 〓 〓 ^k b ⁽²⁾ _k (t-T/2-kT) sinθ ₁ -h・(t-T/2-
kT) cosθ ₁ )...(16) V ⁽¹⁾ _I (t) = 〓 ^k a ⁽¹⁾ _k h ₀ (t-T/2-kT) sin θ ₁ + 〓 ^k b ⁽¹⁾ _k h ₀ ( t-kT) cosθ ₁ + 〓 ^k a ⁽²⁾ _k (h _c (t-kT) sinθ ₁ −h _s (t-kT) cosθ ₁ )− 〓 〓 ^k b ⁽²⁾ _k (h _c (t- T/2-kT) _cosθ1 + _hs (t-T/2
−kT) sinθ ₁ ) is obtained. However, h ₀ (t), h _c (t), and h _s (t) are functions defined below. h _p (t)=g(t)*g(t) h _c (t)=g(t) cos2π/Tt*g(t) h _s (t)=g(t) sin2π/Tt*g(t ) Since g(t) is the impulse response of a filter having root Nyquist characteristics, for an integer value m, h _p (mT) = 1 when m = 0, 0 when m≠0, and moreover, as intended. It is already known that h _s (mT)=0 and h _c (mT+T/2)=0 hold for an integer m. Therefore, in equations (16) and (17), θ ₁ =0, and V ⁽¹⁾ _R (t),
If V ⁽²⁾ _R (t) is sampled at the timings of t=mT+T/2 and t=mT, then V ⁽¹⁾ _R (mT+T/2)=a ⁽¹⁾ _n V ⁽¹⁾ _I (mT)=b ⁽¹⁾ _n , and it can be seen that the in-phase data and orthogonal data of the first channel can be correctly restored. 2nd to 4th
Channel in-phase data and quadrature data can also be correctly restored in exactly the same manner. However, since there is generally a carrier phase shift and a timing shift on the receiving side, it is necessary to control these shifts. The present invention focuses on the fact that each channel is a staggered QAM in such an orthogonal QAM transmission system and that all channels operate with a synchronous clock, and can accurately control the carrier phase shift and timing shift described above. A carrier timing coupling control method is provided. Now, in Fig. 4, the true timing shift on the receiving side is τ ^* , and the true carrier phase shifts for the 1st to 4th QAM channels are respectively θ ^* ₁ , θ ^* ₂ , θ ^* ₃ , θ
^* _Four
shall be. The transmission complex envelope of the kth channel,
Letting the received complex envelopes be β _k (t) and γ _k (t), respectively, then γ _k (t)=β _k (t+τ ^* )e ^j 〓 ^k* +β _k-1 (t+τ
^* )e ^-jw0(t+ 〓 ^*)+j 〓 ^k* +β _k+1 (t+τ ^* )e ^jw0(t+ 〓 ^*)
+j 〓 ^k* +n _k (t)k=1, 2, 3, 4 (19). However, β ₀ (t)=β _s (t)=0. Also, w ₀ =2π/T, and n _k (t) is the complex baseband noise of the k-th channel. Since the second and third terms in equation (19) are inter-channel interference components independent of the desired signal in the first term, they can be approximately considered Gaussian noise, and γ _k (t) can be expressed as Write it like this. γ _k (t) = β _k (t + τ ^* ) e ^j 〓 ^k + N _k (t)...(20) However, N _k (t) is the sum of all terms other than the 12th term in equation (19). . If the likelihood function in this system is L(τ, θ ₁ , θ ₂ , θ ₃ , θ ₄ ), then L(τ, θ ₁ , θ ₂ , θ ₃ , θ ₄ )= {transmission complex envelope set β ₁ (t+τ)e ^j 〓 ¹ , β ₂
(t+τ)e ^j 〓 ² , β ₃ (t+τ)e ^j 〓 ³ , β ₄ (t+τ)
e ^j 〓 ⁴
The posterior probability of receiving the received complex envelope set γ ₁ (t), γ ₂ (t), γ ₃ (t), γ ₄ (t) when sending . Here, the noise N ₁ (t), N ₂ (t), N ₃
(t) and N ₄ (t) can be regarded as mutually uncorrelated Gaussian noise, so L=(τ, θ ₁ , θ ₂ , θ ₃ , θ ₄ ) = { ₄ 〓 ^k=1 {β _k (t+τ)e ^j = posterior probability of receiving γ _k (t) when sending ^k }, and the log likelihood function Ω (τ,
θ ₁ , θ ₂ , θ ₃ , θ ₄ ) sent Ω(τ, θ ₁ , θ ₂ , θ ₃ , θ ₄ ) = ₄ 〓 ^k=1 log{β _k (t+τ)e ^j 〓 ^k The posterior probability of receiving γ _k (t) at time} is expressed as follows. If Ω(τ, θ ₁ , θ ₂ , θ ₃ , θ ₄ ) is averaged over the data, then = ₄ 〓 ^k=1 _k (τ, θ _k )...(21) However, _k When k is an odd number, _k (τ, θ _k ) = constant × 〓 ^m V ² _1k (mT + T / 2 + τ) 〓 ⁿ V ² _{2 k} (nT + τ) ... (22), and when k is an even number, (τ, θ _k ) = constant × 〓 ^m V ² _1,k (mT + τ) 〓 ⁿ V ² _2,k (nT + T / 2 + τ) ... (23). Here, V ₁ , _k (t) and V ₂ , _k (t) are respectively V ₁ , _k (t)=∫Re{γ _k (s)e ^-j 〓 ^k }g(s-t)ds V ₂ , _k (t)=∫Im{γ _k (s)e ^-j 〓 ^k }g(s-t)ds, and the signal corresponding to this is output terminal 1 in Fig. 4.
01-108. That is, the output end 10
1,102, the in-phase component V _1,1 (t) and quadrature component ₂ V _2,1 (t) of the first channel demodulated baseband signal are obtained, and the output terminals 103, 104 have V _1,2 (t ), V _2,2
(t), but the output terminals 105 and 106 have V _1,3 (t),
V _2,3 (t), but V _1,4 at the output terminals 107 and 108
(t) and V _2,4 (t) are obtained, respectively. From the above equation (21), the following control method for the timing deviation τ and the carrier phase deviations θ ₁ , θ ₂ , θ ₃ , and θ ₄ can be obtained. That is, the current estimate τ ⁽ⁱ⁾ ,
^The _next ^estimated _value ^{τ (} _{i +} ^1
) , θ ₁ ⁽ⁱ⁺¹⁾ ,
To obtain θ ₂ ⁽ⁱ⁺¹⁾ , θ ₃ ⁽ⁱ⁺¹⁾ , and θ ₄ ⁽ⁱ⁺¹⁾ , let K〓, K〓 be the loop gain, And it is sufficient. In the orthogonal QAM transmission system, each channel is staggered QAM, so as mentioned above, NDA control is possible with respect to timing and carrier phase. Furthermore, since the carrier frequency offset amount is common to all channels, the amount of temporal variation in phase difference information detected in each channel is common to all channels. Furthermore, regarding timing, since all channels are in a synchronous relationship, it is understood that common control can be applied by adding up all the timing control signals obtained from each channel. FIG. 6 is a block diagram showing an embodiment of the present invention based on the above principle. That is, Fig. 6 shows the fourth
This is an example in which the offset suppressing carrier timing coupling control method according to the present invention is applied to a system in which all of the receiving section processing is digitalized, and 122 is a digital circuit consisting of a demultiplexing circuit, a polyphase circuit, and an inverse offset Fourier transform circuit. This digital signal processing demodulation unit 1 represents a signal processing demodulation unit.
The four outputs of 22 are output terminals 101 to 1 in FIG.
T/ of the first channel complex baseband signal to the fourth channel complex baseband signal obtained in 08
It is a 2 second sampling series. The detailed contents of the demultiplexing circuit, polyphase circuit, and inverse offset Fourier transform circuit can be found in Reference 5 in the Showa era.
See '55 Patent Application No. 28740. In FIG. 6, the sampling circuit 121 has an input terminal 1.
The received signal obtained at 20 is sampled by a sampling clock having a frequency s. Here, s and the baud rate 1/T of each channel usually maintain a simple integer ratio relationship, and therefore, the so-called timing time is determined based on this sampling clock.
Therefore, in this system, the timing control signal controls the VCO 119 which generates the sampling clock. 123 to 126 are phase rotation circuits for the first channel to the fourth channel, and 127 to 1
30 is a phase detection circuit for each channel. The phase rotation circuits 123 to 126 are configured as shown in FIG. 9, for example. That is, in FIG. 9, 160 and 161 are input terminals into which the in-phase and quadrature components of the demodulated baseband signal are input, respectively, 162, 163, 164, and 165 are multipliers, and 166 is an adder. , 167 is a subtracter, 168 is a ROM in which sine wave series and cosine wave series are written, 169 is a 1 sample delay circuit, 170 is an adder, 172, 1
Reference numeral 73 denotes an output terminal from which the in-phase and quadrature components of the baseband signal subjected to phase rotation are output. Now, if a phase control signal φ is input to the control terminal 171, the ROM 168 outputs the 2 phase control signal corresponding to φ.
Addresses are accessed using decimal codes.
Internal data is written in the ROM 168 so that cosφ and sinφ are output in response to an input of φ.
The cosφ obtained as the first output of the ROM 168 is multiplied by the in-phase signal and the quadrature signal by multipliers 162 and 165, respectively. Similarly, the second output is
sinφ is an in-phase signal in multipliers 163 and 164, respectively;
Multiplied by orthogonal signal. Furthermore, the output of the multiplier 162 and the output of the multiplier 163 are summed together in an adder 166, and this sum signal is outputted to the output terminal 172 as a new in-phase signal. A subtracter 167 takes the difference between the output of the multiplier 164 and the output of the multiplier 165, and outputs this difference signal to the output terminal 173 as a new orthogonal signal. By this operation, the input complex signal undergoes a phase rotation of (-φ) and is output. The phase control signal is obtained as the output of the phase difference detection circuits 127, 128, 129, and 130 shown in FIG. 6, and these phase difference detection circuits are configured as shown in FIG. 7, for example. That is, in FIG. 7, 140 and 141 are input terminals into which the in-phase and quadrature components of the complex baseband signal are input, and 142 and 143 are T/2 second delay circuits,
144 and 145 are multipliers, 146 is a subtracter, 147 is a T-second delay circuit, 148 is a subtracter, 149 is an output end of the phase difference signal, and 150 is carrier offset information. is the output end of Which of the input terminals 140 and 141 the in-phase component is input to depends on how the T/2 second delay is applied on the transmitting side. That is, for channels such as the first and third channels in which the in-phase component is delayed by T/2 seconds on the transmitting side, the in-phase component is allocated to the input end 140 and the orthogonal component is allocated to the input end 141. For channels such as the second and fourth channels where the orthogonal portion is delayed by T/2 seconds on the transmitting side, the opposite allocation is performed. In this way, the difference component between the correlation values of the two inputs and the correlation values of the two T/2 seconds delayed inputs is outputted to the output terminal 149 as a phase difference signal. Further, the time variation of the phase difference signal is outputted to the output terminal 150 as carrier offset information. Here, since the carrier frequency offset amount is common to all channels, the amount of change in phase difference information detected in each channel is common to all channels. The phase shifts θ ₁ , θ ₂ , θ ₃ , θ ₄ of each channel are corrected by phase rotation circuits 123, 124, 125, 126, respectively, but the control signals for each phase rotation circuit are sent to adders 132, 133, 134. , 135, respectively. That is, adders 132, 133,
134 and 135 add the static phase difference signal specific to each channel and the carrier offset information common to all channels obtained as the output of the adder circuit 131 to generate a control signal for the corresponding phase rotation circuit. By making all channels common in this way, a carrier offset control signal with a high signal-to-noise ratio can be obtained, so the loop gain can be increased accordingly, and a carrier phase control system with good followability for carrier offset can be obtained. It turns out. Next, the timing deviation τ is 136, 13 in Fig. 6.
The timing deviation detection circuits shown at 7, 138, and 139 each independently detect the detection outputs, and all these detection outputs are added together in the adder circuit 118 and sent to the VCO 119.
will be controlled. Each timing deviation detection circuit is configured as shown in FIG. 8, for example. That is, 151,1 in FIG.
52 is an input terminal into which the in-phase and quadrature components of the phase-corrected complex baseband signal are input;
3,154 is a differential circuit realized as a discrete value system, 155,156 is a multiplier, and 157
is an adder, 159 is a T/2 second delay circuit, and 158 is an output terminal. In a timing shift detection circuit for odd-numbered channels such as 136 and 138 in FIG. 6, in which the in-phase signal is delayed by T/2 seconds with respect to the quadrature signal on the transmitting side, An in-phase signal is input to each. In FIG. 8, the multiplier 155 is connected to the input terminal 151.
The multiplier 156 calculates the correlation between the signal input to the input terminal 152 and its differential signal and the signal obtained by delaying the signal input to the input terminal 152 by T/2 seconds. The correlation value is added by adder 15
After being added in step 7, the signals are outputted to a terminal 158 as a timing shift signal. This timing shift signal is common to all channels except for noise that mixes independently between channels, so the output of the adder circuit in Figure 6 is a timing control signal with a high signal-to-noise ratio. can get. In one embodiment of the carrier timing coupling control method according to the present invention shown in FIG. 6, when the group delay distortion amplitude within the transmission band is sufficiently small compared to T seconds,
It can be applied as is. However, in a normal carrier band, a significantly large group delay distortion may occur, especially near the band edges. For such a line, the embodiment shown in FIG. 6, in which the carrier offset control signal and timing control signal are made common to all channels, cannot be used. However, even for such lines, it is possible to ensure good tracking characteristics for carrier timing by sharing carrier offset control signals and timing control signals with only a few channels whose spectrum is arranged near the center of the band. Can be done. FIG. 10 is a block diagram showing a general embodiment of the offset suppressing carrier timing coupling control method according to the present invention derived from the above viewpoint, and 182 is the digital signal processing demodulation section of FIG. 122. Also, 183 to 1
86 corresponds to 123 to 126 in FIG.
This is a phase rotation circuit for channels to the fourth channel. Furthermore, 187 and 188 are 128 and 1 in Figure 6.
It is a phase difference detection circuit corresponding to 192,
193 is a timing shift detection circuit corresponding to 137 and 138 in FIG. In a general embodiment of the present invention shown in FIG. It has become common. As a result, the timing deviation and carrier deviation of the second channel and the third channel are controlled. On the other hand, the first channel and the fourth channel are not provided with a phase difference detection circuit or a timing shift detection circuit. However, the amount of time variation in timing deviation is common to all channels regardless of line distortion, and this common deviation is
It is compensated by a control loop consisting of a VCO 195, a sampling circuit 181, and the like. In addition, phase rotation circuits 183 and 186 are provided in the first channel and the fourth channel.
are provided, and each phase rotation amount is determined by an adder 1.
Since it is controlled by the control signal obtained as the output of 89, the time variation of the carrier phase shift is also compensated for. That is, even for the first and fourth channels that are not shared, there is no temporal variation in timing or carrier phase in the output complex signals, and only static phase shifts remain. It is known that such static timing deviations and carrier phase deviations can be absorbed by the two-dimensional automatic equalizer shown in Reference 3. As described above, according to the present invention, a carrier timing coupling control system with excellent tracking characteristics can be obtained. Note that the embodiment shown in FIG.
It is clear that this embodiment can be regarded as a special case of the general embodiment shown in FIG.

[Brief explanation of drawings]

FIG. 1 is a block diagram showing a general process of generating a two-dimensional modulated signal, and 3 and 4 are an in-phase baseband shaping filter and an orthogonal baseband shaping filter, respectively. Figure 2 shows T/2 between in-phase and quadrature.
12 is a block diagram showing the configuration of a carrier timing combination control method for staggered QAM signals having a delay difference of seconds;
is a multiplier, 13 is a 90° phase shifter, 15,
18 is a matching filter, 16 is a T/2 second delay circuit,
17 and 25 are VCOs, 20 and 23 are differentiating circuits, 2
4 is an adder. FIG. 3 is a block diagram showing a configuration for realizing the timing/carrier coupling control system of FIG. 2 entirely by digital processing, in which 32 is a sampler, 35 is a ROM, and 38 is a phase rotation circuit. FIG. 4 is a diagram showing the basic configuration of an orthogonal QAM transmission system, and FIG. 5 is a diagram showing an example of spectrum arrangement of orthogonal QAM signals sent to a transmission line in this transmission system. FIG. 6 is a block diagram showing a specific embodiment of the offset suppressing carrier timing coupling control system according to the present invention, in which 122 is a digital signal processing demodulation section;
23 to 126 are phase rotation circuits, 127 to 13
0 is a phase difference detection circuit, 132 to 135 are adders, 136 to 139 are timing shift detection circuits, 118 and 131 are addition circuits, and 119 is an adder.
VCO 121 is a sampling circuit. FIG. 7 is a block diagram showing a specific configuration example of the phase difference detection circuit, and 142 and 143 are T/2 second delay circuits;
147 is a T second delay circuit, 144 and 145 are multipliers, and 146 and 148 are subtracters. FIG. 8 is a block diagram showing a specific example of the configuration of the timing deviation detection circuit, and 153 and 154 are differentiating circuits,
155, 156 are multipliers, 157 is an adder, 15
9 is a T/2 second delay circuit. FIG. 9 is a block diagram showing a specific example of the configuration of the phase rotation circuit, in which 162 to 165 are multipliers, 166 is an adder,
167 is a subtracter, 168 is a ROM, 169 is a one-sample delay circuit, and 170 is an adder. 10th
The figure is a block diagram showing a general embodiment of the offset suppressing carrier timing coupling control system according to the present invention, in which 183 to 186 are phase rotation circuits, 187 and 188 are phase difference detection circuits, and 19
2, 193 is a timing shift detection circuit, 194 is an adder circuit, and 195 is a VCO.

Claims (1)

[Claims] 1. A signal in which the first to Nth orthogonal amplitude modulation signals are orthogonally multiplexed in the frequency domain is received via a transmission path, and this is sampled by a sampling clock to perform analog-to-digital conversion, wave manipulation, and discrete In the reception system of the orthogonal multiplexed signal which is demodulated into the first to Nth complex baseband signals by Fourier transform, a phase rotation operation is performed to independently give a phase rotation to the plurality of complex baseband signals, and a predetermined The sampling clock is controlled by a signal obtained by adding all m timing control signals obtained from the phase-rotated outputs of the k-th to km-th complex baseband signals and their differential signals. In addition to controlling the phase, the k-th to km-th
The amount of frequency offset is obtained by adding up all the time changes of the correlation values between the real part and the imaginary part of each of the kth to kmth complex baseband signals, and the amount of phase rotation for the kth to kmth complex baseband signals is It is controlled by the sum signal of the correlation value between the real part and the imaginary part of each complex baseband signal and the frequency offset amount, and the phase rotation amount for the complex baseband signals other than the k-th to km-th is controlled by the sum signal of the sum signal. An offset suppressing carrier timing coupling control system characterized in that the carrier timing is controlled by at least one of the following.