JPH01268457A - Pulse-width modulation control apparatus of inverter - Google Patents

Abstract

PURPOSE:To reduce an electromagnetic noise by dividing the ON time of a switching element operated repeatedly with an operation period corresponding to a carrier frequency into a plurality of pulses and by ON-controlling each element by means of said pulses. CONSTITUTION:An inverter pulse-width modulation control apparatus is connected with a three-phase winding 2 and equipped with a bridge circuit 4 having a plurality of switching elements, and pulse width-modulates DC by the ON-OFF operation of said respective elements. Further, said control apparatus is provided with an operating means 10 operating the ON time of each switching element with an operation period corresponding to a carrier frequency, a pulse-dividing means 11, and a control means 12, and composed of a one-chip microcomputer 8. Thus, also when said carrier frequency takes a normal value, the ON time (PWM control pattern) of each element is repeatedly operated for every operation period. However, because said ON time is divided into a plurality of pulses, this is the same as the case where PWM control is performed at a high carrier frequency.

Description

DETAILED DESCRIPTION OF THE INVENTION (Industrial Application Field) The present invention relates to a pulse width modulation control device for an inverter, and particularly relates to an improvement in a device that increases the carrier frequency and performs precise waveform control.

(Prior art) In recent years, MOS F has been used as a high-speed switching device.
Elements such as ET (metal oxide gate field effect transistor) have appeared, and if they are adopted for pulse width modulation control of inverters, precise waveform control becomes possible, reducing electromagnetic noise and increasing motor efficiency. It has become possible to obtain such effects.

Therefore, in the past, high carrier frequencies (for example, 20 KIIz
) is known that enables pulse width modulation control using the method described above to ensure the above-mentioned effect of reducing electromagnetic noise and the like. (For example, "Application of high frequency switching to general-purpose inverters J,
(See presenter, Chihiro Okadochi, etc.).

(Problems to be Solved by the Invention) However, the conventional device described above has disadvantages such as a complicated circuit, complicated various adjustments, and high cost.

Therefore, it is conceivable to use a one-chip microcomputer (hereinafter referred to as a microcomputer), which is inexpensive and has a simple circuit configuration. The calculation time is long, for example, about 200 μs, and the carrier frequency remains at about 5Kllz at maximum. For this reason, pulse width modulation control using a high frequency (20 KHz or higher) carrier frequency is generally difficult.

The present invention has been made in view of these points, and its purpose is to create a situation that is apparently equivalent to raising the carrier frequency, while employing a single-chip microcontroller.
The object of the present invention is to enable pulse width modulation control at an equivalently high carrier frequency with a low-cost and simple circuit configuration, thereby achieving effects such as reducing electromagnetic noise and increasing motor efficiency through precise waveform control.

(Means for Solving the Problem) In order to achieve the above object, the invention according to claim (1) changes the generation algorithm of the PWM control pattern (that is, the ON time of a plurality of switching elements provided in the inverter). Even if the calculation time (calculation cycle) of the PWM control pattern is long, each calculated ON time is divided into a plurality of pulses to equivalently increase the carrier frequency.

As shown in FIGS. 1 and 2, the specific solution is a bridge circuit (4) connected to a three-phase winding (2) and having a plurality of switching elements (Tra) to (Trc').
, and each switching element (
The present invention is based on an inverter pulse width modulation control device that applies three-phase AC voltage to the three-phase winding (2) by pulse-width modulating the DC through ON/OFF operations of Tra) to (Trc'). . As shown in FIGS. 6 and 7, there is a calculation means (10) for calculating the ON time of each of the switching elements (Tra) to (Trc') at a calculation period according to the carrier frequency; ON of each switching element (Tra) to (Trc゛) calculated by lO)
dividing means (11) for dividing time into a plurality of pulses;
A control means (12) is provided for controlling each of the switching elements (Tra) to (T'C') to ON using a plurality of pulses divided by the dividing means (11).

In addition, in the invention according to claim (21), the above claim (1)
The dividing means (11) of each switching element (Tra)
The ON time of ~(Trc') is specified to be equally divided into a plurality of preset fixed pulses.

(Function) With the above configuration, in the inventions of claims (1) and (2), each switching element (Tra) to (Tr
The ON time (PWM control pattern) of e') is repeatedly calculated by the calculation means (10) at each calculation period according to the carrier frequency, and the ON time (PWM control pattern) of each switching element (Tra)
Since the ON time of ~(Trc') is divided into a plurality of pulses (for example, 4 pulses) by the dividing means (11), the carrier frequency is multiplied by this number of divisions, and the carrier frequency is equivalently high (for example, The same situation occurs when pulse width modulation control is performed at a frequency of about 20 KHz).

As a result, each of the switching elements (Tra) to (Trc') is controlled by the control means (12) by each of the divided pulses.
), a precise output waveform close to a sine wave is obtained, electromagnetic noise is effectively reduced, and motor efficiency is effectively increased.

Here, the carrier frequency of pulse width modulation control is the normal value (
(approximately 5KHz), and even a single-chip microcontroller with a long calculation time can sufficiently calculate the PWM control pattern.
Pulse width modulation control using a high carrier frequency can be performed at low cost and with a simple circuit configuration.

In particular, in the invention according to claim (2), the dividing means (11)
is O of each switching element (Tra) to (Tre')
Since the N time is configured to be equally divided into a plurality of pulses that are fixed in advance, the switching element (T
The time required to divide the ON time of ra) to (Trc') can be further shortened, the calculation process can be shortened accordingly, and pulse width modulation control using a carrier frequency that is higher than that is possible.

(Example) Embodiments of the present invention will be described below with reference to the drawings.

1 and 2 show a pulse width modulation (hereinafter abbreviated as PWM) control device for an inverter according to the present invention. In each figure, (1) has three windings (2a), (2b),
(2e) is a Y-connected three-phase winding (2); (3) is a voltage-type inverter connected to the induction motor (1); A transistor bridge circuit (4) connected to the three-phase winding (2) of the motor (1) is provided, and the bridge circuit (4) includes free-wheeling diodes (Da) to (Dc'), respectively.
A plurality of (six) transistors (switching elements) such as MOSFETs (Tra), (Tra'), (
Trb), (Trb'), (Trc), (T
rc'). Therefore, the inverter (3) includes:
A DC voltage is applied from a rectifier (6) that rectifies the three-phase AC of the three-phase power supply (5).

Further, (8) is a one-chip microcomputer that forms the ON time of the six transistors (Tra) to (Tre') of the bridge circuit (4), that is, the pwM control pattern. is each of the above transistors (T
The microcomputer (8a) is equipped with a pace driver (8a) that turns ON/OFF ra) to (Trc').
) of the transistor (Tra) ~ (Trc')
The direct current is pulse width modulated by N/OFF control.

Next, the formation of a PWM control pattern by the microcomputer (8) will be explained.

To summarize, this PWM control pattern is formed by determining a PWM control pattern so that the locus of time integration of the output voltage approaches a circular locus.

To explain this in detail, first, let the potentials of the output terminals of the inverter (3) be vH, vb, vc, the potential of the neutral point of the three-phase winding (2) be vn, and the output defined by the following formula. Consider a voltage vector vP and a time integral ap of the voltage vector vP.

Vp -J Manchoφ (va +a2・*b +d 11vc lHowever, 1.
w ej2/3・π ap=J'Vpdt Now, when an angular frequency is applied to the three-phase winding (2) of the induction motor (1), the voltage vector vp and its time integral λP follow a circular locus on the complex plane. draw.

On the other hand, in the voltage source inverter (3), one of the transistors in each phase arm is always in the ON state, so for convenience, the ON state on the + side is set to "1", and the ON state on the - side is set to "0".
”, and “101” for C phase, b phase, C phase, ro
When expressed as llJ, etc., there are eight states of the inverter (3). Voltage vector Vp (P-0~
7) has a magnitude of JT'ITV d (V d voltage across the rectifier (6)) and its direction is the direction shown in FIG. Here, V□, Vl 1; kV
There is a zero vector at □-Vl-0.

The time integral jp of the above voltage vector is d2p/dt −V
Since p, the time integral J when driving the inverter (3)
p is 1Vpl-J″gTv in the direction of voltage vector vp
It moves at a speed of d (however, it stops if the vector is zero).

From the above, the PWM control pattern of the voltage source inverter <3> is such that the voltage vector vP is appropriately selected so that the vector locus on the complex plane of the time integral λP of the voltage vector moves at the angular velocity ω along the circumference of the specified radius R. and decide. (The designated radius R is R-V, /ω, where Vl is the effective value of the line voltage of the fundamental voltage and ω is the angular frequency.

That is, for example, as shown in FIG. 4, the angle φ is O≦φ≦π
/3, using voltage vectors V4, V6 and a zero vector (e.g. vo), stays at point PO for time τ0 (this state is indicated by symbol O), and then changes ■4 to time τ
Let us consider the case where the point q1 is reached by taking 4, and the point P1 is reached by taking vc for a time τ6. In this case, △PO
In QIP+, po P, =V, -T.

Po q + −JT”IT vc+ φτ4Qlpl
-v'manchoVd ・τ6 and τ0+τ4+τ6 ``”TO,
Solving the above equation, the voltage vector '4+Vs within the period TO
, To take time τ4. τ6. τ0 is obtained.

ra/To −ks−8ln(π/3−φ0)76
/ TO-ks-31n φO r□ / T□ -1-ks-3In(φo+π/3)
(3) However, kS is the voltage control rate, and is ks - JTV+ / Vd.

Equation (3) above is a relational expression in the range of angle φ 0≦φ≦π/3, but in other sections, inverter (3) performs symmetrical three-phase operation, so the table shown on the next page By replacing each symbol as follows, a relational expression in the range of 0≦φ≦2π can be obtained.

Next, the ON/Off of each transistor (Tra) to (Trc') is determined based on the time τ of the voltage vector in equation (3) above.
Find the F pattern (PWM control pattern). In this case, the relationship between the voltage vector time τ and the PWM control pattern changes depending on the order in which the voltage vectors are taken. If we add the constraint that the ON/OFF switching of the transistor is only once, the PWM control pattern is represented by 4 nomi turns shown in Figure 5 (a) to (d) (in the figure, τ" is on the + side). ON of the transistor of
time, and τ- indicates the ON time of one side transistor, respectively).

In this embodiment, the PWM control pattern shown in FIG. In the voltage source inverter (3), the PWM control pattern is uniquely determined by knowing the name of the transistor that turns ON at the beginning of the period To and the time at which it turns OFF. Referring to figure (a), the PWM control pattern is determined by the following formula when the angle φ is in the range of 0≦φ≦π13.

r a -/To = 1-JT・(V+/Vd)・
5in (φ0+ π/3) rb −/To −t−v/To (V+ /Vd)
・Slnψ.

τc/To-1 (always ON) ・・・・・・(4) In the relational expression (4) of the PWM control pattern in the range of O≦φ≦π/3, each symbol is replaced in the same manner as above. Then 0
The relational expression is within the range of ≦φ≦2π.

Next, the operation of the 1-chip microcontroller (8) is shown in Figures 6 and 7.
This will be explained based on the control flow shown in the figure with reference to FIG.

The control flow in FIG. 6 is as follows:
This is a calculation flow for the ON time (PWM control pattern) of (Tre'), and the control flow in FIG. 7 is a flow for actually controlling ON of each transistor (Tra) to (Trc'). First, to explain the control flow in FIG. 6, the calculation period To(
For example, it is repeated every 200 μs, and step SAI
After inputting the output voltage digit ωt (-φ0) and the output voltage amplitude V1 in step SA2, each transistor (Tra
) to (Trc'), the ON time τ(n+1) is calculated.

After that, in step sA3, the ON time τ(n
+1) by a preset value N (for example, 4), each ON time τ(n+1) is divided by a plurality of N (4) pulses r'(nil)(r'(n+1) −r(n+1)
)/4).

Then, in step SA4, this divided pulse τ' (
nil) in one switching time register for each phase (in a voltage type inverter, one transistor in each phase arm is always in the ON state, so one for each phase is sufficient), and the process returns.

In addition, in the control flow of FIG. 7, the repetition period To° is earlier than the calculation period To of FIG. 6, and the ON time τ
According to the number of divisions N (4 pieces) of (n+1), To'-To
/N (the number of divisions N is preferably selected to be N-2N-2 (, 2, . . . ), where division can be executed only by shifting). After the divided pulse τ'(n+1) is stored in the switching time register of each phase in the control flow shown in FIG. 6, as shown in FIG. Input the contents of the switching time register in Sol, and in step SB2 divide pulse τ
' (nil) indicates the corresponding transistor (Tra
) to (Trc') are turned ON and the process returns.

Therefore, in the control flow of FIG. 6, step SAI
, SA2, the above PWM control pattern relational expression (4
) based on each transistor (switching element) (Tr
It constitutes a calculation means (10) configured to calculate the ONN time (nil) of a) to (Tre').

Further, in step SA3, the ONN time (nil) of each transistor (Tra) to (Trc') calculated by the calculation means (10) is divided into a plurality of N (N-4) pulses τ' (H+1). Dividing means (11)
It consists of Furthermore, according to the control flow in Figure 7,
A plurality of pieces N (N-1) divided by the above dividing means (1■)
) constitutes a control means (12) configured to turn on each of the transistors (Tra) to (Trc') with a pulse τ' (nil).

Therefore, in the above embodiment, as shown in FIGS. 8 and 9, each transistor (Tra) to ( Trc') ON time τ(n
+1) is calculated by the calculation means (10), each ON time τ(n+1) is divided into a plurality of N(
4) pulses τ' (n+1) (τ'(nil)-
τ(nil)/4), this divided pulse τ
'(n+1) is a.

The switching time is stored in the switching time register for each phase b and e.

Then, in the next calculation period To, as shown in FIG. 8, each transistor (Tra
-Trc') ONN time (n+2) calculation,
At the same time, in this current period TO, every period of T o/N (=T o'), the previous period T
Each of the corresponding transistors (Tra) to (Trc') is controlled to be turned on by the control means (12) using the divided pulse τ' (n+1) in the switching time register for each phase of a, b, and c determined by O. Therefore, the frequency of the high-frequency component can be increased compared to the conventional one (which does not divide the ON time into multiple pulses) as shown in FIG. It can be multiplied by -4> times, and the original carrier frequency (5KIIZ) can be made into a high carrier frequency (20KIIz).In addition, in Figures 8 and 16, the positive side transistors of each phase are This is a description of the case where ON time is calculated.

Here, the original carrier frequency (5Kilz), that is, the calculation period T o (200 μs) of the ON time, is 1
Since the period is long enough to calculate the PWM control pattern even with a chip microcomputer (8), it is possible to perform PWM control at a high carrier frequency (about 20!lz) while using a single chip microcomputer (8). This makes it possible to precisely control the three-phase AC waveform to the induction motor (1) by making use of the ability of high-speed switching elements such as MOSFETs at low cost and with a simple circuit configuration, which reduces electromagnetic noise. It is possible to reduce this and increase motor efficiency.

In addition, the carrier frequency (5Kilz) is the same as before.
If this is sufficient, the calculation time of the one-chip microcomputer (8) can be shortened, and the processing capacity for processes other than PWM control can be increased.

Moreover, in the dividing means (11), each transistor (Tr
a) The ONN time (n+1) of ~Trc') is set to a preset value N (N-4) and a plurality of N (4) equal width pulses τ' (nil) (τ' (n+1)-τ( Since it is divided into 1) and 1) and 1) and 2), it is only necessary to perform division, and the calculation time can be shortened, and the calculation period To of the control flow in Fig. 6, that is, the carrier frequency of PWM control, can be made higher. I can do it.

In the above embodiment, each transistor (Tra) to Tr
c') ON time of PWM control pattern relational expression (4)
After calculating based on , this was divided into a plurality of N (N-4), but the above relational expression (4) was converted to a preset value N (N-4).
The divided pulse τ'(nil
) may be calculated.

Furthermore, the part that converts the contents of the switching registers of each phase into pulse widths may be processed by hardware such as an external pulse width modulation IC. Furthermore, with the configuration shown in FIG. 9, even if the carrier frequency changes due to a change in the switching element, it is sufficient to change only the dividing means (11) and the control means (12). Furthermore, if the switching time register is shared with the register of the pulse width control section (step 582), the process of step SBI in FIG. 7 can be omitted.

Furthermore, the calculation period To in the calculation flow of the PWM control pattern is uniquely determined by the time required to actually calculate the PWM control pattern, but the period To of ON control of the transistor in the control flow of FIG. ° is determined depending on the desired carrier frequency, and for this purpose, the value of the division number N (To/To') of the ON time of each transistor may be set to an appropriate value.

Further, FIGS. 10 to 14 show modified examples of the present invention, and in the above embodiment, a plurality of N (hereinafter, N-4
In this case, each pulse is divided into unequal widths instead of having equal widths. In other words, the ON time τ(n) of the transistor in period To and the ON time τ(n+1
) is the result of linear interpolation.

The above modification will be explained in detail. In the control flow shown in FIG. 10, calculation processing is performed in a period To cycle, and in step Scl, the phase ω and amplitude V1 of the output voltage are input, and in step SC
2 for each previous transistor (Tra) ~ (Trc')
Input the calculation result of the ONN time (n-1) (4 divided pulses).

After that, in step SC3, each transistor (Tra) to (
The ON time τ(n) of “Tre”) is calculated, and the divided pulses τ゛(n) divided into a plurality of N (4 pulses) are calculated from this ON time τ(n). Then, in step Sc4, each divided pulse Divided pulses τ of transistors (Tra) to (Trc').

The previous divided pulse τ'(
An interpolated value Δτn-1 of n-1) is calculated based on the following formula.

Δτn-1"'τ°(n)-τ'(n-1)lhaN;
The number of divisions is then the previous four divided pulses τ'(n-1) as this interpolated value Δτ. -1 to perform gradual interpolation, step SC5
Δ for each divided pulse τ'(n-1) sequentially from the first one.
τ, 2-Δτ, 3・Δτ, 4n-1n-I
n-1 ・Δτ. -1 is added, and in step SC6, each divided pulse is stored in a plurality of N (N-4) switching time registers for each phase, and in step SC7, each divided pulse τ'(n-1) is Remember and return.

Furthermore, the control flow in FIG. 11 is based on the period T during which the divided pulse τ' was calculated and stored as shown in FIG.

Each of the transistors (Tra) to (Trc') is controlled to be turned on by each divided pulse τ° in the second period To from 2nd period To, and the control period To° is equal to l of the calculation period To of the control flow in FIG. /N (N is the number of divisions).

In this control flow, as shown in FIG. 13, the divided pulse τ' for each phase stored in the first switching time register is read in step SDI, and then the switching time register is shifted in step SD2. SD3
Then, the corresponding transistors (Tra) to (Trc') are turned on with the read divided pulse τ' (n-1) and the process returns.Then, in the same manner, the second, Reading the divided pulses for each phase stored in the third and fourth switching time registers and controlling each transistor (Tra) to (Trc') to turn on is repeated.

Therefore, in the above modification, as shown in FIG. 12, for example, in the middle period To, the ON time of the transistor τ(
The divided pulse τ°(n) obtained by dividing n) into four is the first period T
If “0” is output, in the next period T'o, a divided pulse that is larger than this divided pulse by an interpolated value Δτ.-1 is output repeatedly in the control period T°0, so the 14th
As shown in the figure, good reproducibility of the waveform can be achieved with respect to the average value of the output voltage at the control period T°0 corresponding to the equivalent carrier frequency.

Furthermore, FIG. 15 shows another modification example, and it is possible to determine whether the ON time of each transistor (Tra) to (Trc') is divided by equal-width pulses or by unequal-width pulses. The selection is made as appropriate depending on the degree of change in the signal wave and the phase of the signal wave.

In other words, in the figure, in the range where the phase slope of the signal wave (indicated by the broken line in the figure) is large, the ON time is divided by unequal width pulses, and in the range where the slope is small, the ON time is divided by equal width pulses. I try to do it with

In this modified example, in the phase range with a large slope, it is possible to improve the waveform reproducibility compared to dividing by equal width pulses, and in the phase range with a small slope, good phase reproducibility can be ensured. However, compared to the case where pulses are divided by unequal width pulses, the calculation and processing time can be saved by eliminating the need to calculate interpolated values.

In the above embodiment, the PWM control pattern was obtained based on the relational expression (4) in the case of voltage vector control.
Of course, it may be determined based on the relational expression for other PWM control methods such as the triangular wave comparison method.

(Effects of the Invention) As explained above, according to the pulse width modulation control device for an inverter of the present invention, the ON time of the switching element, which is repeatedly calculated at the calculation cycle according to the carrier frequency, is divided into a plurality of pulses. , Since each switching element was controlled to be turned on using this divided pulse, the O of the switching element was
Even if it takes a relatively long time to calculate N hours, the carrier frequency can be equivalently increased, and even if a low-cost, simple-circuit one-chip microcontroller is used, for example, three-phase AC waveforms can be precisely generated. waveform control, reducing electromagnetic noise and increasing motor efficiency.

In particular, each switching element (Tra) ~ (Trc')
When the ON time is equally divided into a plurality of preset fixed pulses, the time required for dividing the ON time can be further shortened, and PWM control can be performed at an even higher carrier frequency.

[Brief explanation of the drawing]

Figures 1 to 15 show embodiments of the present invention. Figure 1 is a general schematic diagram, Figure 2 is an electric circuit diagram, and Figure 3 shows various states of the voltage source inverter using eight types of voltage vectors. The displayed explanatory diagrams, Fig. 4, are explanatory diagrams of voltage vector control to bring the locus of the time integral of the voltage vector on the complex plane closer to a circular locus, and Figs. 5 (a) to (d) each show the angle φ 0≦φ
An explanatory diagram of the types of PWM control patterns that can be taken within the range of ≦π/3, Figures 6 and 7 are flowcharts each showing ON/OFF control of each transistor by a 1-chip microcomputer, and Figure 8 is a diagram showing the carrier frequency FIG. 9 is an explanatory diagram showing an equivalently higher position, and FIG. 9 is an explanatory diagram of the operation. 10 to 14 show modified examples, FIGS. 10 and 11 are flowcharts showing ON/OFF control of each transistor, FIG. 12 is an explanatory diagram of interpolation of divided pulses, and FIG. 13 14 is an explanatory diagram of the operation, and FIG. 14 is an explanatory diagram of the waveform reproducibility. Further, FIG. 15 is an explanatory diagram showing a phase range of a signal wave in which division by equal width pulses and division by unequal width pulses are selected, showing another modification. Furthermore, the first
FIG. 6 is an explanatory diagram showing a conventional example. (2)...Three-phase winding, (3)...Voltage source inverter, (4)...Bridge circuit, (Tra) to (Trc"
)...Transistor, (8)...1-chip microcomputer, (10)...Calculating means, (11)...Dividing means,
(12)...control means. fJ3 figure/ / Figure 4 1st rf! J (Bridge port 2r) Figure 2 (A) (B) Voltage vector Vo + V4YV6 V4
◆VS, Vq@Tora, :/Jisu7Ta011
111b phase)-5);X? Tb OO1011C
Sotra No Jis9TCOOOO○ 1st P(c)(d) 7 V7+V6+V4V6-J-■◆v01 Figure 10 Figure 11 Figure Procedure Amendment (Method) August 4, 1988 1, Incident Display 1988 Year: Patent Application No. 95200 2, Name of the invention: Inverter pulse width modulation control device 3, Relationship with the person making the amendment: Patent applicant address: Umeda Center Building, 2-4-12 Nakazaki Nishi, Kita-ku, Osaka-shi, Osaka Prefecture Name (285) Daikin Industries, Ltd. Representative Yama 1) Minoru Agent 8550den06 (445) 2128&
On date 7 of the amendment order, between the fourth and fifth lines of the second page of the statement of contents of the amendment, "3,
Add a detailed description of the invention. that's all

Claims (2)

[Claims] (1) A bridge circuit (4) connected to the three-phase winding (2) and having a plurality of switching elements (Tra) to (Trc'), each switching element (Tra) of the bridge circuit (4) This is a pulse width modulation control device for an inverter, which applies a three-phase AC voltage to the three-phase winding (2) by pulse width modulating the DC by ON/OFF operation of ~(Trc'), O of each of the above switching elements (Tra) to (Trc') at a calculation period according to
a calculation means (10) for calculating N time;
Each switching element (Tra) to (T
a dividing means (11) for dividing the ON time of rc') into a plurality of pulses, and a plurality of pulses divided by the dividing means (11) to divide the ON time of each of the switching elements (Tra) to (Tr).
1. A pulse width modulation control device for an inverter, characterized in that it comprises a control means (12) for controlling ON of the pulse width modulation control device (c'). (2) The dividing means (11) includes each switching element (Tr
The pulse width modulation control device for an inverter according to claim 1, wherein the ON time of a) to (Trc') is equally divided into a plurality of preset fixed pulses.