JP7335258B2 - Motor drive device and air conditioner - Google Patents

Description

TECHNICAL FIELD The present invention relates to a motor drive device for driving a motor, and an air conditioner provided with the motor drive device.

Patent Document 1 below describes a technology for improving the efficiency of an air conditioner, in which the connection of a stator winding (hereinafter simply referred to as "winding") of a motor is switched between star connection and delta connection. A configuration of a motor drive is disclosed. This patent document 1 proposes a technique for expanding the high-speed rotation range and improving efficiency in the low-speed rotation range by using star connection in the low-speed rotation range and delta connection in the high-speed rotation range. Hereinafter, this type of motor will be referred to as a "connection switching motor" as appropriate.

Japanese Patent No. 4619826

The leakage current is caused by switching control of the switching elements of the inverter circuit, and the high-frequency component of the current flows through the stray capacitance between the compressor and the ground. Leakage current varies depending on the impedance within the compressor. In particular, oil remains in the compressor when the air conditioner is started. Therefore, the impedance of the compressor becomes smaller than when no oil remains in the compressor. This results in a large leakage current, especially during start-up.

When operating an air conditioner provided with a connection switching motor, it is basically started with a delta connection. Delta connections have lower impedance than star connections. Therefore, starting with a delta connection means being placed in a more severe environment with respect to leakage current, and it is necessary to suppress an increase in leakage current during startup.

SUMMARY OF THE INVENTION It is an object of the present invention to provide a motor drive device capable of suppressing an increase in leakage current during start-up when driving a connection switching motor.

In order to solve the above-described problems and achieve the object, a motor drive device according to the present invention has at least one first switching element, converts an AC voltage output from an AC power supply into a DC voltage, and boosts the voltage. A booster circuit and a capacitor for smoothing the DC voltage output from the booster circuit are provided. The motor drive device also includes an inverter circuit that has a plurality of second switching elements, converts the power accumulated in the capacitor into AC power, and supplies the AC power to the motor. The motor driving device reduces the number of times of switching for the second switching element from the prescribed number of times of switching during a period from the start of self-restraint operation of the motor until the number of revolutions of the motor becomes constant.

ADVANTAGE OF THE INVENTION According to this invention, it is effective in the ability to suppress the increase in the leakage current at the time of starting when driving a connection switching motor.

1 is a diagram showing a configuration example of an air conditioner according to an embodiment; FIG. 1 is a circuit diagram showing the configuration of a motor drive device according to an embodiment; A circuit diagram showing a detailed configuration of the inverter circuit shown in FIG. FIG. 3 is a circuit diagram showing a configuration of the inverter circuit shown in FIG. 2 different from that shown in FIG. FIG. 4 is a diagram for explaining another example of the first and second connection states in the first embodiment; FIG. 5 is a diagram for explaining the relationship between the connection state and the efficiency in the connection switching motor of the embodiment; FIG. 4 is a diagram for explaining operation modes of the motor drive device according to the embodiment; FIG. 4 is a diagram showing one current path in passive synchronous rectification mode in the booster circuit of the embodiment; Schematic diagram showing current-loss characteristics in a typical switching element FIG. 11 is a diagram showing one current path in a simple switching mode in the booster circuit of the embodiment; FIG. 4 is a diagram for explaining output voltage control in the booster circuit according to the embodiment; Time chart showing an example of the method of operating the air conditioner according to the embodiment Flowchart for explaining leakage current suppression control in the embodiment FIG. 1 is a first diagram for explaining overmodulation in an embodiment; A second diagram for explaining overmodulation in the embodiment FIG. 2 is a block diagram showing an example of a hardware configuration that implements functions of a control unit according to an embodiment; FIG. 4 is a block diagram showing another example of the hardware configuration that implements the functions of the control unit according to the embodiment;

A motor drive device and an air conditioner according to embodiments of the present invention will be described below with reference to the accompanying drawings. It should be noted that the present invention is not limited by the embodiments shown below. Also, hereinafter, the electrical connection will be simply referred to as "connection".

Embodiment.
FIG. 1 is a diagram showing a configuration example of an air conditioner 200 according to an embodiment. FIG. 1 shows the configuration of a separate type air conditioner as an example. Air conditioner 200 according to the embodiment includes compressor 251 incorporating motor 500 , four-way valve 259 , outdoor heat exchanger 252 , expansion valve 261 , and indoor heat exchanger 257 . These components are connected via refrigerant pipes 262 to form a refrigeration cycle. Motor 500 is driven by motor drive device 100 according to an embodiment, which will be described later.

A compression mechanism 250 that compresses refrigerant and a motor 500 that operates the compression mechanism 250 are provided inside the compressor 251 . Refrigerant circulates between the compressor 251 and the outdoor heat exchanger 252 and between the compressor 251 and the indoor heat exchanger 257 to form a refrigeration cycle for cooling and heating. Note that the configuration shown in FIG. 15 can be applied not only to air conditioners but also to refrigeration cycle devices such as refrigerators and freezers having a refrigeration cycle.

FIG. 2 is a circuit diagram showing the configuration of the motor drive device 100 according to the embodiment. FIG. 3 is a circuit diagram showing a detailed configuration of inverter circuit 18 shown in FIG.

As shown in FIG. 2, the motor drive device 100 according to the embodiment converts AC power supplied from a single-phase AC power supply 1 into DC power. Further, the motor drive device 100 converts the converted DC power into AC power again and supplies the converted AC power to the motor 500 to drive the motor 500 . Motor 500 is a motor incorporated in compressor 251 of air conditioner 200 shown in FIG. The motor 500 is a motor having a structure in which connection switching is possible, as shown in FIG. In this specification, it is appropriately called a "connection switching motor". A detailed configuration of the motor 500 will be described later.

2, the motor driving device 100 includes a booster circuit 3, a capacitor 4, a control section 10, an inverter circuit 18 having a current detector 18S, a connection switching section 60, and a voltage detector as a first voltage detector. It comprises a detector 5 and a voltage detector 7 which is a second voltage detector.

The booster circuit 3 is a booster converter that converts an AC voltage output from the AC power supply 1 into a DC voltage and boosts the DC voltage by opening and closing a switching element, which will be described later. The booster circuit 3 controls, that is, boosts the voltage value of the converted DC voltage when converting the AC voltage into the DC voltage. The AC voltage output from the AC power supply 1 is called "power supply voltage" and represented by "Vs".

The booster circuit 3 includes a reactor 2 , a first leg 31 and a second leg 32 . The first leg 31 and the second leg 32 are connected in parallel. In the first leg 31, a first upper arm element 311 and a first lower arm element 312 are connected in series. In the second leg 32, a second upper arm element 321 and a second lower arm element 322 are connected in series. One end of reactor 2 is connected to AC power supply 1 . The other end of the reactor 2 is connected to a connection point 3 a between the first upper arm element 311 and the first lower arm element 312 on the first leg 31 . A connection point 3 b between the second upper arm element 321 and the second lower arm element 322 is connected to the other end of the AC power supply 1 . In the booster circuit 3, the connection points 3a and 3b form AC terminals.

The first upper arm element 311 includes a switching element Q1 and a diode D1 connected in antiparallel to the switching element Q1. The first lower arm element 312 includes a switching element Q2 and a diode D2 connected in antiparallel to the switching element Q2. Second upper arm element 321 includes switching element Q3 and diode D3 connected in antiparallel to switching element Q3. The second lower arm element 322 includes a switching element Q4 and a diode D4 connected in antiparallel to the switching element Q4. Each of the switching elements Q1, Q2, Q3, Q4 may be called "first switching element".

In FIG. 2, a metal oxide semiconductor field effect transistor (MOSFET) is illustrated as each of the switching elements Q1, Q2, Q3, and Q4, but it is not limited to a MOSFET. A MOSFET is a switching element that allows current to flow bidirectionally between a drain and a source. Any switching element may be used as long as it is a switching element that allows current to flow bidirectionally between the first terminal corresponding to the drain and the second terminal corresponding to the source.

Anti-parallel means that the first terminal corresponding to the drain of the MOSFET and the cathode of the diode are connected, and the second terminal corresponding to the source of the MOSFET and the anode of the diode are connected. A parasitic diode that the MOSFET itself has inside may be used as the diode. A parasitic diode is also called a body diode.

At least one of the switching elements Q1, Q2, Q3, Q4 is not limited to a MOSFET made of a silicon-based material, and is made of a wide bandgap semiconductor such as silicon carbide, gallium nitride, gallium oxide, or diamond. A MOSFET may also be used.

Wide bandgap semiconductors generally have higher withstand voltage and heat resistance than silicon semiconductors. Therefore, by using a wide bandgap semiconductor for at least one of the switching elements Q1, Q2, Q3, Q4, the voltage resistance and allowable current density of the switching elements are increased, and the semiconductor module incorporating the switching elements can be miniaturized. can be

Three of the switching elements Q1, Q2, Q3 and Q4 may be diodes instead of switching elements. That is, at least one of the switching elements Q1, Q2, Q3, Q4 should be a switching element. Even with such an alternative configuration, a boosting operation, which will be described later, is possible.

One end of the capacitor 4 is connected to the DC bus 12a on the high potential side. The DC bus 12 a is drawn from a connection point 3 c between the first upper arm element 311 on the first leg 31 and the second upper arm element 321 on the second leg 32 . The other end of the capacitor 4 is connected to the DC bus 12b on the low potential side. The DC bus 12b is drawn from a connection point 3d between the first lower arm element 312 on the first leg 31 and the second lower arm element 322 on the second leg 32 . In the booster circuit 3, the connection points 3c and 3d form DC terminals. Further, in the booster circuit 3, the side on which the connection points 3c and 3d are located is called the "DC side".

The output voltage of the booster circuit 3 is applied across the capacitor 4 . Capacitor 4 is connected to DC buses 12 a and 12 b and smoothes the output voltage of booster circuit 3 . The voltage generated on the DC buses 12a and 12b is called "bus voltage" and is represented by "Vdc". Also, an alternating current flowing between the alternating current power supply 1 and the booster circuit 3 may be referred to as a "first current".

The voltage detector 5 detects the power supply voltage Vs and outputs the detected value of the power supply voltage Vs to the control unit 10 .

Voltage detector 7 detects bus voltage Vdc and outputs the detected value of bus voltage Vdc to control unit 10 .

Next, the circuit configuration of the inverter circuit 18 will be described with reference to FIG. As shown in FIG. 3, the inverter circuit 18 has a leg 18A in which an upper arm element 18UP and a lower arm element 18UN are connected in series, and a leg 18B in which an upper arm element 18VP and a lower arm element 18VN are connected in series. and a leg 18C in which the upper arm element 18WP and the lower arm element 18WN are connected in series. Leg 18A, leg 18B and leg 18C are connected in parallel with each other. A shunt resistor 18DS is inserted in the DC bus 12b. Shunt resistor 18DS is a component of current detector 18S shown in FIG. The upper arm elements 18UP, 18VP, 18WP and the lower arm elements 18UN, 18VN, 18WN are sometimes called "second switching elements".

FIG. 3 illustrates a case where the upper arm elements 18UP, 18VP, 18WP and the lower arm elements 18UN, 18VN, 18WN are insulated gate bipolar transistors (IGBTs), but the present invention is not limited to this. A MOSFET may be used instead of the IGBT.

Upper arm element 18UP includes transistor 18a and diode 18b connected in antiparallel to transistor 18a. Other upper arm elements 18VP, 18WP and lower arm elements 18UN, 18VN, 18WN have the same configuration. Anti-parallel means that the anode side of the diode is connected to the first terminal corresponding to the emitter of the IGBT, and the cathode side of the diode is connected to the second terminal corresponding to the collector of the IGBT, as in the booster circuit 3 . means.

Shunt resistor 18 DS detects current flowing between capacitor 4 and inverter circuit 18 . A detected value of the current flowing through the shunt resistor 18 DS is sent to the control section 10 .

Although FIG. 3 shows a configuration including three legs in which the upper arm element and the lower arm element are connected in series, the configuration is not limited to this. The number of legs may be four or more.

When the transistors 18a of the upper arm elements 18UP, 18VP, 18WP and the lower arm elements 18UN, 18VN, 18WN are MOSFETs, at least one of the upper arm elements 18UP, 18VP, 18WP and the lower arm elements 18UN, 18VN, 18WN is , silicon carbide, gallium nitride, gallium oxide, or a wide bandgap semiconductor such as diamond. By using a MOSFET formed of a wide bandgap semiconductor, it is possible to enjoy the effects of withstand voltage and heat resistance.

A connection point 26a between the upper arm element 18UP and the lower arm element 18UN is connected to a first phase (for example, U phase) of the motor 500 (not shown in FIG. 3), and a connection point between the upper arm element 18VP and the lower arm element 18VN. 26b is connected to the second phase (eg, V phase) of the motor 500, and the connection point 26c between the upper arm element 18WP and the lower arm element 18WN is connected to the third phase (eg, W phase) of the motor 500. FIG. In the inverter circuit 18, the connection points 26a, 26b, and 26c constitute AC terminals.

Further, an inverter circuit 18X shown in FIG. 4 may be used instead of the inverter circuit 18 shown in FIG. FIG. 4 is a circuit diagram showing a configuration of the inverter circuit shown in FIG. 2 different from that shown in FIG.

In the inverter circuit 18X shown in FIG. 4, a lower arm shunt resistor 18US connected in series to the lower arm element 18UN is provided between the lower arm element 18UN and the DC bus 12b. Thus, the leg 18A is composed of an upper arm element 18UP, a lower arm element 18UN, and a lower arm shunt resistor 18US connected in series.

Other legs are similar. Specifically, a lower arm shunt resistor 18VS connected in series to the lower arm element 18VN is provided between the lower arm element 18VN and the DC bus 12b. Thus, the leg 18B is composed of an upper arm element 18VP, a lower arm element 18VN and a lower arm shunt resistor 18VS which are connected in series. A lower arm shunt resistor 18WS connected in series to the lower arm element 18WN is provided between the lower arm element 18WN and the DC bus 12b. Thus, the leg 18CB is composed of an upper arm element 18WP, a lower arm element 18WN and a lower arm shunt resistor 18WS which are connected in series.

The lower arm shunt resistor 18US detects the current flowing through the U-phase lower arm. The lower arm shunt resistor 18VS detects the current flowing through the V-phase lower arm. The lower arm shunt resistor 18WS detects the current flowing through the W-phase lower arm. Detected values of the currents flowing through the lower arm shunt resistors 18 US, 18 VS, and 18 WS are sent to the controller 10 .

FIG. 3 shows a circuit configuration of a so-called one-shunt current detection system. The circuit of FIG. 3 is preferably used mainly for a motor for driving a compressor. Further, FIG. 4 shows a circuit configuration of a so-called 3-shunt current detection system. The circuit of FIG. 4 is preferably used mainly for fan drive motors.

Next, motor 500 in Embodiment 1 will be described. Note that the number of revolutions of the motor 500 is appropriately called "motor number of revolutions". Further, the voltage applied to the motor 500 by the inverter circuit 18 is appropriately called "motor applied voltage" or simply "applied voltage".

In FIG. 2, motor 500 includes U-phase winding 502U, V-phase winding 502V, and W-phase winding 502W. A U-phase winding 502U, a V-phase winding 502V, and a W-phase winding 502W are three windings that the motor 500 has.

Both ends of U-phase winding 502U are open. The same applies to the V-phase winding 502V and the W-phase winding 502W. The connection switching unit 60 switches the connection state of the three windings of the motor 500 between a first connection state and a second connection state. The second connection state is a state in which the voltage applied to the motor is lower than that in the first connection state under the same motor rotation speed condition. Therefore, under the condition of the same motor rotation speed, the voltage applied to the motor in the first connection state is higher than the voltage applied to the motor in the second connection state. In the case of the motor 500 of FIG. 2, the first connection state is star connection, and the second connection state is delta connection.

The connection switching unit 60 has a function of switching the connection state of the windings of the motor between the star connection and the delta connection by changing the connection destination of each open winding. To realize this function, the connection switching section 60 includes a U-phase switch 62U, a V-phase switch 62V, and a W-phase switch 62W. U-phase switch 62U is a switching unit that switches the connection destination of U-phase winding 502U. The V-phase switch 62V is a switching unit that switches the connection destination of the V-phase winding 502V. The W-phase switch 62W is a switching unit that switches the connection destination of the W-phase winding 502W.

Contacts of the U-phase switch 62U, the V-phase switch 62V, and the W-phase switch 62W are individually switched by switching signals CS1 to CS3 from the control section .

The current contacts of each phase switch are in a state to connect each phase winding of motor 500 to the star connection. That is, the default contact is a state in which each phase winding of the motor 500 is connected in star connection.

In addition, although each phase switch is described as a c-contact switch in FIG. 2, it is not limited to these examples. Each phase switch may be a switch that can be opened and closed in both directions. For example, each phase switch may be configured by combining an a-contact switch or a b-contact switch. Also, each phase switch may be a semiconductor switch.

Each phase switch preferably has a small conduction loss when turned on, and a mechanical switch such as a relay or a contactor can be used. Further, a switching element formed of a wide bandgap semiconductor may be used instead of the mechanical switch. By using a switching element formed of a wide bandgap semiconductor, it is possible to obtain the effects of low on-resistance, low loss, and low element heat generation.

In addition, in FIG. 2, although the case where the 1st connection state was star connection and the 2nd connection state was delta connection was illustrated, it is not limited to this. For example, the two connection states shown in FIG. 5 may be switched. FIG. 5 is a diagram for explaining another example of the first and second connection states in the first embodiment.

FIG. 5 schematically shows a wiring state of the motor 500 shown in FIG. 2, which is different from that shown in FIG. The upper portion of FIG. 5 shows an example of series connection in which windings of respective phases in star connection are connected in series. The lower portion of FIG. 5 shows an example of parallel connection in which each phase winding in star connection is connected in parallel.

The impedance of U-phase winding 33U1 in series connection is greater than the impedance of U-phase winding 33U2 in parallel connection. Similarly, the impedance of V-phase winding 33V1 in series connection is greater than the impedance of V-phase winding 33V2 in parallel connection, and the impedance of W-phase winding 33W1 in series connection is greater than the impedance of W-phase winding 33W2 in parallel connection. Greater than impedance. Therefore, if the phase currents are the same, the induced voltage induced in each phase winding is greater in series connection than in parallel connection. Therefore, the series connection shown in the upper part of FIG. 5 is a connection state in which the voltage applied to the motor is higher than that in the parallel connection, and corresponds to the first connection state described above. Conversely, the parallel connection shown in the lower part of FIG. 5 is a connection state in which the voltage applied to the motor is lower than that in the series connection, and corresponds to the above-described second connection state.

In FIG. 5, illustration of a component corresponding to the connection switching unit 60 shown in FIG. 2 is omitted. and the second connection state.

Returning to FIG. 2, the control unit 10 generates control signals S311 to S322 for controlling each switching element in the booster circuit 3 based on the detection values of the voltage detectors 5 and 7. FIG. The control signal S311 is a control signal for controlling the switching element Q1, and the control signal S322 is a control signal for controlling the switching element Q4. The switching elements Q2 and Q3 are also controlled by the control signal from the control section 10. FIG. The control signals S311 to S322 generated by the control section 10 are input to a gate drive circuit (not shown) within the booster circuit 3 .

Further, the control unit 10 generates control signals S1 to S6 for controlling each switching element in the inverter circuit 18 based on the detected values of the voltage detector 5, the voltage detector 7 and the current detector 18S. . Control signal S1 is a control signal for controlling upper arm element 18UP, and control signal S6 is a control signal for controlling lower arm element 18WN. Other upper arm elements 18VP, WP and other lower arm elements 18UN, VN are also controlled by control signals from the controller 10. FIG. The control signals S1 to S6 generated by the control section 10 are input to a gate driving circuit (not shown) within the inverter circuit 18 .

Next, the features of the connection switching motor used in the embodiment will be described. In the following description, the operating efficiency of motor 500 is simply referred to as "efficiency". The “efficiency” referred to here is the ratio of the mechanical output of motor 500 to the input power to motor 500 . Moreover, in some drawings, star connection is described as "Y connection", and delta connection is described as "Δ connection".

FIG. 6 is a diagram for explaining the relationship between the connection state and efficiency in the connection switching motor of the embodiment. The horizontal axis indicates the rotation speed of the motor 500, and the vertical axis indicates the efficiency of the motor 500. FIG.

As shown in FIG. 6, the efficiency of the motor 500 when the connection state is the star connection is better than that of the delta connection in the low speed region where the rotation speed is small, that is, in the light load region, but in the high speed region where the rotation speed is high. , that is, it decreases in the high load region or the overload region.

On the other hand, the efficiency of the motor 500 when the connection state is the delta connection is inferior to that of the star connection in the low speed region where the rotation speed is low, but it is improved in the high speed region where the rotation speed is high.

Therefore, the star connection is more efficient than the delta connection in the low speed region, and the delta connection is more efficient than the star connection in the high speed region. Therefore, there are switching points shown in FIG. 6, and if the connection state is switched at these switching points, efficient operation becomes possible. Note that the switching rotation speed at the switching point may be referred to as "first rotation speed".

Here, one of the indexes related to energy saving in air conditioners is annual performance factor (APF). The efficiency of the air conditioner at intermediate load contributes greatly to the APF. It should be noted that the above-described low-speed region or light-load region may be considered to be almost synonymous with intermediate load referred to in APF.

Next, the operation of the main part of the booster circuit 3 will be described with reference to FIGS. 7 to 10 as appropriate.

FIG. 7 is a diagram for explaining operation modes of motor drive device 100 in the embodiment. FIG. 7 shows three modes of operation: passive synchronous rectification mode, simple switching mode and Pulse Width Modulation (PWM) control mode. FIG. 8 is a diagram showing one current path in the passive synchronous rectification mode in the booster circuit 3 of the embodiment. FIG. 9 is a diagram schematically showing current-loss characteristics in a general switching element. FIG. 10 is a diagram showing one current path in the simple switching mode in booster circuit 3 of the embodiment.

The upper part of FIG. 7 shows the power supply voltage and power supply current in the passive synchronous rectification mode. This operation mode is a mode in which synchronous rectification is performed without boosting. Non-boosting means not performing a power supply short-circuit operation. The power supply short-circuit operation will be described later. Synchronous rectification is a control method for turning ON a switching element connected in anti-parallel to a diode in accordance with the timing at which current flows through the diode.

FIG. 8 shows the charging path for the capacitor 4 when the power supply voltage is positive and synchronous rectification is performed. As shown in FIG. 8, it is assumed that the polarity of the power supply voltage is positive when the upper terminal of the AC power supply 1 is at a positive potential. It is also assumed that the polarity of the power supply voltage is negative when the upper terminal of the AC power supply 1 is at a negative potential.

In FIG. 8, when the capacitor 4 is charged by the current supplied from the AC power supply 1, when the switching elements Q1 and Q4 are not turned on, the AC power supply 1, the reactor 2, the diode D1, the capacitor 4, the diode D4, the AC power supply Current flows in the order of 1. A diode does not conduct unless a voltage corresponding to the voltage drop is applied in the direction of current flow, that is, in the forward direction. Therefore, as shown in the upper part of FIG. 7, current flows in a period T2 shorter than the half cycle T1 in the period of the positive half cycle T1 of the power supply voltage. In the passive synchronous rectification mode, during the period T2, the switching elements Q1 and Q4 are controlled to be ON in accordance with the conduction timing of the diodes D1 and D4. Therefore, in the period T2, current flows through the AC power supply 1, the reactor 2, the switching element Q1, the capacitor 4, the switching element Q4, and the AC power supply 1 in this order.

A similar operation is performed in the half cycle in which the power supply voltage is negative. However, during the period T3 in the negative half cycle of the power supply voltage, the switching elements Q2 and Q3 are controlled to be ON in accordance with the conduction timing of the diodes D2 and D3.

FIG. 9 shows the loss characteristics of the diode and the loss characteristics when the switching element is on. As shown in FIG. 9, in a region A where the current is smaller than the current value I0, the diode loss is greater than the switching element loss. Using this characteristic, the motor driving device 100 can be operated with high efficiency by using synchronous rectification that turns on the switching element connected in anti-parallel to the diode in accordance with the timing at which the current flows through the diode.

The middle part of FIG. 7 shows the power supply voltage and power supply current in the simple switching mode. This operation mode is an operation mode in which the power supply short-circuit operation is performed once or several times during the period of the half cycle of the power supply voltage to cause the booster circuit 3 to perform the boosting operation. It should be noted that in the example of the middle part of FIG. 7, the power supply short-circuit operation is performed once during the period of the half cycle of the power supply voltage.

FIG. 10 shows a short-circuit path of the AC power supply 1 via the reactor 2 when the power supply voltage is positive and synchronous rectification is performed. As shown in FIG. 10, the switching elements Q1 and Q3 are turned ON during the period T4. In this way, current flows through the AC power supply 1, the reactor 2, the switching element Q1, the switching element Q3, and the AC power supply 1 in this order, and electrical energy is accumulated in the reactor 2.

After the period T4, the operation in the passive synchronous rectification mode shown in the upper part of FIG. 7 is performed. Immediately after period T<b>4 , the sum of the voltage of AC power supply 1 and the voltage generated in reactor 2 is applied to booster circuit 3 . Therefore, the diodes D1 and D4 of the booster circuit 3 become conductive. Then, the switching elements Q1 and Q4 are turned ON in synchronization with the conduction timing of the diodes D1 and D4, and the power supply current flows.

In FIG. 10, the switching elements Q1 and Q3 are ON-operated, but instead of this, the switching elements Q2 and Q4 may be ON-operated. In this case, the current flows through the AC power supply 1, the reactor 2, the switching element Q2, the switching element Q4, and the AC power supply 1 in that order.

The same is true for the negative half-cycles, after one or several power supply short-circuits, passive synchronous rectification occurs. In the power supply short-circuit operation, the switching elements Q1 and Q3 may be turned ON, or the switching elements Q2 and Q4 may be turned ON.

The lower part of FIG. 7 shows the power supply voltage and power supply current in the PWM control mode. In this operation mode, a power supply short-circuit operation for accumulating electrical energy in reactor 2 and a charging operation for charging capacitor 4 using the electrical energy accumulated in reactor 2 are alternately repeated. Switching between the power supply short-circuit operation and the charging operation is performed at a high frequency of several kHz to several tens of kHz. As a result, as shown in the lower part of FIG. 7, the power supply current is controlled to a sinusoidal current. In addition, the time for the boosting operation is longer than in the simple switching mode shown in the middle part, and a boosted voltage higher than that in the simple switching mode can be obtained.

The three modes described above are switched according to operating conditions and load conditions. This allows the motor drive device 100 to operate with high efficiency. In addition, this makes it possible to operate the air conditioner 200 with high efficiency.

The air conditioner 200 according to the embodiment is driven by the motor drive device 100 configured and operated as described above. Since the motor driving device 100 has the above characteristics, the operating efficiency of the air conditioner 200 can be improved.

However, the motor 500 incorporated in the compressor 251 of the air conditioner 200 is a connection-switching motor, and the connection-switching motor is started with a low-impedance delta connection at the time of start-up. Become.

Therefore, below, a method for reducing leakage current when the air conditioner 200 is started will be described. It should be noted that the major premise of this method is not to simply reduce the leakage current at the time of startup, but to operate the air conditioner 200 with high efficiency.

FIG. 11 is a diagram for explaining output voltage control in the booster circuit 3 of the embodiment.

FIG. 11 shows the induced voltage when the motor 500 is connected in the star connection and the induced voltage when the motor 500 is connected in the delta connection. The horizontal axis indicates the rotation speed of the motor 500, and the vertical axis indicates various voltages. In FIG. 11, in view of the efficiency characteristics shown in FIG. 6, the connection state of the motor 500 is star connection in the low speed region and delta connection in the high speed region. Moreover, in FIG. 11, it is assumed that the star connection and the delta connection are switched at the first rotation speed shown in FIG.

The induced voltage between terminals in star connection is √3 times the induced voltage between terminals in delta connection. Therefore, changing the connection state from delta connection to star connection is equivalent to multiplying the number of winding turns by √3. Therefore, the slope of the induced voltage with respect to the rotational speed is √3 times the induced voltage of the delta connection.

In FIG. 11, the levels of the rectified voltage and the two boosted voltages, the first voltage and the second voltage, are indicated by dashed lines. The rectified voltage is the output voltage of the booster circuit 3 when the booster circuit 3 is not boosted. In other words, the rectified voltage is the output voltage of the booster circuit 3 without the opening/closing operation of the switching element of the booster circuit 3 .

Here, in the embodiment, two boost modes are defined. One is a boosting mode in which the boosting circuit 3 is operated to boost and output a first voltage. This boost mode is defined as "first boost mode". The other is a boosting mode in which the boosting circuit 3 performs a boosting operation to output a second voltage. This boost mode is defined as "second boost mode".

In the first boosting mode, the boosting circuit 3 operates in the above-described simple switching mode to generate the first voltage as shown in FIG. The first voltage is the output voltage of the booster circuit 3 boosted by the opening/closing operation of the switching element of the booster circuit 3 .

In the second boosting mode, the boosting circuit 3 operates in the PWM control mode described above to generate the second voltage as shown in FIG. The second voltage is the output voltage of the booster circuit 3 boosted by the switching operation of the switching element of the booster circuit 3, and is a voltage higher than the first voltage. Note that if the level difference between the second voltage and the first voltage is small, the second voltage may be generated in the first boost mode, that is, the simple switching mode.

In a connection-switching motor, it is necessary to pay attention to voltage shortages in both the star connection and the delta connection so as not to cause a voltage shortage. It is possible to increase the induced voltage by increasing the number of turns of the winding. However, the conventional booster circuit has a large loss at the time of boosting, and there are restrictions on increasing the number of turns.

On the other hand, in the motor driving device of the present embodiment, synchronous rectification is performed in the booster circuit 3, so that the loss caused by the conventional diode rectification can be improved. In addition, since synchronous rectification is performed in both boosting modes, the loss due to the boosting operation can be offset by the loss improvement due to the synchronous rectification during the boosting operation. As a result, efficiency can be improved without impairing the effect of increasing the number of turns by the connection switching motor.

FIG. 12 is a time chart showing an example of the operating method of the air conditioner 200 according to the embodiment. In FIG. 12, the initial or default connection of the windings of motor 500 is star connection. Also, in FIG. 12, the thick solid line represents the number of revolutions, and the thick dashed line represents the bus voltage. In FIG. 12, the portion surrounded by the dashed ellipse, that is, the period from time t3 to time t6 is part of the self-restraint operation period by feedback control, and is called "first period" here. The first period can be rephrased as a period during which the number of revolutions of the motor 500 is constant after self-restraint operation is started. In addition, self-restraint operation is operation control performed based on the detection value of the current detector 18S, if it is an embodiment. Instead of the detection value of the current detector 18S, the detection may be performed based on the detection value of the motor current, which is the current flowing through the motor 500, or based on the detection value of the rotational position of the motor 500.

First, at time t1, energization of air conditioner 200 is started, and at time t2, motor 500 is started. Between time t1 and time t2, the connection state of the windings is switched from star connection to delta connection.

During the period from time t2 to time t3, open-loop control, that is, multi-control operation is performed. An example of the multi-regulation operation is V/f control in which a voltage command corresponding to the commanded rotation speed is generated to control the inverter circuit 18 .

Between time t3 and time t6, that is, in the first period, control is performed to suppress an increase in leakage current according to a flowchart described later. This control is hereinafter referred to as "leakage current suppression control".

In the first period, motor 500 is accelerated. Further, at time t4 when future voltage shortage is expected, the first boosting is performed, and the bus voltage is changed to the first voltage. The above-described leakage current suppression control is performed before and after the bus voltage is changed to the first voltage.

Also, at time t5, the voltage is boosted for the second time, and the bus voltage is changed to the second voltage. The first boosting is performed in the first boosting mode, and the second boosting is performed in the second boosting mode. Leakage current suppression control is also performed before and after the bus voltage is changed to the second voltage.

At time t6, the rated load is reached, and control for constant rotation speed is performed between time t6 and time t7. Also, between time t6 and time t7, the control unit 10 of the motor drive device 100 determines whether the absolute value of the temperature difference between the target temperature and the room temperature is less than the threshold. If the temperature difference is less than the threshold, the deceleration operation is performed for restarting. In addition, in the example of FIG. 12, the deceleration operation is started at time t7.

At time t8 between time t7 and time t9, the boosting operation is stopped and the bus voltage becomes the rectified voltage in order to improve efficiency. At time t9, the operation is stopped and the connection state of the windings is switched from delta connection to star connection.

It is restarted at time t10, and motor 500 is accelerated between time t10 and time t12. At time t11 when the voltage shortage is expected, boosting is performed and the bus voltage is changed to the first voltage. At time t12, the engine reaches an intermediate load, and control is performed to keep the rotation speed constant from time t12 to time t13. In addition, between time t12 and time t13, operation is performed with an intermediate load, and boosting to the second voltage is unnecessary.

At time t13, for example, a stop command is input from a remote controller (not shown), and the deceleration operation is started. At time t14 between time t13 and time t15, the boosting operation is stopped and the bus voltage becomes the rectified voltage in order to improve efficiency. The operation is stopped at time t15, and the energization ends at time t16.

In the example of FIG. 12, the leakage current suppression control is not performed during the restart, but this is not intended to prevent the leakage current suppression control during the restart.

For example, when restarting after switching the connection state of the windings to delta connection, leakage current suppression control is performed. Also, even if the windings are star-connected, the leakage current suppression control may be performed in an unusual state such as a large change in the load state.

Leakage current suppression control in the embodiment can be implemented, for example, by a flowchart shown in FIG. FIG. 13 is a flowchart for explaining leakage current suppression control in the embodiment. The processing shown in FIG. 13 can be performed under the control of the control unit 10 during the first period (period from time t3 to time t6) shown in FIG.

First, the motor 500 is started, and the motor 500 is operated in a controlled manner (step S101). The control unit 10 determines whether or not self-restraint driving can be performed (step S102). If the self-restrained operation cannot be performed (step S102, No), the determination process of step S102 is repeated while continuing the other-restrained operation. On the other hand, if the self-restraint operation is possible (step S102, Yes), the other control operation is changed to the self-restraint operation.

Further, the control unit 10 determines whether or not it is necessary to boost the bus voltage (step S104). If boosting of the bus voltage is unnecessary (step S104, No), the determination process of step S104 is repeated while continuing the self-restraint operation. On the other hand, if it is necessary to boost the bus voltage (step S104, Yes), the control unit 10 selects the boost mode (step S105), causes the booster circuit 3 to boost (step S106), and performs overmodulation control. (Step S107).

The control unit 10 determines whether or not the operation of the motor 500, which is the load, is stable (step S108). If the operation of the motor 500 is not stable (step S108, No), the process returns to step S104. Then, the processing from step S104 to step S106 is repeated. On the other hand, if the operation of the motor 500 is stable (step S108, Yes), the process of this flowchart is terminated.

The above is an example of the leakage current suppression control in the air conditioner 200 according to the embodiment. Some operations are supplemented below.

The decision as to whether or not the self-controlled operation can be performed in step S102 can be made based on whether or not voltage detection by the current detector 18S is possible. The determination of whether the load is stable in step S108 can be made based on the number of revolutions, the temperature of the air blown from the indoor unit, the modulation factor, and the like.

Further, the determination of whether or not boosting is necessary in step S104 is a process of predicting the shortage of the bus voltage. Insufficient bus voltage may be determined by comparing the number of revolutions with a threshold value, or by comparing the modulation rate with a threshold value. The "number of revolutions" and the "modulation rate" referred to here are examples of determination indices. Further, whether or not it is less than the threshold or whether or not it is greater than or equal to the threshold is a condition determined by the determination index. Here, this condition is called "first condition". In other words, the determination of whether or not the pressure should be increased in step S104 is an example of determination processing of whether or not the determination index satisfies the first condition.

When it is predicted that the bus voltage will be insufficient, in step S105, the control unit 10 determines whether to select the first boost mode or the second boost mode. When the first boost mode is selected, the booster circuit 3 generates the first voltage in step S106. Further, when the second boost mode is selected, the booster circuit 3 generates the second voltage in step S106.

In the overmodulation control in step S107, the modulation rate is set to a value exceeding 1 when the inverter circuit 18 is operated.

In the flowchart of FIG. 13, overmodulation control is performed during boosting operation, but overmodulation control may be performed during non-boosting operation.

FIG. 14 is a first diagram for explaining overmodulation in the embodiment, and FIG. 15 is a second diagram for explaining overmodulation in the embodiment. FIG. 14 shows an example of non-overmodulation and FIG. 15 shows an example of overmodulation.

The upper part of FIG. 14 shows an example of the voltage command Vm and the carrier used when generating the PWM signal. Voltage command Vm is a voltage command generated based on voltage amplitude command V*, bus voltage Vdc, and modulation factor. Although FIGS. 14 and 15 show the case where the voltage command is a sine wave, it may also be a rectangular wave. The modulation factor is obtained by dividing the voltage amplitude command V* by the bus voltage Vdc, as shown in FIGS. Note that the voltage amplitude command V* is calculated by multiplying various coefficients inside the control unit 10 depending on the method of vector control, the difference in the configuration of the inverter circuit, and the like. Therefore, depending on how it is defined, the meaning of modulation factor=1 may differ.

FIG. 14 is an example of modulation factor=1, and FIG. 15 is an example of modulation factor=1.2. 14 and 15, the waveform of the inverter output voltage applied from the inverter circuit 18 to the motor 500 is shown. The inverter output voltage is a voltage pulse train.

As shown in the upper part of FIG. 15, when the modulation factor exceeds 1, the peak value of the voltage command _Vm becomes larger than the carrier amplitude. Therefore, as shown in the lower part of FIG. 15, in a region A where the voltage command _Vm is larger than the carrier peak, a wide single voltage pulse is generated. Also, in regions B1 and B2 located on both sides of region A, a voltage pulse train is generated in which the pulse width gradually widens as region A is approached. Therefore, when overmodulation with a modulation factor exceeding 1 is performed, the number of switching times of the switching elements in the inverter circuit 18 can be reduced.

Note that the value of the modulation factor is generated inside the control unit 10 or is given to the control unit 10 from the outside of the control unit 10 .

As described above, the leakage current is caused by the high-frequency component of the current caused by switching control of the switching elements of the inverter circuit, which flows through the stray capacitance between the compressor and the ground. As the number of switching times increases, the high-frequency component of the current due to switching control also increases. Therefore, by reducing the number of times of switching, it is possible to reduce leakage current.

Here, the value of the modulation factor has a preferable range determined by the motor characteristics such as the capacity and impedance of the motor 500 . Therefore, it is important to determine the value of the modulation rate based on the effect of reducing the number of switching times and the characteristics of the motor. In any case, leakage current can be reduced by performing overmodulation during boosting operation.

In this embodiment, a method of reducing the number of times of switching by overmodulation control has been described in order to reduce leakage current, but the present invention is not limited to this. Techniques for reducing the number of times of switching include, for example, two-phase modulation and variable carrier, and these techniques may be used. Specifically, during control for reducing leakage current, it is conceivable to perform control to reduce the number of switching times for the switching elements in the inverter circuit from the prescribed number of switching times. Leakage current can also be reduced by such a technique. The prescribed number of switching times is determined by the amplitude and frequency of the voltage command and the amplitude and frequency of the carrier. The prescribed number of switching times can be stored in a memory, which will be described later.

Next, a hardware configuration for realizing the functions of the control unit 10 according to the embodiment will be described with reference to FIGS. 16 and 17. FIG. FIG. 16 is a block diagram showing an example of a hardware configuration that implements the functions of the control unit 10 according to the embodiment. FIG. 17 is a block diagram showing another example of the hardware configuration that implements the functions of the control unit 10 according to the embodiment.

When realizing all or part of the functions of the control unit 10 in the embodiment, as shown in FIG. The configuration may include an interface 304 for inputting and outputting signals.

The processor 300 may be an arithmetic means such as an arithmetic unit, a microprocessor, a microcomputer, a CPU (Central Processing Unit), or a DSP (Digital Signal Processor). The memory 302 includes non-volatile or volatile semiconductor memories such as RAM (Random Access Memory), ROM (Read Only Memory), flash memory, EPROM (Erasable Programmable ROM), EEPROM (registered trademark) (Electrically EPROM), Examples include magnetic discs, flexible discs, optical discs, compact discs, mini discs, and DVDs (Digital Versatile Discs).

The memory 302 stores programs for executing all or part of the functions of the control unit 10 . The processor 300 controls the booster circuit 3 and the inverter circuit 18 by exchanging necessary information via the interface 304 and executing a program stored in the memory 302 .

Also, the processor 300 and memory 302 shown in FIG. 16 may be replaced with a processing circuit 305 as shown in FIG. The processing circuit 305 corresponds to a single circuit, a composite circuit, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination thereof.

It should be noted that the configuration shown in the above embodiment shows an example of the contents of the present invention, and it is possible to combine it with another known technique, and the configuration can be changed without departing from the gist of the present invention. It is also possible to omit or change part of

1 AC power supply 2 Reactor 3 Booster circuit 3a, 3b, 3c, 3d, 26a, 26b, 26c Connection point 4 Capacitor 5, 7 Voltage detector 10 Control unit 12a, 12b DC bus 18, 18X Inverter circuit, 18A, 18B, 18C leg, 18a transistor, 18b, D1, D2, D3, D4 diode, 18DS shunt resistor, 18S current detector, 18UN, 18VN, 18WN lower arm element, 18UP, 18VP, 18WP upper arm element , 18US, 18VS, 18WS lower arm shunt resistor, 31 first leg, 32 second leg, 33U1, 33U2, 502U U-phase winding, 33V1, 33V2, 502V V-phase winding, 33W1, 33W2, 502W W-phase Winding 60 Connection switching unit 62U U-phase switch 62V V-phase switch 62W W-phase switch 100 Motor drive device 200 Air conditioner 250 Compression mechanism 251 Compressor 252 Outdoor heat exchanger 257 Indoor heat Exchanger 259 Four-way valve 261 Expansion valve 262 Refrigerant piping 300 Processor 302 Memory 304 Interface 305 Processing circuit 311 First upper arm element 312 First lower arm element 321 Second upper arm element, 322 second lower arm element, 500 motor, Q1, Q2, Q3, Q4 switching element.

Claims (8)

a booster circuit that has at least one first switching element and converts an AC voltage output from an AC power supply into a DC voltage and boosts it;
a capacitor for smoothing the DC voltage output from the booster circuit;
an inverter circuit having a plurality of second switching elements, converting the power accumulated in the capacitor into AC power and supplying the AC power to the motor;
with
The motor has a plurality of windings,
Both ends of the winding are open, and by changing connection destinations of the both ends, the connection state of the winding can be switched between a first connection state and a second connection state,
The impedance in the first connection state is higher than the impedance in the second connection state,
The motor is started in the second connection state, and after the motor is started, the operation of the motor is switched from the other-controlled operation to the self-controlled operation, and from the start of the self-controlled operation until the rotation speed of the motor becomes constant. During the period of, leakage current suppression control is performed to suppress the leakage current flowing between the equipment containing the motor and the ground, and then the operation of the motor is temporarily stopped, and the connection state of the windings is changed to the third Switching from the connection state of 2 to the first connection state and restarting,
The motor driving device, wherein the leakage current suppression control reduces the number of times of switching to the second switching element from a prescribed number of times of switching. 2. The motor drive device according to claim 1, wherein the number of times of switching is reduced when the connection state of the windings is the second connection state. The motor drive device according to claim 2, wherein the first connection state is star connection and the second connection state is delta connection. When reducing the switching times,
The motor drive device according to claim 2 or 3, wherein the inverter circuit is operated in an overmodulation manner. The motor drive device according to any one of claims 1 to 4 , wherein the number of times of switching is reduced when the number of revolutions of the motor is equal to or greater than a threshold. The motor driving device according to any one of claims 1 to 4 , wherein the number of times of switching is reduced when the modulation factor of the inverter circuit is equal to or greater than a threshold. When the self-restrained operation is performed, it is determined whether or not the bus voltage needs to be boosted, and the determination of whether or not the boost is necessary is made based on the number of revolutions of the motor or the modulation rate when the inverter circuit is operated. A motor driving device according to any one of claims 1 to 6 . An air conditioner comprising the motor drive device according to any one of claims 1 to 7 .

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