JPWO2020066029A1 - Motor drive and air conditioner - Google Patents

Abstract

The motor drive device (100) has at least one first switching element, and is a booster circuit (3) and a booster circuit (3) that convert and boost the AC voltage output from the AC power supply (1) to a DC voltage. An inverter that has a capacitor (4) that smoothes the DC voltage output from the capacitor and a plurality of second switching elements, converts the power stored in the capacitor (4) into AC power, and supplies it to the motor (500). The circuit (18) is provided. The motor drive device (100) determines the number of times of switching to the second switching element with respect to the specified number of times of switching in the period from the start of the self-control operation of the motor (500) to the constant rotation speed of the motor (500). To reduce.

Description

The present invention relates to a motor drive device for driving a motor and an air conditioner including the motor drive device.

In Patent Document 1 below, as a technique for improving efficiency in an air conditioner, the connection of a stator winding (hereinafter, simply referred to as "winding") of a motor is switched between a star connection and a delta connection. The configuration of the motor drive device is disclosed. Patent Document 1 proposes a technique for expanding a high-speed rotation range and improving efficiency in a low-speed rotation range by making a star connection in a low-speed rotation range and a delta connection in a high-speed rotation range. Hereinafter, this type of motor is appropriately referred to as a "connection switching motor".

Japanese Patent No. 4619826

Leakage current is a high-frequency component of the current that flows through the stray capacitance between the compressor and ground due to switching control of the switching element of the inverter circuit. The leakage current changes according to the impedance in the compressor. In particular, when the air conditioner is started, oil remains in the compressor. Therefore, the impedance of the compressor is smaller than that in the state where no oil remains in the compressor. For this reason, the leakage current becomes particularly large during startup.

When operating an air conditioner equipped with a connection switching motor, it is basically started by delta connection. Delta connections have lower impedance than star connections. Therefore, starting with a delta connection means that the environment is more severe with respect to the leakage current, and it is necessary to suppress an increase in the leakage current during startup.

The present invention has been made in view of the above, and an object of the present invention is to obtain a motor driving device capable of suppressing an increase in leakage current during startup when driving a connection switching motor.

In order to solve the above-mentioned problems and achieve the object, the motor drive device according to the present invention has at least one first switching element, and converts and boosts the AC voltage output from the AC power supply to the DC voltage. It includes a booster circuit and a capacitor that smoothes the DC voltage output from the booster circuit. Further, the motor drive device includes a plurality of second switching elements, and includes an inverter circuit that converts the electric power stored in the capacitor into AC power and supplies it to the motor. The motor drive device reduces the number of times of switching to the second switching element with respect to the specified number of times of switching in the period from the start of the self-control operation of the motor to the constant rotation speed of the motor.

According to the present invention, it is possible to suppress an increase in leakage current during startup when driving a connection switching motor.

The figure which shows the structural example of the air conditioner which concerns on embodiment A circuit diagram showing a configuration of a motor drive device according to an embodiment. A circuit diagram showing a detailed configuration of the inverter circuit shown in FIG. A circuit diagram showing a configuration different from that of FIG. 3 of the inverter circuit shown in FIG. The figure which is provided for the explanation of another example of the 1st and 2nd connection state in Embodiment 1. The figure provided for explaining the relationship between the connection state and efficiency in the connection switching motor of the embodiment. The figure provided for the description of the operation mode of the motor drive device in an embodiment. The figure which shows one of the current paths in the passive synchronous rectification mode in the step-up circuit of embodiment. The figure which shows typically the current-loss characteristic in a general switching element. The figure which shows one of the current paths in the simple switching mode in the step-up circuit of an embodiment. The figure which provides the explanation of the output voltage control in the booster circuit of an embodiment. A time chart showing an example of an operation method of the air conditioner according to the embodiment. Flow chart used to explain the leakage current suppression control in the embodiment The first figure which provides the explanation of the overmodulation in an embodiment. The second figure which provides the explanation of the hypermodulation in embodiment A block diagram showing an example of a hardware configuration that embodies the functions of the control unit in the embodiment. A block diagram showing another example of a hardware configuration that embodies the function of the control unit in the embodiment.

The motor drive device and the air conditioner according to the embodiment of the present invention will be described below with reference to the accompanying drawings. The present invention is not limited to the embodiments shown below. Further, in the following, the electrical connection will be described simply as "connection".

Embodiment.
FIG. 1 is a diagram showing a configuration example of the air conditioner 200 according to the embodiment. FIG. 1 shows the configuration of a separate type air conditioner as an example. The air conditioner 200 according to the embodiment includes a compressor 251 having a built-in motor 500, a four-way valve 259, an outdoor heat exchanger 252, an expansion valve 261 and an indoor heat exchanger 257. These components are connected via a refrigerant pipe 262 to form a refrigeration cycle. The motor 500 is driven by the motor drive device 100 according to the embodiment described later.

Inside the compressor 251 is provided a compression mechanism 250 for compressing the refrigerant and a motor 500 for operating the compression mechanism 250. A refrigeration cycle is configured in which the refrigerant circulates between the compressor 251 and the outdoor heat exchanger 252 and between the compressor 251 and the indoor heat exchanger 257 to perform heating and cooling. The configuration shown in FIG. 15 can be applied not only to an air conditioner but also to a refrigerating cycle apparatus including a refrigerating cycle such as a refrigerator and a freezer.

FIG. 2 is a circuit diagram showing the configuration of the motor drive device 100 according to the embodiment. FIG. 3 is a circuit diagram showing a detailed configuration of the inverter circuit 18 shown in FIG.

As shown in FIG. 2, the motor drive device 100 according to the embodiment converts the AC power supplied from the single-phase AC power supply 1 into DC power. Further, the motor drive device 100 converts the converted DC power into AC power again, supplies the converted AC power to the motor 500, and drives the motor 500. The motor 500 is a motor built in the compressor 251 of the air conditioner 200 shown in FIG. The motor 500 is a motor having a structure capable of switching connections as shown in FIG. In the present specification, it is appropriately referred to as a "connection switching motor". The detailed configuration of the motor 500 will be described later.

In FIG. 2, the motor drive device 100 includes a booster circuit 3, a capacitor 4, a control unit 10, an inverter circuit 18 including a current detector 18S, a connection switching unit 60, and a voltage which is a first voltage detector. It includes a detector 5 and a voltage detector 7 which is a second voltage detector.

The booster circuit 3 is a boost converter that converts an AC voltage output from the AC power supply 1 into a DC voltage and boosts the voltage by opening and closing the switching element described later. When the AC voltage is converted into a DC voltage, the booster circuit 3 controls, that is, boosts the voltage value of the converted DC voltage. The AC voltage output from the AC power supply 1 is called "power supply voltage" and is represented by "Vs".

The booster circuit 3 includes a reactor 2, a first leg 31, and a second leg 32. The first leg 31 and the second leg 32 are connected in parallel. In the first leg 31, the first upper arm element 311 and the first lower arm element 312 are connected in series. In the second leg 32, the second upper arm element 321 and the second lower arm element 322 are connected in series. One end of the reactor 2 is connected to the AC power supply 1. The other end of the reactor 2 is connected to a connection point 3a between the first upper arm element 311 and the first lower arm element 312 in the first leg 31. The connection point 3b between the second upper arm element 321 and the second lower arm element 322 is connected to the other end of the AC power supply 1. In the booster circuit 3, the connection points 3a and 3b form an AC terminal.

The first upper arm element 311 includes a switching element Q1 and a diode D1 connected in antiparallel to the switching element Q1. The first lower arm element 312 includes a switching element Q2 and a diode D2 connected in antiparallel to the switching element Q2. The second upper arm element 321 includes a switching element Q3 and a diode D3 connected in antiparallel to the switching element Q3. The second lower arm element 322 includes a switching element Q4 and a diode D4 connected in antiparallel to the switching element Q4. In addition, each of the switching elements Q1, Q2, Q3, and Q4 may be referred to as a "first switching element".

In FIG. 2, metal oxide semiconductor field effect transistors (Metal Oxide Semiconductor Field Effect Transistors: MOSFETs) are illustrated for each of the switching elements Q1, Q2, Q3, and Q4, but are not limited to MOSFETs. A MOSFET is a switching element capable of passing a current in both directions between a drain and a source. Any switching element may be used as long as it is a switching element capable of bidirectionally flowing a current between the first terminal corresponding to the drain and the second terminal corresponding to the source.

Further, antiparallel means that the first terminal corresponding to the drain of the MOSFET and the cathode of the diode are connected, and the second terminal corresponding to the source of the MOSFET and the anode of the diode are connected. As the diode, a parasitic diode that the MOSFET itself has inside may be used. Parasitic diodes are also called body diodes.

Further, at least one of the switching elements Q1, Q2, Q3, and Q4 is not limited to the MOSFET formed of the silicon-based material, but is formed of a wide bandgap semiconductor such as silicon carbide, gallium nitride, gallium oxide, or diamond. It may be a MOSFET.

In general, wide bandgap semiconductors have higher withstand voltage and heat resistance than silicon semiconductors. Therefore, by using a wide bandgap semiconductor for at least one of the switching elements Q1, Q2, Q3, and Q4, the withstand voltage resistance and the allowable current density of the switching element are increased, and the semiconductor module incorporating the switching element can be miniaturized. Can be converted.

Note that, three of the switching elements Q1, Q2, Q3, and Q4 may use diodes instead of switching elements. That is, at least one of the switching elements Q1, Q2, Q3, and Q4 may be switching elements. Even with such an alternative configuration, the boosting operation described later is possible.

One end of the capacitor 4 is connected to the DC bus 12a on the high potential side. The DC bus 12a is drawn from the connection point 3c between the first upper arm element 311 in the first leg 31 and the second upper arm element 321 in the second leg 32. The other end of the capacitor 4 is connected to the DC bus 12b on the low potential side. The DC bus 12b is drawn from the connection point 3d between the first lower arm element 312 in the first leg 31 and the second lower arm element 322 in the second leg 32. In the booster circuit 3, the connection points 3c and 3d form a DC terminal. Further, in the booster circuit 3, the side where the connection points 3c and 3d are located is called the "DC side".

The output voltage of the booster circuit 3 is applied to both ends of the capacitor 4. The capacitor 4 is connected to the DC bus 12a and 12b and smoothes the output voltage of the booster circuit 3. The voltage generated on the DC bus 12a and 12b is called "bus voltage" and is represented by "Vdc". Further, the alternating current flowing between the alternating current power supply 1 and the booster circuit 3 may be referred to as a "first current".

The voltage detector 5 detects the power supply voltage Vs and outputs the detected value of the power supply voltage Vs to the control unit 10.

The voltage detector 7 detects the bus voltage Vdc and outputs the detected value of the bus voltage Vdc to the control unit 10.

Next, the circuit configuration of the inverter circuit 18 will be described with reference to FIG. As shown in FIG. 3, the inverter circuit 18 includes a leg 18A in which the upper arm element 18UP and the lower arm element 18UN are connected in series, and a leg 18B in which the upper arm element 18VP and the lower arm element 18VN are connected in series. And a leg 18C in which the upper arm element 18WP and the lower arm element 18WN are connected in series. The legs 18A, 18B and 18C are connected in parallel to each other. A shunt resistor 18DS is inserted in the DC bus 12b. The shunt resistor 18DS is a component of the current detector 18S shown in FIG. The upper arm elements 18UP, 18VP, 18WP and the lower arm elements 18UN, 18VN, 18WN may be referred to as "second switching elements", respectively.

FIG. 3 illustrates a case where the upper arm elements 18UP, 18VP, 18WP and the lower arm elements 18UN, 18VN, 18WN are insulated gate bipolar transistors (Insulated Gate Bipolar Transistors: IGBTs), but the present invention is not limited thereto. MOSFETs may be used instead of the IGBTs.

The upper arm element 18UP includes a transistor 18a and a diode 18b connected in antiparallel to the transistor 18a. The other upper arm elements 18VP and 18WP and the lower arm elements 18UN, 18VN and 18WN have the same configuration. In antiparallel, as in the case of the booster circuit 3, the anode side of the diode is connected to the first terminal corresponding to the emitter of the IGBT, and the cathode side of the diode is connected to the second terminal corresponding to the collector of the IGBT. means.

The shunt resistor 18DS detects the current flowing between the capacitor 4 and the inverter circuit 18. The detected value of the current flowing through the shunt resistor 18DS is sent to the control unit 10.

Note that FIG. 3 has a configuration including three legs in which the upper arm element and the lower arm element are connected in series, but the configuration is not limited to this. The number of legs may be four or more.

When the transistor 18a of the upper arm elements 18UP, 18VP, 18WP and the lower arm elements 18UN, 18VN, 18WN is a MOSFET, at least one of the upper arm elements 18UP, 18VP, 18WP and the lower arm elements 18UN, 18VN, 18WN is , Silicon carbide, gallium nitride, gallium oxide or diamond may be formed of a wide bandgap semiconductor. If a MOSFET formed of a wide bandgap semiconductor is used, the effects of withstand voltage and heat resistance can be enjoyed.

The connection point 26a between the upper arm element 18UP and the lower arm element 18UN is connected to the first phase (for example, U phase) of the motor 500 (not shown in FIG. 3), and the connection point between the upper arm element 18VP and the lower arm element 18VN. The 26b is connected to the second phase (for example, V phase) of the motor 500, and the connection point 26c between the upper arm element 18WP and the lower arm element 18WN is connected to the third phase (for example, W phase) of the motor 500. In the inverter circuit 18, the connection points 26a, 26b, and 26c form an AC terminal.

Further, instead of the inverter circuit 18 shown in FIG. 3, the inverter circuit 18X shown in FIG. 4 may be used. FIG. 4 is a circuit diagram showing a configuration different from that of FIG. 3 of the inverter circuit shown in FIG.

In the inverter circuit 18X shown in FIG. 4, a lower arm shunt resistor 18US connected in series with the lower arm element 18UN is provided between the lower arm element 18UN and the DC bus 12b. As a result, the leg 18A is composed of an upper arm element 18UP, a lower arm element 18UN, and a lower arm shunt resistor 18US connected in series with each other.

The same is true for other legs. Specifically, a lower arm shunt resistor 18VS connected in series with the lower arm element 18VN is provided between the lower arm element 18VN and the DC bus 12b. As a result, the leg 18B is composed of an upper arm element 18VP, a lower arm element 18VN, and a lower arm shunt resistor 18VS connected in series with each other. Further, a lower arm shunt resistor 18WS connected in series with the lower arm element 18WN is provided between the lower arm element 18WN and the DC bus 12b. As a result, the leg 18CB is composed of an upper arm element 18WP, a lower arm element 18WN, and a lower arm shunt resistor 18WS connected in series with each other.

The lower arm shunt resistor 18US detects the current flowing through the U-phase lower arm. The lower arm shunt resistor 18VS detects the current flowing through the lower arm of the V phase. The lower arm shunt resistor 18WS detects the current flowing through the lower arm of the W phase. The detected value of the current flowing through the lower arm shunt resistors 18US, 18VS, 18WS is sent to the control unit 10.

FIG. 3 shows a circuit configuration of a so-called one-shunt current detection method. The circuit of FIG. 3 is preferably used mainly for a motor for driving a compressor. Further, FIG. 4 shows a circuit configuration of a so-called three-shunt current detection method. The circuit of FIG. 4 is preferably used mainly for a motor for driving a fan.

Next, the motor 500 according to the first embodiment will be described. The rotation speed of the motor 500 is appropriately referred to as a "motor rotation speed". Further, the voltage applied to the motor 500 by the inverter circuit 18 is appropriately referred to as "motor applied voltage" or simply "applied voltage".

In FIG. 2, the motor 500 includes a U-phase winding 502U, a V-phase winding 502V, and a W-phase winding 502W. The U-phase winding 502U, the V-phase winding 502V, and the W-phase winding 502W are three windings included in the motor 500.

Both ends of the U-phase winding 502U are open. The same applies to the V-phase winding 502V and the W-phase winding 502W. The connection switching unit 60 switches the connection states of the three windings included in the motor 500 between the first connection state and the second connection state. The second connection state is a state in which the applied voltage of the motor is lower than that of the first connection state under the condition that the motor rotation speed is the same. Therefore, under the same conditions of the motor rotation speed, the motor applied voltage in the first connection state is higher than the motor applied voltage in the second connection state. In the case of the motor 500 of FIG. 2, the first connection state is the state of being connected to the star connection, and the second connection state is the state of being connected to the delta connection.

The connection switching unit 60 has a function of switching the connection state of the windings of the motor between the star connection and the delta connection by changing the connection destinations at both ends of each open winding. In order to realize this function, the connection switching unit 60 includes a U-phase switch 62U, a V-phase switch 62V, and a W-phase switch 62W. The U-phase switch 62U is a switching unit for switching the connection destination of the U-phase winding 502U. The V-phase switch 62V is a switching unit for switching the connection destination of the V-phase winding 502V. The W-phase switch 62W is a switching unit for switching the connection destination of the W-phase winding 502W.

The contacts of the U-phase switch 62U, the V-phase switch 62V, and the W-phase switch 62W are individually switched by the switching signals CS1 to CS3 from the control unit 10.

The current contacts of each phase switch are in a state of connecting each phase winding of the motor 500 to a star connection. That is, the default contact is a state in which each phase winding of the motor 500 is connected to the star connection.

In FIG. 2, each phase switch is described as a c-contact switch, but the present invention is not limited to these examples. Each phase switch may be a switch that can be opened and closed in both directions. For example, each phase switch may be configured by combining an a-contact switch or a b-contact switch. Further, each phase switch may be a semiconductor switch.

It is preferable that each phase switch has a small conduction loss when it is turned on, and a mechanical switch such as a relay or a contactor can be used. Further, instead of the mechanical switch, a switching element formed of a wide bandgap semiconductor may be used. By using a switching element formed of a wide bandgap semiconductor, it is possible to enjoy the effects of low on-resistance, low loss, and low element heat generation.

Note that FIG. 2 illustrates a case where the first connection state is a star connection and the second connection state is a delta connection, but the present invention is not limited to this. For example, the two connection states shown in FIG. 5 may be switched. FIG. 5 is a diagram provided for explaining another example of the first and second connection states in the first embodiment.

FIG. 5 schematically shows a connection state of the motor 500 shown in FIG. 2, which is different from that in FIG. The upper part of FIG. 5 shows an example of series connection in which the windings of each phase in the star connection are connected in series. Further, in the lower part of FIG. 5, an example of parallel connection in which each phase winding in the star connection is connected in parallel is shown.

The impedance of the U-phase winding 33U1 in the series connection is larger than the impedance of the U-phase winding 33U2 in the parallel connection. Similarly, the impedance of the V-phase winding 33V1 in the series connection is larger than the impedance of the V-phase winding 33V2 in the parallel connection, and the impedance of the W-phase winding 33W1 in the series connection is that of the W-phase winding 33W2 in the parallel connection. Greater than impedance. Therefore, for the same phase current, the induced voltage induced in each phase winding is larger in the series connection than in the parallel connection. Therefore, the series connection shown in the upper part of FIG. 5 is a connection state in which the motor applied voltage is higher than the parallel connection, and corresponds to the above-mentioned first connection state. On the contrary, the parallel connection shown in the lower part of FIG. 5 is a connection state in which the motor applied voltage is lower than the series connection, and corresponds to the second connection state described above.

Although the configuration portion corresponding to the connection switching unit 60 shown in FIG. 2 is not shown in FIG. 5, the first connection state can be obtained by appropriately combining the a contact switch, the b contact switch, or the c contact switch. And the second connection state can be switched.

Returning to FIG. 2, the control unit 10 generates control signals S311 to S322 for controlling each switching element in the booster circuit 3 based on the detection values of the voltage detector 5 and the voltage detector 7. The control signal S311 is a control signal for controlling the switching element Q1, and the control signal S322 is a control signal for controlling the switching element Q4. The switching elements Q2 and Q3 are also controlled by the control signal from the control unit 10. The control signals S311 to S322 generated by the control unit 10 are input to a gate drive circuit (not shown) in the booster circuit 3.

Further, the control unit 10 generates control signals S1 to S6 for controlling each switching element in the inverter circuit 18 based on the detected values of the voltage detector 5, the voltage detector 7, and the current detector 18S. .. The control signal S1 is a control signal for controlling the upper arm element 18UP, and the control signal S6 is a control signal for controlling the lower arm element 18WN. The other upper arm elements 18VP and WP and the other lower arm elements 18UN and VN are also controlled by the control signal from the control unit 10. The control signals S1 to S6 generated by the control unit 10 are input to a gate drive circuit (not shown) in the inverter circuit 18.

Next, the features of the connection switching motor used in the embodiment will be described. In the following description, the operating efficiency of the motor 500 is simply referred to as "efficiency". The "efficiency" referred to here is the ratio of the mechanical output of the motor 500 to the input power to the motor 500. Further, in some drawings, the star connection is referred to as "Y connection" and the delta connection is referred to as "Δ connection".

FIG. 6 is a diagram for explaining the relationship between the connection state and the efficiency in the connection switching motor of the embodiment. The horizontal axis shows the rotation speed of the motor 500, and the vertical axis shows the efficiency of the motor 500.

As shown in FIG. 6, the efficiency of the motor 500 when the connection state is star connection is better in the low speed region where the rotation speed is small, that is, in the light load region, as compared with the delta connection, but in the high speed region where the rotation speed is large. That is, it decreases in the high load region or the overload region.

On the other hand, the efficiency of the motor 500 when the connection state is delta connection is inferior to that of star connection in the low speed region where the rotation speed is low, but is improved in the high speed region where the rotation speed is high.

Therefore, in the low speed region, the star connection is more efficient than the delta connection, and in the high speed region, the delta connection is more efficient than the star connection. Therefore, there is a switching point shown in FIG. 6, and if the connection state is switched at this switching point, efficient operation becomes possible. The switching rotation speed at the switching point may be referred to as a "first rotation speed".

Here, one of the indexes related to energy saving in the air conditioner is the annual energy consumption efficiency (APF). The efficiency of the air conditioner at the intermediate load greatly contributes to the APF. The low-speed region or the light load region described above may be considered to be substantially synonymous with the intermediate load referred to in APF.

Next, the operation of the main part of the booster circuit 3 will be described with reference to the drawings of FIGS. 7 to 10 as appropriate.

FIG. 7 is a diagram for explaining the operation mode of the motor drive device 100 according to the embodiment. FIG. 7 shows three operation modes: a passive synchronous rectification mode, a simple switching mode, and a pulse width modulation (PWM) control mode. FIG. 8 is a diagram showing one of the current paths in the passive synchronous rectification mode in the booster circuit 3 of the embodiment. FIG. 9 is a diagram schematically showing a current-loss characteristic in a general switching element. FIG. 10 is a diagram showing one of the current paths in the simple switching mode in the booster circuit 3 of the embodiment.

The upper part of FIG. 7 shows the power supply voltage and the power supply current in the passive synchronous rectification mode. This operation mode is a mode in which synchronous rectification is performed without boosting. Non-boosting means that the power supply short-circuit operation is not performed. The power short-circuit operation will be described later. Further, synchronous rectification is a control method in which a switching element connected in antiparallel to a diode is turned on at the timing when a current flows through the diode.

FIG. 8 shows a charging path for the capacitor 4 when the power supply voltage is positive and synchronous rectification is performed. As shown in FIG. 8, it is assumed that the polarity of the power supply voltage is positive when the upper terminal of the AC power supply 1 has a positive potential. Further, it is assumed that the polarity of the power supply voltage is negative when the upper terminal of the AC power supply 1 has a negative potential.

In FIG. 8, when the capacitor 4 is charged by the current supplied from the AC power supply 1, when the switching elements Q1 and Q4 are not turned on, the AC power supply 1, the reactor 2, the diode D1, the capacitor 4, the diode D4, and the AC power supply are used. Current flows in the order of 1. The diode does not conduct unless a voltage corresponding to the voltage drop is applied in the direction in which the current flows, that is, in the forward direction. Therefore, as shown in the upper part of FIG. 7, the current flows in the period T2 in which the power supply voltage is positive in the half cycle T1 and shorter than the half cycle T1. In the passive synchronous rectification mode, the switching elements Q1 and Q4 are controlled to be ON in the period T2 according to the conduction timing of the diodes D1 and D4. Therefore, in the period T2, the current flows in the order of the AC power supply 1, the reactor 2, the switching element Q1, the capacitor 4, the switching element Q4, and the AC power supply 1.

The same operation is performed in the half cycle when the power supply voltage is negative. However, in the period T3 in the half cycle in which the power supply voltage is negative, the switching elements Q2 and Q3 are controlled to be ON according to the conduction timing of the diodes D2 and D3.

FIG. 9 shows the loss characteristic of the diode and the loss characteristic when the switching element is turned on. As shown in FIG. 9, in the region A where the current is smaller than the current value I0, the loss of the diode is larger than the loss of the switching element. By utilizing this characteristic and using synchronous rectification that turns on the switching element connected to the diode in antiparallel to the timing when the current flows through the diode, the motor drive device 100 can be operated with high efficiency.

Further, in the middle part of FIG. 7, the power supply voltage and the power supply current in the simple switching mode are shown. This operation mode is an operation mode in which the booster circuit 3 is boosted by performing one or several power short-circuit operations during a half-cycle period of the power supply voltage. In the example of the middle part of FIG. 7, the power supply short-circuit operation is performed once in the half-cycle period of the power supply voltage.

FIG. 10 shows a short-circuit path of the AC power supply 1 via the reactor 2 when the power supply voltage is positive and synchronous rectification is performed. As shown in FIG. 10, the switching elements Q1 and Q3 are turned on during the period T4. In this way, the current flows in the order of the AC power supply 1, the reactor 2, the switching element Q1, the switching element Q3, and the AC power supply 1, and the electric energy is stored in the reactor 2.

After the period T4, the operation is performed in the passive synchronous rectification mode shown in the upper part of FIG. Immediately after the period T4, the sum of the voltage of the AC power supply 1 and the voltage generated in the reactor 2 is applied to the booster circuit 3. Therefore, the diodes D1 and D4 of the booster circuit 3 are conductive. Then, the switching elements Q1 and Q4 are turned on in accordance with the conduction timing of the diodes D1 and D4, and the power supply current flows.

In FIG. 10, the switching elements Q1 and Q3 are turned on, but instead of this, the switching elements Q2 and Q4 may be turned on. In this case, the current flows in the order of the AC power supply 1, the reactor 2, the switching element Q2, the switching element Q4, and the AC power supply 1.

The same applies to the negative half cycle, and after one or several power short-circuit operations, the passive synchronous rectification operation is performed. In the power short-circuit operation, the switching elements Q1 and Q3 may be turned on, or the switching elements Q2 and Q4 may be turned on.

Further, the lower part of FIG. 7 shows the power supply voltage and the power supply current in the PWM control mode. In this operation mode, a power short-circuit operation in which electric energy is stored in the reactor 2 and a charging operation in which the capacitor 4 is charged by using the electric energy stored in the reactor 2 are alternately repeated. Switching between the power short-circuit operation and the charging operation is performed at a high frequency of several kHz to several tens of kHz. As a result, as shown in the lower part of FIG. 7, the power supply current is controlled to a sinusoidal current. Further, the boosting operation time is longer than that of the simple switching mode shown in the middle stage, and a higher boosting voltage can be obtained than that of the simple switching mode.

The above-mentioned three modes can be switched according to the operating conditions and the load conditions. This makes it possible to operate the motor drive device 100 with high efficiency. Further, this makes it possible to operate the air conditioner 200 with high efficiency.

The air conditioner 200 according to the embodiment is driven by a motor driving device 100 configured and operating as described above. Since the motor drive device 100 has the above-mentioned characteristics, the operating efficiency of the air conditioner 200 can be improved.

However, the motor 500 built in the compressor 251 of the air conditioner 200 is a connection switching motor, and the connection switching motor is started with a delta connection having a low impedance at the time of starting, so that the above-mentioned leakage current is large. Become.

Therefore, a method for reducing the leakage current at the time of starting the air conditioner 200 will be described below. It is a major premise that this method is performed while operating the air conditioner 200 with high efficiency, rather than simply reducing the leakage current at the time of starting.

FIG. 11 is a diagram for explaining output voltage control in the booster circuit 3 of the embodiment.

FIG. 11 shows an induced voltage when the connection state of the motor 500 is star connection and an induced voltage when the connection state of the motor 500 is delta connection. The horizontal axis shows the rotation speed of the motor 500, and the vertical axis shows various voltages. In FIG. 11, in view of the efficiency characteristics shown in FIG. 6, the connection state of the motor 500 is a star connection in the low speed region and a delta connection in the high speed region. Further, in FIG. 11, it is assumed that the star connection and the delta connection are switched at the first rotation speed shown in FIG.

The induced voltage between the terminals in the star connection is √3 times the induced voltage between the terminals in the delta connection. Therefore, changing the connection state from delta connection to star connection is equivalent to multiplying the number of winding turns by √3. Therefore, the slope of the induced voltage with respect to the rotation speed is √3 times the induced voltage of the delta connection.

In FIG. 11, the rectified voltage and the levels of the first voltage and the second voltage, which are the two boost voltages, are shown by broken lines. The rectified voltage is the output voltage of the booster circuit 3 when the booster circuit 3 is not boosted. In other words, the rectified voltage is the output voltage of the booster circuit 3 that does not involve the opening / closing operation of the switching element of the booster circuit 3.

Here, in the embodiment, two boost modes are defined. One is a boosting mode in which the boosting circuit 3 is boosted to output a first voltage. This boost mode is defined as a "first boost mode". The other is a boost mode in which the boost circuit 3 is boosted to output a second voltage. This boost mode is defined as a "second boost mode".

In the first boost mode, the boost circuit 3 operates in the simple switching mode described above and generates a first voltage as shown in FIG. The first voltage is the output voltage of the booster circuit 3 which is boosted by the opening / closing operation of the switching element of the booster circuit 3.

Further, in the second boost mode, the boost circuit 3 operates in the PWM control mode described above, and generates a second voltage as shown in FIG. The second voltage is the output voltage of the booster circuit 3 boosted by the opening / closing operation of the switching element of the booster circuit 3, and is higher than the first voltage. When the level difference between the second voltage and the first voltage is small, the second voltage may be generated in the first boost mode, that is, the simple switching mode.

In the connection switching motor, it is necessary to pay attention to the voltage shortage of both sides so that the voltage shortage in the star connection and the voltage shortage in the delta connection do not occur. It is possible to increase the induced voltage by increasing the number of turns of the winding. However, the conventional booster circuit has a large loss at the time of boosting, and there is a limitation in increasing the number of turns.

On the other hand, in the motor drive device of the present embodiment, since synchronous rectification is performed in the booster circuit 3, the loss generated by the conventional diode rectification can be improved. Further, since synchronous rectification is performed even in the two boost modes, the loss due to the boost operation can be compensated for by the loss improvement due to the synchronous rectification during the boost operation. This makes it possible to improve the efficiency without impairing the effect of increasing the number of turns by the connection switching motor.

FIG. 12 is a time chart showing an example of the operation method of the air conditioner 200 according to the embodiment. In FIG. 12, the initial connection state of the winding of the motor 500, that is, the default connection state is the star connection. Further, in FIG. 12, the thick solid line represents the rotation speed, and the thick broken line represents the bus voltage. Further, in FIG. 12, the portion surrounded by the broken line ellipse, that is, the period from the time t3 to the time t6 is a part of the self-control operation period by the feedback control, and is referred to here as the “first period”. The first period can be rephrased as a period in which the rotation speed of the motor 500 becomes constant from the start of self-control operation. In the embodiment, the self-control operation is an operation control performed based on the detection value of the current detector 18S. Instead of the detection value of the current detector 18S, it may be performed based on the detection value of the motor current, which is the current flowing through the motor 500, or may be performed based on the detection value of the rotation position of the motor 500.

First, at time t1, energization of the air conditioner 200 is started, and at time t2, the motor 500 is started. The connection state of the winding is switched from the star connection to the delta connection between the time t1 and the time t2.

From time t2 to time t3, open-loop control is performed. An example of other control operation is V / f control that controls the inverter circuit 18 by generating a voltage command corresponding to the command rotation speed.

Between time t3 and time t6, that is, in the first period, control is performed to suppress an increase in leakage current according to a flowchart described later. Hereinafter, this control is referred to as "leakage current suppression control".

In the first period, the motor 500 is accelerated. Further, at the time t4 when the voltage shortage is expected in the future, the first boost is performed and the bus voltage is changed to the first voltage. Even before and after the bus voltage is changed to the first voltage, the above-mentioned leakage current suppression control is carried out.

Further, at time t5, the boost is performed for the second time, and the bus voltage is changed to the second voltage. The first boost is performed in the first boost mode and the second boost is performed in the second boost mode. Leakage current suppression control is also implemented before and after the bus voltage is changed to the second voltage.

At time t6, the rated load is reached, and control at a constant rotation speed is performed between time t6 and time t7. Further, between the time t6 and the time t7, the control unit 10 of the motor drive device 100 determines whether or not the absolute value of the temperature difference between the target temperature and the room temperature is less than the threshold value. If the temperature difference is less than the threshold value, the deceleration operation is started to restart. In the example of FIG. 12, the deceleration operation is started at time t7.

At time t8 between time t7 and time t9, the boosting operation is stopped and the bus voltage becomes the rectified voltage in order to improve efficiency. At time t9, the operation is stopped, and the connection state of the winding is switched from the delta connection to the star connection.

It is restarted at time t10, and the motor 500 accelerates between time t10 and time t12. At time t11 when a voltage shortage is expected, boosting is performed and the bus voltage is changed to the first voltage. At time t12, the intermediate load is reached, and control with a constant rotation speed is performed between time t12 and time t13. It should be noted that the operation of the intermediate load is performed between the time t12 and the time t13, and the boosting to the second voltage is unnecessary.

At time t13, for example, a stop command is input from a remote controller (not shown), and the deceleration operation is started. At time t14 between time t13 and time t15, the boosting operation is stopped and the bus voltage becomes the rectified voltage in order to improve efficiency. The operation is stopped at time t15, and the energization ends at time t16.

In the example of FIG. 12, the leakage current suppression control is not executed during the restart, but it does not mean that the leakage current suppression control is not prevented during the restart.

For example, when restarting after switching the connection state of the winding to delta connection, leakage current suppression control is performed. Further, even if the connection state of the winding is a star connection, leakage current suppression control may be performed in a state different from the usual state such as a large fluctuation in the load state.

The leakage current suppression control in the embodiment can be carried out, for example, by the flowchart shown in FIG. FIG. 13 is a flowchart provided for explaining the leakage current suppression control in the embodiment. The process shown in FIG. 13 can be performed under the control of the control unit 10 in the first period (the period from time t3 to time t6) shown in FIG.

First, the motor 500 is started, and the other control operation for the motor 500 is carried out (step S101). The control unit 10 determines whether or not self-control driving can be performed (step S102). If the self-control operation cannot be performed (steps S102, No), the determination process of step S102 is repeated while continuing the other control operation. On the other hand, if self-control driving can be carried out (step S102, Yes), the mode shifts from other control operation to self-control operation.

Further, the control unit 10 determines whether or not it is necessary to boost the bus voltage (step S104). If it is not necessary to boost the bus voltage (steps S104, No), the determination process of step S104 is repeated while continuing the self-control operation. On the other hand, when it is necessary to boost the bus voltage (step S104, Yes), the control unit 10 selects the boost mode (step S105), and while the boost circuit 3 is boosted (step S106), overmodulation control is performed. (Step S107).

The control unit 10 determines whether or not the operation of the motor 500, which is a load, is stable (step S108), and if the operation of the motor 500 is not stable (steps S108, No), returns to step S104. Then, the processes of steps S104 to S106 are repeated. On the other hand, when the operation of the motor 500 is stable (step S108, Yes), the process of this flowchart ends.

The above is an example of leakage current suppression control in the air conditioner 200 according to the embodiment. The following is a supplement to some operations.

The determination as to whether or not the self-control operation is possible in step S102 can be performed depending on whether or not the voltage can be detected by the current detector 18S. The determination of load stability in step S108 can be made based on the rotation speed, the temperature of the blown air of the indoor unit, the modulation rate, and the like.

Further, the determination of the necessity of boosting in step S104 is a process of predicting the shortage of the bus voltage. For the lack of bus voltage, the number of revolutions may be compared with the threshold value, or the modulation factor may be compared with the threshold value. The "rotation speed" and "modulation rate" referred to here are examples of determination indexes. Further, whether or not it is below the threshold value or whether or not it is above the threshold value is a condition determined by the determination index. Here, this condition is referred to as a "first condition". That is, the determination of the necessity of boosting in step S104 is an example of the determination process of whether or not the determination index satisfies the first condition.

When it is predicted that the bus voltage will be insufficient, in step S105, the control unit 10 determines whether to select the first boost mode or the second boost mode. When the first boost mode is selected, in step S106, the boost circuit 3 generates the first voltage. When the second boost mode is selected, the boost circuit 3 generates a second voltage in step S106.

In the overmodulation control in step S107, the modulation factor when operating the inverter circuit 18 is set to a value exceeding 1.

In the flowchart of FIG. 13, overmodulation control is performed during the boosting operation, but overmodulation control may be performed during the non-boosting operation.

FIG. 14 is a first diagram for explaining overmodulation in the embodiment, and FIG. 15 is a second diagram for explaining overmodulation in the embodiment. FIG. 14 shows an example of non-overmodulation, and FIG. 15 shows an example of overmodulation.

In the upper part of FIG. 14, a voltage command Vm and an example of a carrier used when generating a PWM signal are shown. The voltage command Vm is a voltage command generated based on the voltage amplitude command V *, the bus voltage Vdc, and the modulation factor. 14 and 15 show the case where the voltage command is a sine wave, but it may be a rectangular wave. The modulation factor is determined by dividing the voltage amplitude command V * by the bus voltage Vdc, as shown in FIGS. 14 and 15. The voltage amplitude command V * is calculated by multiplying various coefficients inside the control unit 10 depending on the vector control method, the difference in the configuration of the inverter circuit, and the like. Therefore, the meaning of modulation factor = 1 may differ depending on the definition method.

FIG. 14 is an example of modulation factor = 1, and FIG. 15 is an example of modulation factor = 1.2. Further, in the lower part of FIGS. 14 and 15, the waveform of the inverter output voltage applied from the inverter circuit 18 to the motor 500 is shown. The inverter output voltage is a voltage pulse train.

As shown in the upper part of FIG. 15, when the modulation factor exceeds 1, the peak value of the voltage command V _m becomes larger than the carrier amplitude. Therefore, as shown in the lower part of FIG. 15, a wide and single voltage pulse is generated in the region A where the voltage command V _m is larger than the peak of the carrier. Further, in the regions B1 and B2 located on both sides of the region A, voltage pulse trains in which the pulse width gradually increases as the region A approaches are generated. Therefore, when the modulation factor exceeds 1, the number of switchings of the switching element in the inverter circuit 18 can be reduced.

The value of the modulation factor is generated inside the control unit 10 or is given to the control unit 10 from the outside of the control unit 10.

As described above, the leakage current is a high-frequency component of the current controlled by switching to the switching element of the inverter circuit, which flows through the stray capacitance between the compressor and the ground. As the number of switchings increases, so does the high-frequency component of the current due to switching control. Therefore, by reducing the number of switchings, it is possible to reduce the leakage current.

Here, the value of the modulation factor has a preferable range determined by the characteristics of the motor such as the capacitance and impedance of the motor 500. Therefore, it is important that the value of the modulation factor is determined by the effect of reducing the number of switchings and the characteristics of the motor. In any case, it is possible to reduce the leakage current by performing overmodulation during the boosting operation.

In the present embodiment, a method of reducing the number of switchings by overmodulation control has been described in order to reduce the leakage current, but the present invention is not limited to this. Methods for reducing the number of switchings include, for example, two-phase modulation and carrier variable, and these methods may be used. Specifically, at the time of control for reducing the leakage current, it is conceivable to perform control to reduce the number of switchings for the switching element in the inverter circuit with respect to the specified number of switchings. Even with such a method, it is possible to reduce the leakage current. The specified number of switchings is a number determined by the amplitude and frequency of the voltage command, the amplitude and frequency of the carrier. The specified number of switchings can be stored in a memory described later.

Next, the hardware configuration for realizing the function of the control unit 10 in the embodiment will be described with reference to the drawings of FIGS. 16 and 17. FIG. 16 is a block diagram showing an example of a hardware configuration that embodies the function of the control unit 10 in the embodiment. FIG. 17 is a block diagram showing another example of a hardware configuration that embodies the function of the control unit 10 in the embodiment.

When all or a part of the functions of the control unit 10 in the embodiment are realized, as shown in FIG. 16, a processor 300 that performs an operation, a memory 302 that stores a program read by the processor 300, and a memory 302 that stores a program read by the processor 300, and It can be configured to include an interface 304 that inputs and outputs signals.

The processor 300 may be an arithmetic unit, a microprocessor, a microcomputer, a CPU (Central Processing Unit), or a DSP (Digital Signal Processor). Further, the memory 302 includes a non-volatile or volatile semiconductor memory such as a RAM (Random Access Memory), a ROM (Read Only Memory), a flash memory, an EPROM (Erasable Program ROM), or an EEPROM (registered trademark) (Electrically EPROM). Examples thereof include magnetic disks, flexible disks, optical discs, compact disks, mini disks, and DVDs (Digital Versaille Discs).

The memory 302 stores a program that executes all or a part of the functions of the control unit 10. The processor 300 sends and receives necessary information via the interface 304, and the processor 300 executes a program stored in the memory 302 to control the booster circuit 3 and the inverter circuit 18.

Further, the processor 300 and the memory 302 shown in FIG. 16 may be replaced with the processing circuit 305 as shown in FIG. The processing circuit 305 corresponds to a single circuit, a composite circuit, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination thereof.

The configuration shown in the above-described embodiment shows an example of the content of the present invention, can be combined with another known technique, and is configured without departing from the gist of the present invention. It is also possible to omit or change a part of.

1 AC power supply, 2 reactors, 3 booster circuits, 3a, 3b, 3c, 3d, 26a, 26b, 26c connection points, 4 capacitors, 5,7 voltage detectors, 10 controls, 12a, 12b DC bus, 18, 18X Inverter circuit, 18A, 18B, 18C leg, 18a transistor, 18b, D1, D2, D3, D4 diode, 18DS shunt resistance, 18S current detector, 18UN, 18VN, 18WN lower arm element, 18UP, 18VP, 18WP upper arm element , 18US, 18VS, 18WS Lower arm shunt resistance, 31 1st leg, 32 2nd leg, 33U1, 33U2, 502U U-phase winding, 33V1, 33V2, 502V V-phase winding, 33W1, 33W2, 502W W phase Winding, 60 connection changeover, 62U U-phase switch, 62V V-phase switch, 62W W-phase switch, 100 motor drive, 200 air conditioner, 250 compression mechanism, 251 compressor, 252 outdoor heat exchanger, 257 indoor heat Exchanger, 259 four-way valve, 261 expansion valve, 262 refrigerant pipe, 300 processor, 302 memory, 304 interface, 305 processing circuit, 311 first upper arm element, 312 first lower arm element, 321 second upper arm Element, 322 second lower arm element, 500 motor, Q1, Q2, Q3, Q4 switching element.

Claims (7)

A booster circuit that has at least one first switching element and converts and boosts the AC voltage output from the AC power supply to a DC voltage.
A capacitor that smoothes the DC voltage output from the booster circuit,
An inverter circuit having a plurality of second switching elements, converting the electric power stored in the capacitor into AC power and supplying it to the motor.
With
A motor drive device that reduces the number of times of switching for the second switching element with respect to a specified number of times of switching in the period from the start of self-control operation of the motor to the constant rotation speed of the motor. The motor has multiple windings
Both ends of the winding are open, and by changing the connection destinations at both ends, the connection state of the winding can be switched between the first connection state and the second connection state.
The impedance in the first connection state is higher than the impedance in the second connection state.
The motor drive device according to claim 1, wherein the number of times of switching is reduced when the connection state of the winding is the second connection state. When reducing the number of switchings
The motor drive device according to claim 1 or 2, wherein the inverter circuit is overmodulated. The motor drive device according to claim 3, wherein when the inverter circuit performs an overmodulation operation, the voltage applied to the motor becomes larger than the output voltage of the booster circuit. The motor drive device according to any one of claims 1 to 4, which reduces the number of switching times when the rotation speed of the motor is equal to or higher than a threshold value. The motor drive device according to any one of claims 1 to 4, which reduces the number of switchings when the modulation factor of the inverter circuit is equal to or higher than a threshold value. An air conditioner including the motor drive device according to any one of claims 1 to 6.

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