JP6984399B2 - Power converter controller - Google Patents

Description

The present invention relates to a power converter control device.

Conventionally, there is known a technique for reducing a direct current component contained in a three-phase alternating current energized from a power converter to a load. For example, Patent Document 1 describes in an inverter device that "when the three-phase AC voltage becomes unbalanced, a direct current flows through the load and the transformer or motor becomes demagnetized. In this case, the direct current component is superimposed on the three-phase alternating current. It becomes a waveform. " Therefore, in the embodiment shown in FIG. 4 of Patent Document 1, each phase current is input to the integrator, and if the output of the integrator contains a DC component, each phase is removed so as to remove the DC component. The voltage command is biased and corrected.

Further, the control device of the multi-phase rotary machine disclosed in FIGS. 1 and 2 of Patent Document 2 multiplies the current of each phase by the gain K corresponding to the virtual resistance of the stator and negatively feeds back to the voltage command of each phase. However, the DC current component is reduced by increasing the apparent stator resistance in a controlled manner. Further, the polyphase rotation control device disclosed in FIGS. 3 and 4 of Patent Document 2 detects a direct current component by using a direct current component detecting means such as a low-pass filter, and outputs a direct current component by the direct current component detecting means. Multiply only by the virtual resistance gain K.

Japanese Unexamined Patent Publication No. 9-117154 Japanese Unexamined Patent Publication No. 2003-88192

In the prior art of Patent Document 1, the cause of the generation of the DC component is the imbalance of the power converter or the control unit. In the prior art of Patent Document 1, since the current for one cycle of the electric angle is integrated by using an integrator in order to extract the DC component, there is a problem that the responsiveness at the time of transient is deteriorated.

In the prior art of Patent Document 2, the cause of the DC component is the variation of the on-resistance of the switching element and the variation of the inverter DC intermediate voltage. In the prior art of Patent Document 2, since a virtual resistor is used, the voltage applied to the load decreases, and the voltage utilization rate decreases. Further, in the configuration of FIG. 4 of Patent Document 2, in order to extract the DC component, a filter is provided to process the current for one cycle of the electric angle, so that there is a problem that the responsiveness at the time of transition is lowered.

Further, if an initial current flows through the inductance of the circuit during the transient response when the output of the power converter changes, a DC offset current is generated and an imbalance of the three-phase current occurs. However, neither of Patent Documents 1 and 2 mentions the DC offset current.

The present invention has been created in view of these points, and an object of the present invention is to provide a power converter control device capable of suppressing a three-phase current imbalance generated by a DC offset current during a transient response to a high response. There is something in it.

The present invention is a power converter control device that converts DC power into three-phase AC power and controls the operation of the power converter (60) that supplies the load to the motor (80) having resistance and inductance. This power converter control device includes a basic voltage command value calculation unit (20), a compensation voltage command value calculation unit (40), and a gate signal generation unit (56).

The basic voltage command value calculation unit calculates the dq-axis voltage command value (Vq ^* , Vd ^* ) based on the dq-axis current command value (iq ^* , id ^* ), and further converts the dq-axis voltage command value into three phases. Calculate the basic voltage command value (Vu _b ^* , Vv _b ^*). The compensation voltage command value calculation unit is a three-phase current ^{which is a deviation between the three-phase current command value (iu *} , iv ^* ) obtained by converting the dq-axis current command value into three phases and the three-phase current value (iu, iv). The compensation voltage command value (Vu _offset ^* , Vv _offset ^* ) is calculated based on the deviation (Δiu, Δiv) and the resistance or inductance which is the equipment constant of the motor. The gate signal generation unit generates a gate signal commanded to the power converter based on ^{the voltage command value (Vu *} , Vv ^* ) obtained by adding the basic voltage command value and the compensation voltage command value.

In the present invention, by adding the compensation voltage command value to the basic voltage command value, it is possible to suppress the imbalance of the three-phase current generated by the DC offset current during the transient response of the power converter. At this time, the compensation voltage command value calculation unit can instantly and directly control the DC offset component based on the three-phase current deviation. Therefore, it is possible to improve the responsiveness for reducing the DC component as compared with the prior arts of Patent Documents 1 and 2 that need to process the current for one cycle of the electric angle.

For example if the compensation voltage command value calculating unit uses the resistance value proportional gain of the PD controller that receives the three-phase current deviation, it calculates a compensation voltage command value using the inductance value in the differential gain. This makes it possible to calculate the voltage command value for following the command value fastest.

Further, preferably, the compensation voltage command value calculation unit further calculates the compensation voltage command value based on the three-phase current value. Specifically, if the three-phase current deviation is the same, the compensation voltage command value calculation unit increases the compensation voltage command value as the three-phase current value increases. That is, when the load is low and the three-phase current value is small, the influence of the current imbalance is small, so that the compensation voltage command value is set small with an emphasis on control stability. On the other hand, when the load is high and the three-phase current value is large, the compensation voltage command value is set large with an emphasis on high responsiveness in order to suppress the current imbalance.

The overall block diagram of the motor drive system to which the inverter control device by each embodiment is applied. The figure explaining the three-phase current imbalance which occurs when the torque command fluctuates. The circuit model figure which extracted the single-phase component of a three-phase inverter. The current waveform diagram at the time of the transient response (inverter output change) in the inverter control device of the comparative example. Circuit model diagram separated into alternating current (AC) component and direct current (DC) component. Current waveform diagram during transient response when dead beat control is applied. The control block diagram of the inverter control device according to 1st Embodiment. The control block diagram of the inverter control device according to 2nd Embodiment. The control block diagram of the inverter control device according to 3rd Embodiment. (A), (b) a diagram showing the relationship between the three-phase current and the gain of the compensation voltage calculator, (c) the diagram showing the relationship between the three-phase current and the three-phase current deviation and the compensation voltage command value. The control block diagram of the inverter control device according to 4th Embodiment. FIG. 5 is a control block diagram of a basic voltage command value calculation unit according to a fifth embodiment. The figure which shows the switching of the DC component extraction composition by the motor rotation speed. The control block diagram of the inverter control device of the comparative example.

Hereinafter, a plurality of embodiments of the power converter control device will be described with reference to the drawings. In a plurality of embodiments and comparative examples, substantially the same configurations are designated by the same reference numerals and description thereof will be omitted. The power converter control device of the present embodiment controls the operation of the power converter that converts DC power into three-phase AC power and supplies it to a load having resistance and inductance. In this embodiment, the motor corresponds to the load and the inverter corresponds to the "power converter". Further, the inverter control device corresponds to the "power converter control device".

First, with reference to FIG. 1, the overall configuration of the motor drive system 90 will be described. In the inverter 60, six switching elements 61-66 of the upper and lower arms are bridge-connected. Specifically, the switching elements 61, 62, and 63 are U-phase, V-phase, and W-phase upper arm switching elements, respectively, and the switching elements 64, 65, and 66 are below the U-phase, V-phase, and W-phase, respectively. It is a switching element of the arm. The switching element 61-66 is composed of, for example, an IGBT, and a freewheeling diode that allows a current from the low potential side to the high potential side is connected in parallel.

The smoothing capacitor 15 is provided at the input portion of the inverter 60 and smoothes the voltage of the battery 10. A boost converter may be provided between the battery 10 and the inverter 60. The inverter 60 converts the DC power of the battery 10 into three-phase AC power by operating the switching elements 61-66 according to the gate signals UH, UL, VH, VL, WH, and WL commanded by the inverter control device 50. .. Then, the inverter 60 applies the three-phase voltages Vu, Vv, and Vw to the phase windings 81, 82, and 83 of the motor 80.

The motor 80 is, for example, a permanent magnet type synchronous three-phase AC motor, and is typically a motor generator capable of power running and regenerative operation. A current sensor for detecting the phase current value is provided in the current path connected to the two-phase winding of the three-phase windings 81, 82, and 83 of the motor 80. In the example of FIG. 1, current sensors 71 and 72 for detecting the U-phase current value iu and the V-phase current value iv are provided in the current paths connected to the U-phase winding 81 and the V-phase winding 82, respectively. .. The rotation angle sensor 85 is a rotation angle sensor such as a resolver, and detects the electric angle θ of the motor 80. Further, the angular velocity in which the electric angle θ is time-differentiated by the differentiator 86 is converted, and the rotation speed ω is calculated.

The inverter control device 50 calculates a voltage command value to be applied to the motor 80 based on a torque command from a higher-level ECU (not shown) and feedback information from the current sensors 71 and 72 and the rotation angle sensor 85. Then, the gate signals UH, UL, VH, VL, WH, and WL are generated based on the calculated voltage command value, and the gate signals are commanded to the switching elements 61-66 of the inverter 60.

Here, FIG. 14 shows a configuration of a well-known inverter control device 509 that performs dq-axis current feedback control as a comparative example. The configuration related to the dq-axis current feedback control includes the other-phase current calculation unit 31, the three-phase / dq conversion unit 32, the dq-axis current deviation calculation unit 36, the controllers 371 and 372, and the like. The other-phase current calculation unit 31 calculates the current value of the other one phase (for example, iw) from the current value of two of the three phases (for example, iu, iv) using Kirchhoff's law.

The three-phase / dq conversion unit 32 dq-converts the three-phase current values iu, iv, and iwa into the dq-axis current values iq and id based on the electric angle θ. Hereinafter, according to the order of the top and bottom in the figure, the q-axis current and the like are described first as the description order of the symbols such as the dq-axis current value. The dq-axis current deviation calculation unit 36 calculates the dq-axis current deviations Δiq and Δid, which are deviations between the ^{dq-axis current command values iq *} and id ^{* and the dq-axis current values iq and id.} The controllers 371 and 372 calculate the ^{q-axis voltage command value Vq *} and the d-axis voltage command value Vd ^* , respectively, by PI control or the like.

Further, in the comparative example of FIG. 14, a non-interference control unit 38 that calculates a non-interference term so as to compensate the interference voltage between the dq axes based on the dq axis current values iq and id is provided. The non-interference term calculated by the non-interference control unit 38 is added to the outputs of the controllers 371 and 372 by the non-interference term addition unit 39. It should be noted that the distinction between the symbols of ^{the dq-axis voltage command values Vq *} and Vd ^* before and after the addition of the non-interference term is omitted.

The dq / 3-phase conversion unit 24 converts the dq-axis voltage command values Vq ^* and Vd ^* into three phases based on the electric angle θ, and calculates the U-phase and V-phase voltage command values Vu ^* and Vv ^{*. The other-phase voltage calculation unit 55 calculates the voltage command value flow value Vw *} of the other one phase, the W phase, from the U phase, the V phase voltage command value Vu ^* , and Vv ^*. The PWM signal generation unit 56 generates a ^{gate signal by PWM control based on the three-phase voltage command values Vu *} , Vv ^* , and Vw ^* , and outputs the gate signal to the inverter 60.

In the inverter control device 509 of the comparative example, when the voltage command value fluctuates with the fluctuation of the torque command, an imbalance of the three-phase current occurs. Refer to FIG. 2 for this three-phase current imbalance. As for the three-phase current, the solid line, the broken line, and the alternate long and short dash line indicate the current of each phase. Before the time tx, the current command is constant at i1, and the three-phase current is balanced. When the torque command fluctuates at time tx and the current command changes from i1 to i2, an imbalance occurs in the three-phase current, and an overcurrent occurs in one phase shown by the solid line. As a result, the inverter 60 may be damaged or torque ripple may occur. In addition, reactive current is also generated in the battery current on the DC side, and the life of the battery 10 and the smoothing capacitor 15 may be shortened due to heat generation.

The causes of the three-phase imbalance and their solutions will be described with reference to FIGS. 3 to 6. FIG. 3 shows a circuit model in which a single-phase component of a three-phase inverter is extracted by taking the U phase as an example. Vu represented by an AC symbol is an inverter output, and eu is an induced voltage. The U-phase current iu flows through the resistor R and the inductance L of the circuit. When the initial current flows through the inductance L, a voltage VL opposite to the U-phase current iu is generated. At this time, the inverter output Vu is represented by the equation (1). The symbol P in the mathematical formula is a differential operator. In addition, regarding the notation of subscripts of symbols in the following mathematical formulas, the part corresponding to the subscripts may be described in normal characters in the specification or in the figure depending on the legibility.

Here, as shown in FIG. 4, it is assumed that the inverter output Vu changes. Va is the amplitude of Vu. When the U-phase current iu is 0 at tx when the inverter output Vu changes, the center of the current amplitude remains 0 even after the time tx. On the other hand, when the U-phase current iu is not 0 at tx when the inverter output Vu changes, an offset current ΔIdc is generated in the negative direction in the example of FIG. 4, and the center of the current amplitude is offset from 0.

Next, FIG. 5 shows a circuit model in which the U-phase current iu is separated into an alternating current (AC) component iu_ac and a direct current (DC) component iu_dc. The equation (1) of the inverter output Vu is separated into the equation (2.1) of the AC voltage Vu and the equation (2.2) of the DC offset voltage Vdc.

The AC component iu_ac of the U-phase current changes in a sinusoidal shape. Further, in the configuration of the comparative example, which is a well-known technique, the DC component iu_dc is attenuated according to the LR time constant as shown in the equation (3) with the offset current −ΔIdc as the initial value as shown by the solid line.

As described above, when the initial current flows through the inductance L at the time of transient, a DC offset current is generated. That is, in a three-phase inverter, a DC offset current is always generated at the time of transient, so that a three-phase current imbalance occurs. Therefore, we focus on the point that the DC offset current is controlled by the DC offset voltage from the DC component equation (2.2). Then, as shown by the broken line in the circuit model of the DC component in FIG. 5, the DC offset voltage Vdc is determined so that the DC offset current becomes 0 at the next sample timing of tx at the time of change. Control that suppresses the DC offset current in this way is called "dead beat control". The dead beat control is expressed by the equation (4).

FIG. 6 shows a current waveform when a voltage obtained by adding an AC voltage Vu and a DC offset voltage Vdc determined by deadbeat control is applied to the inverter. The two-dot chain line is a current waveform when the dead beat control shown in FIG. 4 is not performed. Even if the U-phase current iu is not 0 at the change tx of the inverter output Vu, the amplitude center of the U-phase current iu coincides with 0 at the next sample timing by performing the dead beat control at the change tx.

In this way, by applying the dead beat control to the DC offset component ΔIdc, it is possible to control the U-phase current iu so as to reach the target value instantly. Therefore, the three-phase current imbalance at the time of transient response due to the DC offset current is suppressed. As a result, it is possible to appropriately prevent damage to the inverter, generation of torque ripple, heat generation of the battery and the smoothing capacitor, and the like due to overcurrent.

The inverter control device 50 of the present embodiment uses the above principle to suppress a three-phase current imbalance during a transient response in which a torque command fluctuates. Subsequently, the configuration and operation of the inverter control device 50 will be described for each embodiment. The code of the inverter control device of each embodiment is assigned the number of the embodiment in the third digit following "50".

(First Embodiment)
The first embodiment will be described with reference to FIG. 7. The inverter control device 501 of the first embodiment includes a basic voltage command value calculation unit 201, a compensation voltage command value calculation unit 401, a PWM signal generation unit 56 as a “gate signal generation unit”, and the like. Here, "1" is added to the reference numerals of the basic voltage command value calculation unit and the compensation voltage command value calculation unit of the first embodiment, respectively, in the third digit following "20" and "40", respectively. As for the reference numerals of the basic voltage command value calculation unit and the compensation voltage command value calculation unit of the following embodiments, when the configurations are substantially the same as those of the above-described embodiment, the reference numerals of the above-described embodiments are used. On the other hand, when the configuration is different from the above-described embodiment, the number of the embodiment is newly added to the third digit following "20" or "40".

Further, in the present embodiment, the current sensors 71 and 72 correspond to the detection of the two-phase current values iu and iv of the U-phase and the V-phase, and the two-phase of the three-phase current values of the U-phase and the V-phase The current value is controlled, and the current value of the other one phase, W phase iw, is calculated using Kirchhoff's law. In other embodiments, any two phases may be used, and the current values of the three phases may be detected and controlled. In the specification, the two-phase current value or voltage value of U phase and V phase are also described by using the terms "three-phase current value" and "three-phase voltage value".

Basic voltage command value calculating section 201 includes a FF control calculation unit 21 and dq / 3-phase conversion unit 24, ^* the basic voltage command values Vu _b by the feed forward control, it calculates the Vv _b ^*. The FF control calculation unit 21 calculates the dq-axis voltage command values Vq ^* and Vd ^* by the voltage equation shown in the equation (5). dq / 3-phase conversion unit 24, dq-axis voltage command value based on the electric angle theta Vq ^*, to convert the Vd ^* 3-phase, calculated U-phase, the basic voltage command values V phase Vu _b ^*, the Vv _b ^* do.

The compensation voltage command value calculation unit 401 includes a dq / 3-phase conversion unit 41, a 3-phase current deviation calculation unit 42, and 3-phase current controllers 431 and 432. The dq / 3-phase conversion unit 41 converts the dq-axis current command values iq ^* and id ^* into three phases based on the electric angle θ, and calculates the three-phase current command values iu ^* and iv ^*. The three-phase current deviation calculation unit 42 calculates ^{the three-phase current deviations Δiu and Δiv, which are deviations between the three-phase current command values iu *} and iv ^* and the three-phase current values iu and iv detected by the current sensors 71 and 72. do.

_{The U-phase current controller 431 and the V-phase current controller 432 calculate the compensation voltage command values Vu offset} ^* and Vv _offset ^* of the U-phase and V-phase based on the current deviations Δiu and Δiv of the U-phase and V-phase, respectively. .. The current deviations Δiu and Δiv correspond to DC offset components at the time of transient response. That is, the technical significance of the compensation voltage command value calculation unit 401 is that the DC offset component is directly controlled by the three-phase current controllers 431 and 432.

And encompasses each phase current deviation .DELTA.i, represents a compensation voltage to encompass command value V _offset ^*, described compensating voltage command value V _offset ^* computation methods. The calculation of the compensation voltage command value V _offset ^* is generally performed by PD control according to the equation (6.1) or P control according to the equation (6.2). Here, Kp is a proportional gain, Kd is a differential gain, and (Δi / Δt) is a time differential value of the three-phase current deviation Δi. The P control equation (6.2) corresponds to the one in which the differential term Kd (Δi / Δt) of the PD control equation (6.1) is regarded as 0.

Further, in the above-mentioned dead beat control, the DC offset voltage Vdc is determined so as to suppress the DC offset current Δidc by the equation (4) including the resistance R and the inductance L, which are the equipment constants of the motor 80. The three-phase current controller 431 and 432 may calculate the _{compensation voltage command value V offset} ^* by the equation (7) obtained by multiplying the right side of the equation (4) by the gain K (K ≦ 1). By using the equipment constant of the motor 80, it is possible to calculate the voltage command value for following the command value as soon as possible.

The equation (7) can be regarded as an equation in which the differential gain Kd in the PD control equation (6.1) is (K × L) and the proportional gain Kp is (K × R). Further, if the differential term of the equation (7) is ignored, the equation of P control with the proportional gain Kp as (K × R) is obtained.

The compensation voltage addition unit 54 adds the basic voltage command values Vu _b ^* and Vv _b ^* and the compensation voltage command values Vu _offset ^* and Vv _offset ^* for each U phase and V phase, and the added voltage command value Vu ^*. , Vv ^* is output. ^{The other-phase voltage calculation unit 55 calculates the voltage command value flow value Vw *} of the other one phase, the W phase, from the U phase, the V phase voltage command value Vu ^* , and Vv ^*.

The PWM signal generation unit 56 generates a ^{gate signal by PWM control based on the three-phase voltage command values Vu *} , Vv ^* , and Vw ^* , and outputs the gate signal to the inverter 60. In another embodiment, the gate signal generation unit is not limited to PWM control, and may generate a gate signal by a pulse pattern method or the like that selects an appropriate pattern from a plurality of preset voltage output patterns.

As described above, in the first embodiment, the AC component is controlled by the feedforward control by the basic voltage command value calculation unit 201, and the DC offset component is controlled by the three-phase current feedback control by the compensation voltage command value calculation unit 401. High response is possible.

(Second Embodiment)
The second embodiment will be described with reference to FIG. In the inverter control device 502 of the second embodiment, the basic voltage command value computing section 202, the basic voltage command values Vu _b ^* by conjugation with a feed-forward control and the dq-axis current feedback control, and calculates the Vv _b ^*. The configuration of the compensation voltage command value calculation unit 401 is the same as that of the first embodiment.

The configuration of the basic voltage command value calculation unit 202 related to feedforward control is the same as that of the basic voltage command value calculation unit 201 of the first embodiment. ^{The FF control calculation unit 21 calculates the feedforward terms Vq *} _ff and Vd ^* _ff of the dq-axis voltage command value by the voltage equation.

The configuration related to the dq-axis current feedback control includes the other-phase current calculation unit 31, the 3-phase / dq conversion unit 32, the low-pass filter (“LPF” in the figure and the following text) 34, the dq-axis current deviation calculation unit 36, and control. Vessels 371, 372 and the like are included. Except for LPF34, the configuration is substantially the same as that of the comparative example of FIG.

The LPF 34 has a high frequency component of a predetermined frequency or higher of the dq axis current values iq and id, and outputs _{the dq axis current values iq LPF and id LPF after the LPF processing.} In other words, the LPF 34 feeds back only the AC component to the dq-axis current command values iq ^* and id ^* , excluding the DC offset components of the dq-axis current values iq * and id. That is, the LPF 34 corresponds to a "DC component extraction unit" that extracts DC components of dq-axis current values iq and id.

The dq-axis current deviation calculation unit 36 calculates dq-axis current deviations Δiq and Δid, which are deviations between the ^{dq-axis current command values iq *} and id ^* and the dq-axis current values iq _LPF and id _{LPF after LPF processing.} The controllers 371 and 372 calculate the feedback term Vq ^* _fb ^{of the q-axis voltage command value and the feedback term Vd *} _fb of the d-axis voltage command value, respectively, by PI control or the like. The voltage command value adding unit 22 adds the feedforward terms Vq ^* _ff and Vd ^* _ff and the feedback terms Vq ^* _fb and Vd ^* _fb of the dq axis voltage command value.

As described above, the basic voltage command value calculation unit 202 of the second embodiment can improve the controllability in the steady state by feeding back only the AC component excluding the DC offset component by the LPF 34. That is, since the dq-axis currents iq and id vibrate when the DC offset current is generated, only the DC component is extracted using a filter, and the dq-axis currents iq and id are controlled to realize a desired torque. Therefore, in the second embodiment, the feedback control of the current excluding the DC offset portion by the compensation voltage command value calculation unit 401 and the feedback control of the DC offset current by the basic voltage command value calculation unit 20 are combined to provide high response and high response. Control without constant deviation can be realized.

(Third Embodiment)
The third embodiment will be described with reference to FIGS. 9 and 10. The compensation voltage command value calculation unit 403 of the inverter control device 503 of the third embodiment has a U-phase current controller 431 and a V-phase current controller 432 with respect to the compensation voltage command value calculation unit 401 of the first and second embodiments. The U-phase current value iu and the V-phase current value iv are input to, respectively, and the other configurations are the same. The U-phase current controller 431 calculates the _{U-phase compensation voltage command value Vu offset} ^* based on the U-phase current value iu in addition to the U-phase current deviation Δiu. The V-phase current controller 432 calculates the _{V-phase compensation voltage command value Vv offset} ^* based on the V-phase current value iv in addition to the V-phase current deviation Δiv.

For example, in the configuration in which the dead beat control is performed by the equation (7), the compensation voltage command value calculation unit 403 has gain K of the three-phase current controllers 431 and 432 according to the U-phase current value iu and the V-phase current value iv. Is changed in the range of 0 to 1. Preferably, the larger the three-phase current values iu and iv are, the larger the gain K is set.

That is, when the load is low and the three-phase current values iu and iv are small, the influence of the current imbalance is small, so the gain K decreases so _{that the compensation voltage command value V offset} ^{* becomes small with an emphasis on control stability.} Will be done. On the other hand, when the load is high and the three-phase current values iu and iv are large, the gain K is increased so _{that the compensation voltage command value V offset} ^{* is increased with an emphasis on high responsiveness in order to suppress the current imbalance.} To.

10 (a) and 10 (b) show the change pattern of the gain K according to the three-phase current values iu and iv. In the example of FIG. 10A, in the region where the three-phase current values iu and iv are less than the saturation current value Isat, the gain K monotonically increases to 1 as the three-phase current values iu and iv increase. The gain K is 1 in the region where the three-phase current values iu and iv are equal to or higher than the saturation current value Isat. Further, in the example of FIG. 10B, the gain K is 0 in the region where the three-phase current values iu and iv are equal to or less than the critical value Icrt. The fact that K is 0 is equivalent to not performing DC offset current suppression control.

Further, in the configuration in which the PD control is carried out according to the equation (6.1), the proportional gain Kp and the differential gain Kd are set larger as the three-phase current values iu and iv are larger, respectively. In the configuration in which P control is performed according to the equation (6.2), the larger the three-phase current values iu and iv are, the larger the proportional gain Kp is set.

Further, the compensation voltage command value V _offset ^* is not limited to the configuration calculated by the mathematical formula using the gain K, and the three-phase current values iu and iv and the three-phase current deviations Δiu and Δiv as shown in FIG. 10 (c) are used. It may be calculated directly by the map whose input is. In this map, the larger the three-phase current values iu and iv, and the larger the three-phase current deviations Δiu and Δiv, the larger the compensation voltage command value V _offset ^* . The characteristic line of the map is not limited to the straight line as shown in FIG. 10 (c), but may be a polygonal line or a curved line.

(Fourth Embodiment)
The fourth embodiment will be described with reference to FIG. Inverter controller 504 in the basic voltage command value calculating portion 204 of the fourth embodiment does not perform the feedforward control, the basic voltage command values Vu _b ^* only by dq-axis current feedback control, and calculates the Vv _b ^*. The configuration of the compensation voltage command value calculation unit 401 is the same as that of the first embodiment.

The configuration related to the dq-axis current feedback control is the same as that of the second and third embodiments. Further, in the example of FIG. 11, as in the comparative example of 14, a non-interference control unit 38 that calculates the non-interference term so as to compensate the interference voltage between the dq axes based on the dq axis current values iq and id is provided. ing. In FIG. 11, the portion Ex surrounded by the frame including the LPF 34 and the dq-axis current deviation calculation unit 36 is a portion quoted in the description of the fifth embodiment shown in FIG. In the fourth embodiment, the LPF34 as the "DC component extraction unit" provides feedback control of the AC component excluding the DC offset component, whereby the controllability in the steady state can be improved.

(Fifth Embodiment)
A fifth embodiment will be described with reference to FIGS. 12 and 13. The basic voltage command value calculation unit 205 of the fifth embodiment differs from the basic voltage command value calculation unit 204 of the fourth embodiment only in the configuration corresponding to the Ex unit of FIG. Since the other configurations are the same as those in FIG. 11, the illustration is omitted. The basic voltage command value calculation unit 205 is provided with a DC component extraction unit 330 including a switching determination unit 33, an LPF 34, and a moving average calculation unit 35.

The LPF 34 performs LPF processing for removing high frequency components having a predetermined frequency or higher of the dq axis current values iq and id as in the fourth embodiment, and outputs the dq axis current values iq _LPF and id _LPF after the LPF processing. The moving average calculation unit 35 performs a moving average process for calculating a moving average in a predetermined electric angle range of the dq axis current values iq and id or at a predetermined time, and outputs _{the dq axis current values id MVAVR} and iq _{MVAVR after the moving average process.} do. The switching determination unit 33 switches between the LPF process and the moving average process as the DC component extraction process according to the rotation speed of the motor.

In the fourth embodiment having only LPF34 as the DC component extraction configuration, the cutoff frequency is fixed. Therefore, in the frequency region where the frequency of the AC component of the dq axis current values iq and id, which is the sixth component of the rotation frequency ω, is equal to or lower than the cutoff frequency, the DC component cannot be extracted, and steady-state control is performed. It becomes unstable. Further, if the cutoff frequency is lowered too much in order to make the LPF34 function in a wide rotation speed region, the convergence after the transient response deteriorates. Therefore, as shown in FIG. 13, in the low rotation speed region where the frequency of the AC component of the dq-axis current values iq and id is equal to or lower than the cutoff frequency of the LPF 34, the switching determination unit 33 switches from the LPF processing to the moving average processing. To determine. The section of the moving average processing is set to a predetermined electric angle range or a predetermined time corresponding to one cycle of, for example, the sixth component of the rotation frequency.

As described above, in the fifth embodiment, in the dq-axis current feedback control of the basic voltage command value calculation unit 205, LPF processing is performed in the high rotation speed region larger than the cutoff frequency and LPF processing is performed in the low rotation speed region according to the rotation speed of the motor. The processing is switched so that the DC component is extracted by the moving average processing. Therefore, it is possible to appropriately balance the control stability in the steady state and the convergence after the transient response.

On more than, the present invention is not intended to be limited to the above embodiments can be implemented in various forms without departing from the scope of the invention.

201, 202, 204, 205 ... Basic voltage command value calculation unit,
401, 403 ... Compensated voltage command value calculation unit,
50 (501-504) ... Inverter control device (power converter control device),
56 ... PWM signal generation unit (gate signal generation unit),
60 ... Inverter (power converter),
80 ... Motor (load).

Claims (4)

A power converter control device that converts DC power into three-phase AC power and controls the operation of the power converter (60) that is supplied to the motor (80), which is a load having resistance and inductance.
dq-axis current command value ^(iq *, ib) dq-axis voltage command value based on ^{(Vq *,} Vd *) is calculated and further the dq-axis voltage command value 3-phase converted basic voltage command values _(Vu ^{b *} , Vv _b ^* ) and the basic voltage command value calculation unit (201, 202, 204, 205),
The three-phase current deviation (Δiu, Δiv), ^{which is the deviation between the three-phase current command value (iu *} , iv ^* ) obtained by converting the dq-axis current command value into three phases and the three-phase current value (iu, iv). The compensation voltage command value calculation unit (401, 403) that calculates the compensation voltage command value _{(Vu offset} ^* , Vv _offset ^* ) based on the resistance or inductance that is the equipment constant of the motor, and
A gate signal generation unit (56) that generates a gate signal to be commanded to the power converter based on ^{a voltage command value (Vu *} , Vv ^* ) obtained by adding the basic voltage command value and the compensation voltage command value.
A power converter controller equipped with. The compensation voltage command value calculation unit (403) is
The power converter control device according to claim 1 , further calculating the compensation voltage command value based on the three-phase current value. The basic voltage command value calculation unit (202, 204, 205) is
It has a DC component extraction unit (34, 330) that extracts a DC component of the dq axis current value (iq, id) dq-converted from the three-phase current value.
The basic voltage command value is obtained by feedback control based on the dq-axis current deviation (Δiq, Δid), which is the deviation between the dq-axis current command value and the dq-axis current value excluding the DC component by the DC component extraction unit. The power converter control device according to claim 1 or 2 for calculation. Before Stories basic voltage command value calculating portion and the DC component extraction unit (205) (330), depending on the rotational speed of the motor, and a low-pass filter processing for removing a predetermined frequency or higher frequency components of the dq-axis current value The power converter control device according to claim 5, wherein the dq-axis current value is switched between a predetermined electric angle range and a moving average process for calculating a moving average in a predetermined time.

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