JP6976645B2 - Characteristic measuring device - Google Patents

Description

The present invention relates to a characteristic measuring device for measuring the characteristics of an object, for example, a characteristic measuring device including a surface acoustic wave (SAW) sensor.

A surface acoustic wave sensor, for example, is used as a characteristic sensor in a characteristic measuring device for measuring the characteristics of a liquid as an object, for example, concentration, viscosity, conductivity, dielectric constant, and the like. A surface acoustic wave sensor is a sensor in which a comb-shaped (IDT: InterDigital Transducer) electrode and surface acoustic wave element are arranged on a piezoelectric substrate such as a crystal (see, for example, Patent Document 1 and the like), and is an object. It is a sensor that measures the characteristics of a liquid by dropping a certain liquid onto a surface acoustic wave sensor, inputting an input signal to the surface acoustic wave sensor, and detecting the amplitude and phase of the output signal from the surface acoustic wave sensor. ..

Specifically, for example, the characteristic measuring device 100 shown in FIG. 6 includes a digital circuit 120 and an analog circuit 130, and the output signal from the oscillator 140, which is the source of the entire device, is used as a clock source in the digital circuit 120. However, in the analog circuit 130, it is used as a reference signal of the PLL circuit (phase-locked loop) 131. The PLL circuit 131 is an oscillator that generates and outputs a local carrier signal used for modulation / demodulation of transmission / reception in the analog circuit 130. That is, it is composed of a phase comparator and a voltage controlled oscillator, and a stable signal having a low frequency is used as a reference signal and used as an oscillation circuit of a high frequency signal. Here, the local frequency of the local carrier signal is 261.5 MHz. The local carrier signal from the PLL circuit 131 is input to the quadrature modulator 134 and the mixer circuit (mixer) 137.

On the other hand, the baseband portion (10 MHz CW Baseband Signal) 121 of the digital circuit 120 converts the 10 MHz complex baseband signal into two values (I value) based on the 10 MHz complex carrier signal generated by the NCO (numerical controlled oscillator) 122. And Q value). Only the 10 MHz signal, which is the fundamental wave, is passed through the first LPF (low-pass filter) 132 and the second LPF 133 of the analog circuit 130, respectively, and the 10 MHz sine wave and the cosine wave are orthogonal to each other. It is input to the modulator 134.

The quadrature modulator 134 transitions the baseband signal to a frequency of 251.5 MHz by multiplying the complex baseband signal of 10 MHz and the local carrier signal, that is, by the difference between the two. The output signal from the orthogonal modulator 134 is power-amplified by the power amplifier 135 and input to the SAW sensor (surface acoustic wave sensor) 150. The SAW sensor 150 is a SAW filter as a device, and is configured and designed to pass only a signal near 251.5 MHz. Therefore, by combining it with a narrow band inspection signal as a sensor, it is possible to suppress noise in a frequency band other than the inspection signal and obtain a high SN ratio (signal-to-noise ratio).

The output signal from the SAW sensor 150 is input to the mixer circuit 137 after being power-amplified by the LNA 136. A 261.5 MHz local carrier signal from the PLL circuit 131 and a 251.5 MHz signal from the LNA 136 are input to the mixer circuit 137, and by multiplying these signals, that is, by the difference between the two, 10 MHz. It is divided into a baseband signal and a harmonic component. As described above, in order to generate the 251.5 MHz signal which is the inspection signal of the SAW sensor 150 and the 10 MHz signal which is the baseband signal in the digital circuit 120, the PLL circuit 131 has a 261.5 MHz local carrier signal. Is generated, and multiplication processing is performed in the mixer circuit 137.

Subsequently, after removing the harmonic component with the third LPF138, the baseband signal is input to the A / D converter 139 for conversion into a digital signal. The sample rate of the A / D converter 139 is usually required to be at least twice the frequency of the input signal according to the Nyquist sampling theorem. That is, when the input signal is 10 MHz, a high sample rate of 20 MHz or more is required. The inspection signal converted into a digital signal by the A / D converter 139 is multiplied by the 10 MHz complex carrier signal generated by the NCO 122 by the multiplication circuits 123 and 124 of the digital circuit 120 to include amplitude information and phase information. However, it is decomposed into a DC component and a harmonic component. Subsequently, the harmonic component is removed by the fourth LPF125 and the fifth LPF126, and the DC component is taken out.

Here, each data (DC component) contains noise, and a sufficient amount of data is stored and stored in the RAM 127 in order to satisfy the required accuracy. After that, the host control unit (CPU) 160 collects the data stored in the RAM 127, performs an averaging process, calculates and acquires the amplitude and phase of the inspection signal that has passed through the SAW sensor 150. Then, the amplitude and phase of the acquired inspection signal are converted into density and various other information according to the type of the SAW sensor 150.

Japanese Unexamined Patent Publication No. 2010-181178

By the way, when commercializing such a characteristic measuring device as a battery-powered portable type, it is required to reduce the circuit scale, reduce power consumption, reduce the size, and reduce the price. For that purpose, it is desirable to mount all the circuit configurations on one chip IC. However, the characteristic measuring device as described above has a complicated circuit configuration and high power consumption, and it is necessary to simplify the circuit configuration and reduce the power consumption.

Therefore, an object of the present invention is to provide a characteristic measuring device capable of simplifying a circuit configuration and reducing power consumption.

In order to solve the above problems, the invention according to claim 1 is an input signal generation means that receives an oscillation signal from an oscillator and generates an input signal based on a reference frequency of the oscillation signal, and receives the input signal. A characteristic sensor that outputs an output signal according to the characteristics of the object to be measured, a mixer means that multiplies the input signal and the output signal, and outputs a baseband signal of a predetermined frequency, and the baseband signal. Is undersampled, converted into a digital signal at a predetermined sample rate, and output, and the digital signal is divided into an in-phase component signal and an orthogonal component signal according to the predetermined sample rate, and the in-phase component signal and the said quadrature component signal and respectively and a quadrature detection means for integrating a predetermined time, while when the time interval of a given sample rate, accuracy by reducing the influence of multiple reflection occurring on the characteristics the sensor is Ru is ensured Therefore, it is a characteristic measuring apparatus characterized in that it is a value obtained by subtracting a quarter of the time of the period of the baseband signal.

According to the present invention, when the input signal generated by the input signal generation means is input to the characteristic sensor, the characteristic sensor outputs an output signal according to the characteristics of the object to be measured. This output signal is multiplied by the input signal by the mixer means, frequency-shifted, undersampled by the conversion means, and converted into a digital signal at a predetermined sample rate. At this time, the time interval of the predetermined sample rate is a value obtained by subtracting a quarter of the period of the baseband signal from the predetermined time for ensuring the accuracy of the characteristic sensor. This value is the characteristic sensor. The sample rate is lower than the frequency of the output signal from. The converted digital signal is divided into an in-phase component signal and an orthogonal component signal according to a predetermined sample rate by the orthogonal detection means, and further, the in-phase component signal and the orthogonal component signal are integrated for a predetermined time.

The invention according to claim 2 is characterized in that, in the characteristic measuring apparatus according to claim 1, the characteristic sensor is a surface acoustic wave sensor that detects a change in the propagation characteristics of a surface acoustic wave of the input signal. And.

According to the invention of claim 1, the output signal from the characteristic sensor is frequency-shifted by the mixer means, undersampled at a sample rate lower than the frequency of the output signal from the characteristic sensor, and converted into a digital signal. The conversion means can be configured by an A / D converter having a low sample rate. Therefore, it becomes possible to reduce the power consumption. Further, since the amplitude and phase of the output signal from the surface acoustic wave sensor can be acquired without providing the orthogonal modulator, NCO, multiplication circuit, etc., the configuration can be simplified. According to the invention of claim 1, the time interval of the predetermined sample rate in the conversion means is the time for which the influence of the multiple reflections generated in the characteristic sensor is reduced and the accuracy is ensured. Since the value is obtained by subtracting one-fourth of the period, the phase is rotated by 90 ° for each burst by the orthogonal detection means, so that it is not necessary to provide a multiplication circuit, and the circuit configuration can be simplified. become.

According to the second aspect of the present invention, the characteristic sensor is composed of a surface acoustic wave sensor that detects a change in the propagation characteristics of the surface acoustic wave of the input signal, so that the in-phase component for a predetermined time is used by the orthogonal detection means. It is possible to calculate and acquire the amplitude and phase of the output signal from the surface acoustic wave sensor based on the signal and the orthogonal component signal. This makes it possible to provide a characteristic measuring device for measuring the concentration, viscosity, conductivity, dielectric constant, etc. of the liquid to be measured at low cost.

It is a block block diagram which shows the outline of the characteristic measuring apparatus 1 which concerns on embodiment of this invention. It is a waveform diagram which shows the waveform of the input signal IS, the output signal OS, and the baseband signal BS of FIG. It is a block block diagram which shows the orthogonal detection circuit 5 of FIG. It is an image diagram which shows the calculation by the orthogonal detection circuit 5 of FIG. It is a figure which shows the calculation result (truth value) of the I value and Q value by the orthogonal detection circuit 5 of FIG. It is a block diagram which shows the characteristic measuring apparatus provided with the conventional surface acoustic wave sensor.

Hereinafter, the present invention will be described based on the illustrated embodiment.

FIG. 1 is a block diagram showing a characteristic measuring device 1 according to an embodiment of the present invention. This characteristic measuring device 1 is a device for measuring the characteristics of the object to be measured, for example, the concentration, viscosity, conductivity, dielectric constant, etc. of the liquid, and is mainly an oscillator 2, an analog circuit 3, and a SAW sensor. It includes a (characteristic sensor) 4, a digital circuit 5, and an upper control unit (CPU) 6.

The oscillator 2 is the source vibration of the entire characteristic measuring device 1, for example, an oscillation signal WS of 10 MHz is output, and the oscillation signal WS is input to the PLL circuit (phase-locked loop) 31 of the analog circuit 3 described later as a reference signal. At the same time, it is used as a clock source of the digital circuit 5. Here, since the oscillator 2 is required to have high accuracy, a crystal oscillator is desirable. Further, as will be described later, it is input to the mixer 32 of the analog circuit 3 and multiplied by the output signal of the PLL circuit 31 to be used for transitioning to a frequency of, for example, 250 MHz.

The analog circuit 3 includes a PLL circuit (input signal generation means) 31, a mixer circuit 32, a power amplifier 33, an LNA (low noise amplifier) 34, a mixer circuit (mixer means) 35, and an LPF (low pass filter). ) 36 and an A / D converter (conversion means) 37.

The PLL circuit 31 is a circuit that generates an input signal IS having a predetermined frequency and inputs it to the SAW sensor 4. That is, it is composed of a phase comparator and a voltage control oscillator, and functions as an oscillation circuit of a high frequency signal using a stable oscillation signal WS with a low frequency input from the oscillator 2 as a reference signal, and an input signal IS of a predetermined frequency ( RF signal), in this embodiment, a 260 MHz input signal IS is generated. This input signal IS is input to the mixer circuit 32.

The mixer circuit 32 is a circuit that transitions the input signal IS to a frequency of 250 MHz by multiplying the input signal IS of 260 MHz and the oscillation signal WS of 10 MHz, that is, by the difference between the two. This input signal IS is input to the power amplifier 33, is amplified by the power amplifier 33, and is input to the SAW sensor 4.

In this embodiment, the SAW sensor 4 is actually a SAW filter as a device, and is configured and designed to pass only a signal near 250 MHz. Therefore, by combining with the input signal IS having a narrow band as a sensor, noise in a frequency band other than the input signal IS can be suppressed and a high SN ratio (signal-to-noise ratio) can be obtained. The frequency of the input signal IS input to the SAW sensor 4 and the output signal OS output from the SAW sensor 4 are the same, but the frequency depends on the characteristics of the object to be measured such as being dropped on the SAW sensor 4. Therefore, the amplitude and phase are different (changed), or a delay occurs. Here, the SAW sensor 4 does not have to be a SAW filter, and may be any one that can convert the characteristics / events to be measured as a sensor into the amplitude, phase, or delay of the electric signal. Further, a reference channel (a path only for passing the input signal IS from the power amplifier 33 to the LNA 34) is provided in parallel with the SAW sensor 4, and the inspection signal is passed or not passed through the SAW sensor 4 by switching the switch. Therefore, the characteristics of the object to be measured may be measured (comparative measurement).

The inspection signal that has passed through the SAW sensor 4 is input to the LNA 34 as an output signal OS of 250 MHz, power is amplified by the LNA 34, and then input to the mixer circuit 35. The mixer circuit 35 has the same configuration as the mixer circuit 32, and is obtained by multiplying the output signal OS of 250 MHz and the input signal IS of 260 MHz output from the PLL circuit 31, that is, by the difference between the two, the output signal OS. Is a circuit that outputs a baseband signal BS by transitioning to a frequency of 10 MHz. This baseband signal BS is input to the LPF36. This LPF36 is a filter that removes harmonic components, and improves the SN ratio by passing only a signal near 10 MHz. Therefore, if the predetermined SN ratio is satisfied, the LPF 36 may not be provided. The baseband signal BS that has passed through the LPF 36 is input to the A / D converter 37.

The A / D converter 37 is an analog-to-digital converter that undersamples a 10 MHz baseband signal BS and converts it into a digital signal DS at a time interval of, for example, a sample rate of 11.975 μs. The A / D converter 37 has a built-in analog driver (amplifier), and the band of the analog driver can input a high frequency signal without distortion or with low distortion depending on the required accuracy.

Here, the reason for setting the time interval of this sample rate to 11.975 μs will be described. An input signal IS transitioned to a frequency of 250 MHz is input to the SAW sensor 4 as a burst signal, an object to be measured is dropped, and the amplitude and phase of the output signal OS change according to the characteristics of the object to be measured. Or there will be a delay. As a result, multiple reflections of burst signals occur in the SAW sensor 4, and the time during which this effect becomes constant or less, that is, the predetermined time during which the accuracy of the SAW sensor 4 is ensured is 12 μs. Further, since the frequency of the baseband signal BS input to the A / D converter 37 is 10 MHz, the period is 0.1 μs. However, as will be described later, the digital signal DS is orthogonally modulated and each sampling is performed. Since it rotates by 1/4 frequency (1 quadrant, 90 °), the period of the baseband signal BS for each quadrant is 1/4 (= 0.025 μs) of 0.1 μs. Therefore, the value of 11.975 μs, which is obtained by subtracting the period of 0.025 μs for each quadrant of the baseband signal BS from the time of 12 μs in which the accuracy of the SAW sensor 4 is ensured, is set as the time interval of the sample rate.

FIG. 2 is a waveform diagram showing waveforms of an input signal IS output from the power amplifier 33 of FIG. 1, an output signal OS input to the LNA 34, and a baseband signal BS input to the A / D converter 37. .. As shown in FIG. 2, when the time interval of the sample rate is 11.975 μs, the timing at which the baseband signal BS is input to the A / D converter 37 (the timing at which the waveform of the baseband signal BS is open up and down). ), So it is possible to perform sampling equivalently. The time interval of this sample rate is 11.975 μs if the relationship between the time when the influence of multiple reflections in the SAW sensor 4 becomes constant or less and the frequency of the baseband signal BS as described above is established. Not limited to.

The digital circuit 5 is a circuit that digitally processes the digital signal DS output from the A / D converter 37 of the analog circuit 3, and includes a 2-bit counter 51 and an orthogonal detection circuit (orthogonal detection means) 7. There is. The 2-bit counter 51 counts at predetermined sample rate time intervals, that is, every 11.975 μs, and controls “0”, “1”, “2”, “3”, “0”, “1”, and so on. It is a counter that repeatedly outputs a signal (count value).

The orthogonal detection circuit 7 divides the digital signal DS output from the A / D converter 37 into an in-phase component signal and an orthogonal component signal every 11.975 μs, and the in-phase component signal and the orthogonal component signal are predetermined respectively. It is a circuit that integrates time. Specifically, in this embodiment, as shown in FIG. 3, the first logic circuit 71, the second logic circuit 72, the first flip-flop 73, and the second flip-flop 74 are provided. I have.

The first logic circuit 71 is a logic circuit that integrates the in-phase component signal of the output signal OS output from the A / D converter 37 and converted into the digital signal DS according to the control signal of the 2-bit counter 51 for a predetermined time. Is. That is, the digital signal DS from the A / D converter 37 and the I value stored and stored in the first flip-flop 73 are input, and as shown in FIG. 4, the control signal of the 2-bit counter 51 is "0". (Initial phase θ ₀ + 0 °), the digital signal DS (x ₀ ) is added to _{the I value (x 0} − x ₂ ) of the first flip-flop 73, and the first flip-flop 73 is combined with the digital signal DS (x 0). Output. When the control signal is "1" (initial phase θ ₀ + 90 °), the I value (x ₀ − x ₂ ) of the first flip-flop 73 is directly output to the first flip-flop 73 for control. When the signal is "2" (initial phase θ ₀ + 180 °), the digital signal DS (x _{2 ) is subtracted from the I value (x 0} − x ₂ ) of the first flip-flop 73 to obtain the first flip-flop 73. When the output is output to the flip-flop 73 and the control signal is "3" (initial phase θ ₀ + 270 °), the I value (x ₀ − x ₂ ) of the first flip-flop 73 is used as it is in the first flip-flop 73. Output to.

Similarly, the second logic circuit 72 integrates the orthogonal component signal of the output signal OS output from the A / D converter 37 and converted into the digital signal DS according to the control signal of the 2-bit counter 51 for a predetermined time. It is a logic circuit. That is, when the digital signal DS from the A / D converter 37 and the Q value stored and stored in the second flip-flop 74 are input and the control signal is “0” (initial phase θ ₀ + 0 °). Outputs the Q value (x ₁ − x ₃ ) of the second flip-flop 74 to the second flip-flop 74 as it is. When the control signal is "1" (initial phase θ ₀ + 90 °), the digital signal DS (x ₁ ) is added to _{the Q value (x 1} − x ₃ ) of the second flip-flop 74. When the output is output to the second flip-flop 74 and the control signal is “2” (initial phase θ _{0 + 180 °), the Q value (x 1} − x ₃ ) of the second flip-flop 74 is used as it is in the second flip-flop 74. When the output is output to the flip-flop 74 and the control signal is “3” (initial phase θ ₀ + 270 °), the digital signal DS (x ₃ ) is _{transmitted from the Q value (x 1} − x ₃ ) of the second flip-flop 74. Is divided and output to the second flip-flop 74.

_{In this way, when the control signal is "0" (initial phase θ 0} + 0 °) according to the control signal (control timing) every 11.975 μs of the sample rate time interval, the digital signal DS is in phase with “+”. When the control signal is "1" (initial phase θ ₀ + 90 °), the digital signal DS is divided into "+" orthogonal component signals, and the control signal is "2" (initial phase θ ₀ +180). In the case of °), the digital signal DS is divided into "-" in-phase component signals, and in the case of the control signal "3" (initial phase θ ₀ + 270 °), the digital signal DS is divided into the orthogonal components of "-". It is divided into signals, and the in-phase component signal and the orthogonal component signal are integrated and accumulated for a predetermined time, respectively.

The first flip-flop 73 and the second flip-flop 74 are logic circuit registers composed of a plurality of bits and holding and storing data (1, 0), and the first flip-flop 73 is a first flip-flop 73. The output value from the logic circuit 71 is stored, and the second flip-flop 74 stores the output value from the second logic circuit 72. Only DC components and complex numbers (DC components of Fourier transform) including amplitude information and phase information are stored and stored in both flip flops 73 and 74, and noise and interference components are suppressed. There is. Further, if the integration time, that is, the number of bits of both flip-flops 73 and 74 is lengthened, the SN ratio deteriorated by undersampling can be improved, and the noise contained in each digital signal (sample) is suppressed to satisfy the required accuracy. It is set to the optimum time.

When the integration time is reached, the I value and the Q value stored and accumulated in both the flip-flops 73 and 74 are input to the upper control unit 6 and cleared to zero. That is, the integration and the dump (Integrate and Dump) are repeated every integration time. Here, since the output values of both flip-flops 73 and 74 have already been averaged, the output values (measurement results) may be transferred to the upper control unit 6 at a relatively low rate.

Due to the output values of the first logic circuit 71 and the second logic circuit 72 described above, the I value of the first flip-flop 73 and the Q value of the second flip-flop 74 are output as shown in FIG. The logic. That is, when the control signal is “0” (initial phase θ ₀ + 0 °), the I value of the first flip-flop 73 is 1, and the Q value of the second flip-flop 74 is 0. When the control signal is "1" (initial phase θ ₀ + 90 °), the I value of the first flip-flop 73 is 0, and the Q value of the second flip-flop 74 is 1, and the control is performed. When the signal is "2" (initial phase θ ₀ + 180 °), the I value of the first flip-flop 73 is -1, the Q value of the second flip-flop 74 is 0, and the control signal is In the case of "3" (initial phase θ ₀ + 270 °), the I value of the first flip-flop 73 is 0, and the Q value of the second flip-flop 74 is -1.

The upper control unit 6 calculates the amplitude and phase of the output signal OS output from the SAW sensor 4 based on the I value and the Q value output from both flip-flops 73 and 74, and determines the characteristics of the object to be measured. It is an arithmetic processing unit to measure. That is, the amplitude and phase of the input signal IS passing through the SAW sensor 4 are converted into concentration and other information according to the type of the SAW sensor 4. Here, the higher-level control unit 6 may be configured as long as it plays the role of such higher-level control, and may be configured by a smartphone (multifunctional mobile phone) or a personal computer. Further, it may be mounted in hardware in the digital circuit 5.

Next, the operation of the characteristic measuring device 1 having such a configuration, the characteristic measuring method, and the like will be described. Here, it is assumed that the measurement target is dropped on the SAW sensor 4 and the SAW sensor 4 is exposed to the measurement target.

First, the 260 MHz input signal IS is generated by the PLL circuit 31 of the analog circuit 3, the frequency is changed by the mixer circuit 32, the power is amplified by the power amplifier 33, and the input signal IS is input to the SAW sensor 4. Next, the output signal OS is output by passing through the SAW sensor 4 with its amplitude and phase changed, the output signal OS is power-amplified by the LNA 34, and the baseband signal BS whose frequency is transferred by the mixer circuit 35 is the LPF36. The harmonic component is removed by the method, and a 10 MHz baseband signal BS is input to the A / D converter 37. Subsequently, the A / D converter 37 converts the digital signal DS into a digital signal DS at a sample rate of 11.975 μs at a time interval.

On the other hand, the digital signal is counted by the 2-bit counter 51 of the digital circuit 5 at sample rate time intervals of 11.975 μs, and the digital signal output from the A / D converter 37 according to the control signal (count value) is the orthogonal detection circuit. It is divided into an in-phase component signal and an orthogonal component signal by 7, and each is integrated for a predetermined time. Then, the integrated I value and Q value are output. That is, when the control signal is “0” (initial phase θ ₀ + 0 °), the I value of the first flip-flop 73 is 1, and the Q value of the second flip-flop 74 is 0. When the control signal is "1" (initial phase θ ₀ + 90 °), the I value of the first flip-flop 73 is 0, and the Q value of the second flip-flop 74 is 1, which is controlled. When the signal is "2" (initial phase θ ₀ + 180 °), the I value of the first flip-flop 73 is -1, the Q value of the second flip-flop 74 is 0, and the control signal is In the case of "3" (initial phase θ ₀ + 270 °), the I value of the first flip-flop 73 is 0, and the Q value of the second flip-flop 74 is -1.

After that, the integrated I value and Q value are input to the upper control unit 6, and the amplitude and phase of the inspection signal output from the SAW sensor 4 based on the I value and the Q value in the upper control unit 6 are changed. It is calculated and, as a result, the characteristics of the object to be measured are measured.

As described above, according to the characteristic measuring device 1 having such a configuration, the 250 MHz output signal OS output from the SAW sensor 4 is frequency-shifted by the mixer circuit 35, so that the sample rate is as long as 11.975 μs. Since it is undersampled at time intervals and converted into a digital signal DS, the sample rate of the A / D converter 37 may be low. Therefore, it is possible to reduce the power consumption of the characteristic measuring device 1. Further, since the characteristic measuring device 1 is provided with the quadrature detection circuit 7, it is not necessary to include a quadrature modulator, an NCO, a multiplication circuit, and the like, so that a SAW sensor having a simplified configuration can be realized.

Further, since the A / D converter 37 samples at a time interval of 11.975 μs, which is obtained by subtracting the period 0.025 μs for each quadrant of the baseband signal BS from the time 12 μs in which the accuracy of the SAW sensor 4 is ensured. One sampling is performed for one burst of the input signal IS. As a result, undersampling can be performed equivalently.

Although the embodiment of the present invention has been described above, the specific configuration is not limited to the above-described embodiment, and even if there is a design change or the like within a range that does not deviate from the gist of the present invention. Included in the invention. For example, in the above embodiment, the characteristic sensor is composed of the SAW sensor 4, but if it is a sensor that converts an input signal and outputs an output signal according to the characteristics of the object to be measured, it is a sensor other than the SAW sensor. It may be.

1 Characteristic measuring device 2 Oscillator 3 Analog circuit 31 PLL circuit (input signal generation means)
32 Mixer circuit (mixer)
33 Power Amplifier 34 LNA (Low Noise Amplifier)
35 Mixer circuit (mixer means)
36 LPF (Low Pass Filter)
37 A / D converter (conversion means)
4 SAW sensor (characteristic sensor)
5 Digital circuit 51 2-bit counter 6 Upper control unit (CPU)
7 Orthogonal detection circuit (Orthogonal detection means)
71 First logic circuit 72 Second logic circuit 73 First flip-flop 74 Second flip-flop

Claims (2)

An input signal generation means that receives an oscillation signal from an oscillator and generates an input signal based on the reference frequency of the oscillation signal.
A characteristic sensor that receives the input signal and outputs an output signal according to the characteristics of the object to be measured.
A mixer means that multiplies the input signal and the output signal and outputs a baseband signal having a predetermined frequency.
A conversion means that undersamples the baseband signal, converts it into a digital signal at a predetermined sample rate, and outputs the signal.
The digital signal is divided into an in-phase component signal and an orthogonal component signal according to the predetermined sample rate, and the orthogonal detection means for integrating the in-phase component signal and the orthogonal component signal for a predetermined time is provided.
Time interval of the predetermined sample rate, from the time that accuracy Ru is ensured influence of multiple reflections generated in the characteristics the sensor is reduced, by subtracting 1/4 the time period of the baseband signal values Is,
A characteristic measuring device equipped with a surface acoustic wave sensor. The characteristic sensor is a surface acoustic wave sensor that detects a change in the propagation characteristics of the surface acoustic wave of the input signal.
The characteristic measuring device according to claim 1.

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