JP6917264B2 - Motor control method and motor control device. - Google Patents

Description

The present invention relates to a motor control method and a motor control device.

In a motor, when the rotation speed is controlled, the inductance of the motor may fluctuate due to magnetic non-linearity and magnetic saturation. Therefore, as in the technique disclosed in Patent Document 1, the magnetic flux interlinkage number of each winding with respect to the current value is obtained in advance by analysis such as the non-linear finite element method of the motor, and these current values and the magnetic flux interlinkage number are used. There is known a technique for controlling the torque of a motor or the like by using a data table corresponding to the above. With this technique, control parameters can be optimized and motor control with higher accuracy can be realized.

Japanese Unexamined Patent Publication No. 2008-141835

Here, among the motors, there is a motor called a variable magnetic flux type (or variable leakage flux type) having a structure in which a magnetic barrier such as a physical gap is provided in the rotor. In this variable magnetic flux type motor, a part of the magnetic flux from the permanent magnet to the stator of the rotor is configured to be blocked by a magnetic barrier, and the amount of magnetic flux blocked by this magnetic barrier is the motor. It changes according to the applied current to. Therefore, since the amount of magnetic flux from the rotor to the stator changes according to the magnitude of the applied current, there is a feature that the amount of change in inductance is relatively large.

In this variable magnetic flux type motor, the inductance is greatly changed by the applied current due to the provision of the magnetic barrier. Therefore, the calculation accuracy of the inductance is not sufficiently high even if the analysis method disclosed in Patent Document 1 is used. Therefore, even if the torque, speed, position, etc. are controlled by using the calculated inductance, the calculation accuracy is high. Rotation control may not be possible. Therefore, there is a demand for a technique for further improving the accuracy of rotation control.

The present invention has been made in view of such problems, and an object of the present invention is to provide a motor control method capable of improving the accuracy of motor rotation control, and a motor control device.

In one aspect of the motor control method of the present invention, a magnetic barrier provided in a rotor provided with a permanent magnet can change the magnetic flux from the permanent magnet to the stator according to the current applied to the motor. This is a control method for a variable magnetic flux type motor. In this control method, the gain used for the feedback control of the applied current was calculated at a plurality of calculation timings in the path from the current operating point indicated by the applied current to the target operating point indicated by the target current, and the gain was used. The applied current is controlled by feedback control.

According to one aspect of the present invention, the accuracy of motor rotation control can be improved.

FIG. 1 is a cross-sectional view of the variable magnetic flux type motor of the first embodiment. FIG. 2 is a configuration diagram showing a part of the motor. FIG. 3 is a block diagram of the motor and its control device. FIG. 4 is a diagram showing the gain calculation timing. FIG. 5A is a diagram showing the gain calculation timing in the first modification. FIG. 5B is a diagram showing an example of a change in inductance. FIG. 5C is a diagram showing an example of a change in inductance. FIG. 6A is a diagram showing an example of the gain calculation timing in the second modification. FIG. 6B is a diagram showing another example of the gain calculation timing. FIG. 7 is a diagram showing the relationship between the torque of the motor and the time in the third modification. FIG. 8 is a block diagram of the motor and the control device of the second embodiment. FIG. 9 is a block diagram of the motor and the control device of the third embodiment. FIG. 10 is a diagram showing an operating point when the target current vibrates. FIG. 11 is a diagram showing an operating point when the target current vibrates. FIG. 12 is a diagram showing an operating point when the target current vibrates in the fourth modification. FIG. 13 is a diagram showing an operating point when the target current vibrates.

Hereinafter, embodiments of the present invention will be described with reference to the drawings.

(First Embodiment)
Generally, a motor called a variable magnetic flux type or a variable leakage flux type is configured so that the amount of magnetic flux from the rotor to the stator can be changed due to the physical configuration of the rotor or the like. It is known that the amount of change in this magnetic flux changes according to the current applied to the motor, and the inductance can be changed by controlling the applied current, so that the optimum operating point according to the required load is realized. be able to.

FIG. 1 is a cross-sectional view of the variable magnetic flux type electric motor (rotary electric machine) 100 of the first embodiment perpendicular to the axial direction, showing a quarter of the entire configuration. For the remaining three-quarters of the overall configuration, the partial configuration shown in FIG. 1 is continuously repeated. The motor 100 of the present embodiment includes a stator 1 having an annular shape and a rotor 2 which is concentric with the stator 1 and is arranged so as to have an air gap 13 between the stator 1 and the stator 1. , A plurality of permanent magnets 3 fitted to the rotor 2 are provided to form an electric motor or a generator.

The stator 1 includes a ring-shaped stator core 11, a plurality of teeth 8 protruding from the stator core 11 toward the inner peripheral side, and a slot 9 which is a space between adjacent teeth 8. A stator winding 10 is wound around the teeth 8. The stator core 11 is formed of, for example, an electromagnetic steel plate which is a soft magnetic material.

The rotor 2 has a rotor core 12. The rotor core 12 is formed in a cylindrical shape by a so-called laminated steel plate structure, which is formed by laminating a large number of electromagnetic steel plates formed by punching a metal steel plate having high magnetic permeability in an annular shape in the axial direction. ing. Further, in the vicinity of the peripheral portion of the rotor core 12 facing the stator 1, a plurality of permanent magnets 3 are equidistant to each other along the circumferential direction, and the polarities of the permanent magnets 3 adjacent to each other are different. It is provided to be polar. The rotor core 12 according to the variable magnetic flux type motor 100 of the present embodiment has an octapole structure in which eight permanent magnets 3 are provided along the circumferential direction, as inferred from the partial configuration shown in FIG. Has.

Further, the rotor 2 is a magnetic barrier (also referred to as a flux barrier) which is a space portion formed by punching an electromagnetic steel plate forming the rotor core 12 between the magnetic poles formed by the adjacent permanent magnets 3. ) It has 4 and 5. The magnetic barriers 4 and 5 have a higher reluctance than the magnetic steel sheet. Therefore, the magnetic barriers 4 and 5 act as magnetic flux barriers against the magnetic flux in the magnetic circuit in which the permanent magnet 3 is formed on the rotor 2. As shown in the figure, the magnetic barrier 4 is formed in a substantially triangular shape having a narrower width on the rotation center side than on the outer peripheral side of the rotor 2. The magnetic barrier 5 is formed in a substantially U shape on the rotation center side of the magnetic barrier 4 so as to cover the magnetic barrier 4. However, the shapes of the magnetic barriers 4 and 5 are not limited to the shapes shown in the figure as long as they have the technical effects described later. Further, since the magnetic barriers 4 and 5 are formed on the rotor 2 as shown in the figure, the magnetic flux generated from the permanent magnet 3 constituting one magnetic pole is formed by another permanent magnet 3 adjacent to the rotor 2. A magnetic flux bypass path 6 is formed as a path for leakage to the magnetic pole side. Further, the rotor 2 is provided with a narrow bridge-shaped portion 7 connected to the magnetic flux bypass path 6 on the outer peripheral side of the magnetic barrier 4.

The permanent magnet 3 is fixed to the rotor core 12 by being fitted into a gap formed in the corresponding portion of the rotor core 12. Further, in the permanent magnet 3, the radial direction of the rotor 2 is the magnetization direction. Here, in the present embodiment, the geometric center of the permanent magnet 3 is defined as the d-axis, and the position electrically orthogonal to the d-axis is defined as the q-axis. Since the variable magnetic flux type motor 100 of this embodiment has an 8-pole structure, the position of 22.5 degrees from the d-axis at the mechanical angle is defined as the q-axis. The magnetic barriers 4 and 5 described above are formed on the q-axis.

The above is the basic configuration of the variable magnetic flux type motor 100 of the first embodiment. Here, before the detailed description of the present embodiment, the basic principle of the variable magnetic flux type motor 100 which is the object of the present invention will be described.

The variable magnetic flux type motor 100, which is the object of the present invention, is a part of the magnetic flux (leakage magnetic flux) emitted from the permanent magnet 3 included in the rotor 2 due to the action of the load current (stator current) applied to the stator winding 10. ) Is characterized in that the magnetic path can be changed. Due to this feature, the motor 100 can passively change the stator interlinkage magnetic flux by controlling the stator current, so that the magnetic force of the permanent magnet 3 included in the rotor 2 is apparently variable by the current control. Has the property of being able to. Due to such characteristics, as described above, the variable flux type motor 100 which is the object of the present invention is also called a variable leakage flux type.

FIG. 2 is a configuration diagram showing a part of the motor 100, and is a diagram for explaining an outline of operation in the variable magnetic flux type. The shape and arrangement of each configuration of the variable magnetic flux type motor 100 shown in FIG. 2 are general ones shown for explaining the outline of operation.

The motor 100 includes a stator 1 and a rotor 2 arranged so as to form an air gap 13 between the stator 1. The rotor 2 is connected to the permanent magnets 3, the magnetic barriers 4 and 5, and the magnetic flux bypass path 6 connected via the magnetic barriers 4 and 5 provided between the two adjacent permanent magnets. Has. The direction indicated by the arrow on the left side of the figure indicates the rotation direction of the rotor 2.

The magnetic path direction 15 shown in FIG. 2 indicates the direction in which the leakage flux flows when no current is applied to the stator winding 10 included in the stator 1. As shown in FIG. 2, in the rotor, a part of the magnetic flux emitted from the permanent magnet 3 leaks to the adjacent different pole side through the magnetic flux bypass path 6.

Therefore, the amount of magnetic flux of the main magnetic flux component to the stator 1 side emitted from the permanent magnet 3, that is, the stator interlinkage magnetic flux is relatively reduced, so that the magnetic force of the permanent magnet 3 is apparently weakened. Thereby, the iron loss generated by the stator interlinkage magnetic flux can be reduced. This effect is particularly effective in improving the efficiency of the motor 100 in a low load and high speed region.

On the other hand, the magnetic path direction 20 indicates the direction in which the magnetic flux flows when a current is applied to the stator winding 10. As the stator current flows, the magnetic flux generated from the permanent magnet 3 is attracted to the stator side in the rotation direction of the rotor 2 as indicated by the magnetic path direction 20. Therefore, when the current is not applied to the stator winding 10, the magnetic flux leaking in the rotor 2 as the leakage flux can be efficiently converted into the stator interlinkage magnetic flux. As a result, the motor 100 can output a high torque equivalent to that in the state where the magnetic flux emitted from the permanent magnet 3 is not leaked.

In this way, the magnetic flux bypass path 6 is provided between the magnetic poles formed in the rotor 2, and the stator interlinkage magnetic flux is passively changed by changing the leakage flux in the rotor by controlling the stator current. The motor having the above is called a variable magnetic flux type (variable leakage flux type).

The above is the basic principle of the variable magnetic flux type motor 100 which is the object of the present invention. Hereinafter, the details of the control device that controls such a variable magnetic flux type motor 100 will be described.

FIG. 3 is a block diagram of the motor 100 and the control device 200. The control device 200 is composed of a microcomputer provided with a central arithmetic unit (CPU), a read-only memory (ROM), a random access memory (RAM), and an input / output interface (I / O interface). The control device 200 stores a program, and by executing the stored program, the control described later is executed.

The control device 200 uses the three-phase AC currents I _u , I _v, and I _{w for rotation control based on the d-axis target current I d} ^* and the q-axis target current I _q ^* calculated by the target current calculation unit 230. Is output to the motor 100. _{In the target current calculation unit 230, the d-axis target current I d} ^* and the q-axis target current I _q ^* that realize the desired rotation speed in the motor 100 are set based on the operation of the accelerator, brake, and the like by the driver. Calculated. Further, the rotation speed ω of the motor 100 is fed back to the control device 200.

A subtraction value obtained by subtracting the _{d-axis current I d} output from the inverter 211 from the _{d-axis target current I d} ^* is input to the d-axis PI control unit 201 from the subtractor 202 provided in the previous stage. Further, in the d-axis PI control unit 201, the d-axis gain G _d obtained by the gain map 212 is input. The d-axis gain G _d includes a proportional control gain and an integral control gain, and the d-axis PI control unit 201 performs _{proportional and integral control using the input d-axis gain G d} to control the d-axis. The current _Id is feedback controlled.

A subtractor 203 is provided after the d-axis PI control unit 201, and the d-axis non-interference control unit 204 _{is derived from the feedback-controlled d-axis target current I d} ^{* output from the d-axis PI control unit 201.} The subtraction value obtained by subtracting the d-axis non-interference control current output from is output to the inverter 211.

Regarding the q-axis, the q-axis target current I _q ^* input to the control device 200 is feedback-controlled by the q-axis PI control unit 221 and the subtractor 222, and then the subtractor 223 and the q-axis non-interference. The interference component is canceled by the control unit 224 to be non-interfering and output to the inverter 211. In the process of this calculation, the q-axis gain G _q is input from the gain map 212.

In the inverter 211, the three-phase AC currents Iu, Iv, and Iw are output to the motor 100 by controlling the switching element according to the input d-axis target currents I _d ^* and q-axis target currents I _q ^*. NS. In this way, the rotation control of the motor 100 is performed. An ammeter 101 is provided between the inverter 211 and the motor 100, and the three-phase AC currents Iu, Iv, and Iw measured by the ammeter 101 are converted from the uvw phase to the dq axis. Is performed, and the d-axis current I _d and the q-axis current I _q are fed back to the control device 200.

Here, the processing in the gain map 212 will be described. In the variable magnetic flux type motor 100, the inductance changes greatly according to the applied current. Further, in general, in the motor 100, the responsiveness in the feedback control of the applied current changes according to the inductance. That is, in the variable magnetic flux type motor 100, the responsiveness of the feedback control greatly differs depending on the applied current.

If the gains (proportional control gain and integral control gain) used for feedback control in the d-axis PI control unit 201 and the q-axis PI control unit 221 are large, the time until the current becomes a steady state is short, but vibration occurs. And there is a risk of divergence. On the other hand, if the gain is small, the possibility of divergence is small, but the time until the steady state is reached is long. Therefore, it is assumed that the gain map 212 stores in advance the gain that causes the steady state at the shortest without divergence according to the current applied to the motor 100.

Gain map 212 is. Using such a map stored in advance, the d-axis gain G _d and the q-axis gain G _q are set _{according to the d-axis current I d} and the q-axis current I _{q input to the motor 100.} calculate. The gain calculated by the gain map 212 is performed according to the calculation timing TM generated by the calculation timing generation unit 214. Then, the calculated gain will be used until the next calculation timing.

In the inductance estimation unit 213, the d-axis inductance L _d and the q-axis inductance L _{q of} the motor 100 are obtained _{from the inputs of the d-axis gain G d} and the q-axis gain G _q. This is because the current applied to the motor 100 corresponds to each of the gain and the inductance, so that the inductance can be estimated according to the obtained gain. The inductance can also be estimated from the current applied to the motor 100.

In the calculation timing generation unit 214, the _{current operating point determined by the input d-axis current I d} and the q-axis current I _q , the d-axis target current I _d ^* , and the q-axis target current I _q ^{* are used.} The gain calculation timing is calculated from the determined target operating point, and the calculation timing TM indicating this timing is output to the gain map 212. Operating states such as the control cycle, load conditions, and rotation speed of the motor 100 are input to the calculation timing generation unit 214, and the calculation timing may be calculated using these operating states.

The d-axis non-interference control unit 204 and the q-axis non-interference control unit 224 are a d-axis non-interference control current and a q-axis non-interference control current that realize mutual non-interference in the d-axis and the q-axis, respectively. Is obtained, and each is output to the subtractor 203 and the subtractor 223. The d-axis non-interference control unit 204 controls d-axis non-interference from the rotation speed ω of the motor, the d-axis current I _d output from the inverter 211, and the d-axis inductance L _{d estimated by the inductance estimation unit 213.} Find the current. Similarly, the q-axis non-interference control unit 224 is based on the rotational speed ω of the motor, the q-axis current I _q output from the inverter 211, and the q-axis inductance L _{q estimated by the inductance estimation unit 213.} Find the non-interference control current.

Here, in general motor control, a fixed value is often used for the gain. On the other hand, in the variable magnetic flux type motor 100, since the inductance changes greatly according to the applied current, if a fixed value is used as the gain used for feedback control, the followability deteriorates, and overshoot and stabilization occur. It may take some time. Therefore, by sequentially calculating the gain in the gain map 212 at the calculation timing instructed by the calculation timing generation unit 214, the time until stabilization can be shortened without divergence by the feedback control.

FIG. 4 is a diagram showing the calculation timing by the calculation timing generation unit 214.

In this figure, the d-axis current I _d is shown on the horizontal axis, the q-axis current I _q is shown on the vertical axis, and the operating point of the motor 100 is shown. from the d-axis current I _d and the q-axis current I _q is the initial operating point both become zero P0, which is a target operating point indicated by the d axis target current I _d ^* and q-axis target current I _q ^* to P ^* And the route for changing the applied current are shown. Then, in this example, in the path from the initial operating point P0 to the target operating point P ^* , the gain is calculated by the gain map 212 at a plurality of calculation timings from P1 to P7. The calculation timings from P1 to P7 need not be determined at one time. For example, the sequential calculation timing may be obtained so that P2 is determined in P1 and then P3 is determined in P2.

Specifically, for example, at the initial operating point P0, a gain is calculated from the current value, and feedback control is performed from the initial operating point P0 to the operating point P1 which is the next calculation timing using the calculated gain. It is said. At the operating point P1, the gain is calculated from the current value at the operating point P1, and feedback control is performed from the operating point P1 to the operating point P2, which is the timing for calculating the next gain. By calculating the sequential gain in this way, the control accuracy of the feedback control can be improved.

In the present embodiment, the gain is calculated at seven operating points on the path, but the gain is not limited to this. The sequential gain may be calculated at a plurality of operating points. Further, the calculation timing generation unit 214 can appropriately determine the gain calculation timing, the gain calculation timing, and the calculation cycle according to the operation state such as the control cycle, the load condition, and the rotation speed of the motor 100.

According to the first embodiment, the following effects can be obtained.

The motor 100 controlled by the control method of the first embodiment is a variable magnetic flux type, has a rotor 2 provided with magnetic barriers 4 and 5, and has a rotor 2 according to an applied current. The magnetic flux from the permanent magnet 3 to the stator 1 can be changed. In order to control the motor 100, the gains used for feedback control in the d-axis PI control unit 201 and the q-axis PI control unit 221 using the d-axis _{current I d} and the q-axis current I _{q applied to the motor 100 are gain maps.} Obtained at 212.

The calculation timing generation unit 214 sets a plurality of operating points P1 to P7 on the path as calculation timings in the path from the initial operating point P0 to the target operating point P ^{* indicated by the target current.} Then, the gain map 212 sequentially calculates the gain based on the current of the operating point at the calculation timing. Feedback control is performed by the d-axis PI control unit 201 and the q-axis PI control unit 221 using the gains calculated sequentially in this way, and the currents I _d , I _q (I _u , I) applied to the motor 100 are applied. _v , I _w ) is required.

The variable magnetic flux type motor 100 has a feature that the inductance changes greatly according to the applied current. Further, in general, in the motor 100, the responsiveness in the feedback control of the applied current differs depending on the inductance. That is, in the variable magnetic flux type motor 100, the responsiveness of the feedback control differs greatly depending on the input current, so the d-axis gain G _d and the q-axis gain G _q used for the feedback control are calculated according to the applied current. There is a need to.

Therefore, when controlling the change from the initial operating point P0 to the target operating point P ^* in the motor 100, a plurality of calculation timings are set in the change path, and the gain corresponding to the applied current is set at these calculation timings. Make a calculation. By doing so, the timely gain is calculated even if the inductance changes significantly, so that feedback control is appropriately performed, divergence can be suppressed, and the stabilization time can be shortened.

According to the control method of the motor 100 of the first embodiment, the calculation timing generation unit 214 determines the gain calculation timing according to the operating state such as the control cycle, the load condition, and the rotation speed of the motor 100. By doing so, for example, when the amount of change in the inductance of the motor 100 is large, the optimum feedback control can be realized by shortening the gain calculation interval. On the other hand, when the amount of change in the inductance of the motor 100 is small, an unnecessary change in gain can be suppressed by lengthening the calculation interval.

According to the control method of the motor 100 of the first embodiment, the gain map 212 calculates the gain at the gain calculation timing based on the current value indicating the current calculation timing operating point. By doing so, the gain can be sequentially calculated with relatively simple control, so that the entire process can be simplified.

(First modification)
In the above-mentioned example of the first embodiment, an example in which the gain is corrected at a predetermined calculation cycle based on the current at the current operating point at the gain calculation timing has been described. In this embodiment, a modified example of the method for setting the gain calculation timing will be described.

FIG. 5A is a diagram showing a first modification of the gain calculation timing. In this figure, only the path of change of the operating point is shown in one dimension.

According to this example, the calculation timing generation unit 214 sets the operating points P1 to P6 as the gain calculation timing in the path ^{from the initial operating point P0 to the target operating point P *.} Then, at each gain calculation timing, an intermediate operating point between the operating point of the current calculation timing and the operating point at the next calculation timing is obtained, and the gain is calculated using the current indicating this intermediate operating point.

Specifically, for example, at the initial operating point P0, the calculation timing generation unit 214 sets an intermediate operating point between the initial operating point P0 of the current calculation timing and the operating point P1 at the next calculation timing on the path. It is generated between the initial operating point P0 and the operating point P1. Then, the intermediate operating point is transmitted to the gain map 212 by using the calculation timing TM. In this way, since the gain at the intermediate operating point is calculated in the gain map 212, the feedback control from the initial operating point P0 to the operating point P1 can be performed using the gain.

Here, a case where the inductance at the next calculation timing is smaller than the inductance at the current operating point at the timing of calculating the gain will be examined.

FIG. 5B is a diagram showing the relationship between the current and the inductance when the inductance becomes small. This figure shows the operating point at the current calculation timing and the operating point at the next calculation timing. Then, it is shown that the inductance L decreases from the current operating point to the operating point at the next calculation timing.

Considering the section from the current operating point to the operating point of the next calculation timing, the current operating point, which is the start timing of this section, is based on the intermediate operating point even though the inductance is relatively large. Since the relatively small gain calculated in the above is used for the feedback control, the responsiveness becomes low. On the other hand, at the operating point of the next calculation timing, which is the end timing of this section, a relatively large gain calculated based on the intermediate operating point is used even though the inductance is relatively small. , High responsiveness. As described above, in the section where the initial operating point P0 changes to the operating point P1, the stability is expected to be improved by gradually increasing the responsiveness.

Next, at the timing of calculating the gain, a case where the inductance at the next calculation timing is larger than the inductance of the current operating point will be examined.

FIG. 5C is a diagram showing the relationship between the current and the inductance when the inductance becomes large. In this figure, it is shown that the inductance L increases from the current operating point to the operating point at the next calculation timing.

In the section from the current operating point to the operating point of the next calculation timing, the comparison calculated based on the intermediate operating point is performed at the current operating point, which is the start timing, even though the inductance is relatively small. Since a large gain is used, the responsiveness is high. On the other hand, at the operating point of the next calculation timing, which is the end timing of this section, a relatively small gain calculated based on the intermediate operating point is used even though the inductance is relatively large. , The responsiveness becomes low. As described above, in the section where the operating point changes from P0 to P1, the responsiveness gradually decreases, so that the responsiveness is high at the beginning of the section and the divergence is likely to occur, but the divergence is expected to be suppressed at the end.

According to the first modification, the following effects can be obtained.

According to the control method of the motor 100 of the first modification, the calculation timing generation unit 214 sets an intermediate operating point between the current operating point and the operating point at the next calculation timing as a path between the two in the gain calculation timing. Generate in. Then, the gain map 212 calculates the gain based on the current indicating the intermediate operating point. By doing so, even if the inductance of the motor 100 changes significantly until the next gain calculation timing, feedback is provided by using the gain calculated using the current at the intermediate operating point. It is possible to suppress divergence and prolonged convergence time due to control.

According to the control method of the motor 100 of the first modification, as shown in FIG. 5B, the inductance of the motor 100 decreases from the current operating point to the operating point at the next calculation timing. In such a case, the stability can be improved by gradually increasing the responsiveness in the section from the current operating point to the operating point at the next calculation timing.

According to the control method of the motor 100 of the first modification, as shown in FIG. 5C, the inductance of the motor 100 increases from the current operating point to the operating point at the next calculation timing. In such a case, divergence can be suppressed by gradually lowering the responsiveness in the section from the current operating point to the operating point at the next calculation timing.

(Second modification)
In the examples so far in the first embodiment, an example in which the gain calculation cycle is constant has been described. In the second modification, an example of changing the calculation cycle will be described.

6A and 6B are diagrams showing changes in the operating point in the second modification.

In the case shown in FIG. 6A, the difference in the q-axis current I _q between the initial operating point P0 and the target operating point P ^* is larger than that in the case shown in FIG. 6B. Here, since the q-axis current I _q affects the torque of the motor 100, the case shown in FIG. 6A is from the initial operating point P0 to the target operating point P ^{* as compared with the case shown in FIG. 6B.} The change in torque of the motor 100 between them is small.

Therefore, when the change in torque is relatively large as shown in FIG. 6A, a large change in gain is expected. Therefore, the gain calculation interval is shortened between ^{the initial operating point P0 and the target operating point P *.} Set. On the other hand, when the change in torque is relatively small as shown in FIG. 6B, the calculation interval is set long.

As a result, when the torque fluctuation is large and it is necessary to increase the gain calculation frequency, the gain calculation frequency is increased by shortening the gain calculation interval, and the followability in the feedback control can be improved. .. On the other hand, when the fluctuation of the torque is small and it is not necessary to calculate a large amount of gain, the gain calculation frequency becomes low by lengthening the gain calculation interval, and unnecessary processing can be omitted.

According to the second modification, the following effects can be obtained.

According to the control method of the motor 100 of the second modification, when the change in the q-axis current is large in the path ^{from the initial operating point P0 to the target operating point P *, the calculation timing generation unit 214 calculates the gain calculation timing.} Increase the interval between. By doing so, the gain can be calculated frequently when the torque fluctuation is large, so that the followability can be improved. On the other hand, when the change in the d-axis current is small and the change in the torque is small, it is not necessary to change the gain frequently. Therefore, by lengthening the gain calculation interval, unnecessary processing can be omitted.

(Third modification example)
In the second modification, an example of changing the gain calculation cycle has been described. In the third modification, an example in which a plurality of gain calculation cycles are variably set will be described.

FIG. 7 is a diagram showing the relationship between the torque of the motor 100 and the elapsed time when controlled by the control device 200. The vertical axis shows the torque, and the torque at the initial operating point P0 and the target operating point P ^* is shown as T0 and T ^*. The times at which the torques T0 and T ^* are obtained are indicated as t0 and te.

According to this figure, the torque has a large rate of change when its value is small, and a small rate of change when its value is large. Therefore, with respect to the torque, T1 to T8 are set so that the intervals ^{from the torques T0 to T * are evenly spaced.} Then, the time corresponding to the torques T1 to T8 is set as t1 to t8.

For example, in the section where the torque changes from T0 to T1, the rate of change of the torque is relatively large, so that the gain calculation interval from the time t0 to t1 is relatively short. On the other hand, in the section from the torque change from T8 to the target torque Tf, the torque change rate is relatively small, so that the gain calculation interval from the time t8 to the target time tf is relatively long.

In this way, the calculation timing generation unit 214 sets the gain calculation timings t1 to t8 so that the amount of change in torque is evenly spaced by using the relationship between torque and time as shown in FIG. do. By doing so, when the rate of change in torque is large, the number of times the gain is calculated increases, so that the followability can be improved, and when the rate of change in torque is small, the number of times the gain is calculated decreases. Unnecessary processing can be omitted.

According to the third modification, the following effects can be obtained.

According to the control method of the motor 100 of the third modification, the calculation timing generation unit 214 determines the gain when the torque change rate is relatively large in the path ^{from the initial operating point P0 to the target operating point P *.} Shorten the calculation interval. By doing so, the gain is updated frequently, so that the followability can be improved. On the other hand, when the rate of change in torque is relatively small, it is less necessary to change the gain frequently. Therefore, by lengthening the calculation interval, unnecessary processing can be omitted.

(Second Embodiment)
In the first embodiment, an example in which the inductance estimation unit 213 is provided after the gain map 212 has been described. In this embodiment, an example in which a gain calculator is provided after the inductance estimation unit 213 will be described.

FIG. 8 is a diagram showing the control device 200 of the present embodiment.

According to this figure, the control device 200 of the present embodiment has a d-axis voltmeter 231 added between the subtractor 203 and the inverter 211 as compared with the control device 200 of the first embodiment, and the subtractor. A q-axis voltmeter 232 is added between the 223 and the inverter 211.

Further, to the inductance estimation unit 213, in addition to the d-axis current I _d and the q-axis current I _{q , the d-axis voltage V d} measured by the d-axis voltmeter 231 and the q-axis voltmeter 232 are sent. The q-axis voltage V _q measured is input. The inductance estimation unit 213 obtains the d-axis inductance L _d from the d-axis current I _d and the d-axis voltage V _d, and obtains the q-axis inductance L q from the q-axis current I _q and the q-axis voltage V _q . Then, the d-axis inductance L _d obtained is output to the d-axis gain calculator 233 and the d-axis non-interacting controller 204, the q-axis inductance Lq is, the q-axis gain calculator 234 and the q-axis non-interacting controller 224 Is output to.

The d-axis gain calculator 233 obtains the _{d-axis gain G d} used by the d-axis PI control unit 201 based on _{the d-axis inductance L d.} The q-axis gain calculator 234 obtains the _{q-axis gain G q} used by the q-axis PI control unit 221 based on the q-axis inductance Lq. This is because, in the variable magnetic flux type motor 100, the responsiveness changes according to the inductance, so that the gain used for the feedback control can be determined based on the inductance.

In the d-axis gain calculator 233 and the q-axis gain calculator 234, the calculation timing generated by the calculation timing generation unit 214, that is, the initial stage, is the same as in the case shown in FIG. 4 in the first embodiment. The gain is calculated at a plurality of timings along the path from the operating point P0 to the target operating point P ^*. By calculating the sequential gain in this way, the control accuracy of the feedback control can be improved.

Also in this embodiment, as shown in the first modification to the third modification of the first embodiment, the feedback control can be performed more accurately by determining the gain calculation timing.

According to the second embodiment, the following effects can be obtained.

According to the control method of the second embodiment, the d-axis gain calculator 233 and the q-axis gain calculator 234 calculate the gain based on the inductance calculated by the inductance estimation unit 213. Here, since the responsiveness of the motor 100 changes according to the inductance, the gain can be obtained also based on the inductance. Therefore, in addition to the embodiment in which the gain is calculated based on the d-axis current I _d and the q-axis current I _{q as} in the first embodiment, the gain is set in a different embodiment using inductance to stabilize the feedback control. Therefore, the degree of freedom in design can be improved.

(Third Embodiment)
In the first embodiment and the second embodiment, an example of measuring the inductance by using the feedback input from the motor 100 has been described. In the third embodiment, an example of measuring the inductance by slightly changing the operating point in order to measure the inductance will be described.

FIG. 9 is a diagram showing a control device 200 according to the present embodiment. Compared with the control device 200 of the second embodiment, the control device 200 of the present embodiment is in that the calculation timing generation unit 214 of the control device 200 notifies the target current calculation unit 230 of the gain calculation timing TM. different. When the target current calculation unit 230 receives the calculation timing TM from the calculation timing generation unit 214, the target current calculation unit 230 _{vibrates the d-axis target current I d} ^* with _{an amplitude of ΔI d} for a predetermined time. By doing so, the current applied to the motor 100 also oscillates.

FIG. 10 shows an operating point when the d-axis target current I _d ^{* vibrates.}

According to this figure, it is assumed that the lower limit of the d-axis current is _id1 and the upper limit is _id2 for the operating point that vibrates at a certain gain calculation timing. Here, when the d-axis current is _id1 , the q-axis voltage is V _q1 and the q-axis current is i _q1 . d-axis current to the V _q2, q-axis current q-axis voltage i _q2 Then when it is i _d2, the following expression holds.

However, it is assumed that the resistance of the motor 100 is indicated by Ra and the differential operator is indicated by ρ. Further, the magnet magnetic flux in the motor 100 is ψ _a , and the interlinkage magnetic flux is λ.

By using these equations, without using a map as in the first embodiment and the second embodiment, it is possible to estimate the d-axis inductance L _d and the q-axis inductance Lq.

According to the third embodiment, the following effects can be obtained.

According to the control method of the motor 100 of the third embodiment, the current value indicating the operating point at the calculation timing is vibrated with a predetermined amplitude. In such a case, for example, for two operating points, that is, the operating point corresponding to the upper limit current of vibration and the operating point corresponding to the lower limit current, the respective operating points, the corresponding current values, and the voltage. Can be used to determine the inductance of the motor 100. Therefore, unlike the first and second embodiments, it is not necessary to have the gain map 212, so that the capacity of the storage memory can be reduced.

According to the control method of the motor 100 of the third embodiment, the d-axis target current I _d ^{* of the} target current is vibrated. By doing so, the q-axis target current I _q ^*, which affects the torque of the motor 100, does not change, so that the impedance can be estimated without affecting the rotation control of the motor 100.

In the present embodiment, the d-axis current I _d, which has little effect on the torque on the motor 100, is oscillated, but the present invention is not limited to this. The q-axis current I _q may be oscillated, or the d-axis current indicating the operating point of the calculation timing in order to change both the _{d-axis current I d} and the q-axis current I _{q as shown in FIG.} Further, the current may be changed into a circular locus centered on the q-axis current and having r as a radius.

_{In such a case, the d-axis inductance L d} and the q-axis inductance Lq can be estimated by using the following equations and the equations (2) and (3) instead of the equation (1). In this example, since the q-axis current also changes, the d-axis voltages V _d1 and V _d2 are further used.

(Fourth modification)
In the example of the third embodiment, an example in which the current indicating the operating point at the calculation timing is vibrated with a predetermined amplitude has been described. In the fourth modification, an example in which this amplitude is changed for each operating point will be described.

FIG. 12 shows a plurality of operating points that the d-axis target current I _d ^{* vibrates.}

According to this figure, the amplitude is set to increase in the order of _{Δ d1} , Δ _d2 , and Δ _d3 as the operating points P1, P2, and P3, which are the gain calculation timings, change. The amplitude shall be determined at a constant rate according to the magnitude of the d-axis current at the operating point. That is, when the absolute value of the d-axis target current I _d ^* is relatively small as in the operating point P1, the vibration is performed with a relatively small amplitude Δ _d1. On the other hand, when the absolute value of the d-axis target current I _d ^* is relatively large as in the operating point P3, the vibration is performed with a relatively large amplitude Δ _d3. By doing so, it is possible to vibrate at each operating point with an amplitude of a magnitude corresponding to the current, so that vibration at an unnecessary magnitude can be suppressed.

According to the fourth modification, the following effects can be obtained.

According to the control method of the motor 100 of the fourth modification, when the d-axis current increases as the operating points change to P1, P2, and P3, the amplitude of vibration at each operating point becomes Δ _d1 , Δ _d2. , Δ _d3, and so on. By doing so, the amplitude can be set according to each operating point, and the vibration with an unnecessarily large amplitude is suppressed. Therefore, the vibration for measuring the inductance is the rotation control of the motor 100. The effect on can be suppressed.

_{In this modification, the d-axis current I d,} which has little effect on the torque on the motor 100, is oscillated, but the present invention is not limited to this. The q-axis current I _q may be oscillated, or the d-axis current indicating the operating point of the calculation timing in order to change both the _{d-axis current I d} and the q-axis current I _{q as shown in FIG.} Further, the current may be changed into a circular locus centered on the q-axis current and having r as a radius. In such a case, the radius r changes according to the magnitude of the d-axis current.

Although the embodiments of the present invention have been described above, the above embodiments are only a part of the application examples of the present invention, and the technical scope of the present invention is limited to the specific configurations of the above embodiments. do not have.

100 Motor 200 Controller 201 d-axis PI controller 212 Gain map 213 Inductance estimator 214 Calculation timing generator 221 q-axis PI controller 230 Target current calculation unit

Claims (11)

A variable magnetic flux type motor control method capable of changing the magnetic flux from the permanent magnet to the stator according to the current applied to the motor by a magnetic barrier provided on the rotor provided with the permanent magnet. ,
The gain used for the feedback control of the applied current is calculated at a plurality of calculation timings in the path from the current operating point indicated by the applied current to the target operating point indicated by the target current.
A motor control method in which the applied current is controlled by the feedback control using the gain. In the motor control method according to claim 1,
A motor control method in which the inductance of the motor is obtained using the current of the operating point in the path, and the gain is calculated based on the inductance of the motor. In the motor control method according to claim 2,
A motor control method in which the applied current is vibrated by vibrating the target current, and the inductance of the motor is calculated using a plurality of values of the vibrated applied current. In the motor control method according to claim 3,
A method for controlling a motor, wherein the target current to be vibrated is a d-axis current. In the motor control method according to claim 3 or 4,
A motor control method in which the amplitude of the vibration is determined according to the operating point. In the motor control method according to any one of claims 1 to 5,
A plurality of motor control methods for which the calculation timing is determined according to the operating state of the motor. In the motor control method according to claim 6,
A motor control method in which the calculation timing interval is set so as to become shorter as the amount of torque change in the path increases. In the motor control method according to claim 6,
A motor control method in which the calculation timing interval is set so that the amount of torque change is equal at the interval. In the motor control method according to any one of claims 1 to 8,
A plurality of the calculation timings are determined based on the current current at the operating point.
Motor control method. In the motor control method according to any one of claims 1 to 9,
A motor control method in which an intermediate operating point between the current operating point and the operating point at the next calculation timing is generated at the calculation timing, and the gain is calculated based on the current at the intermediate operating point. .. A variable magnetic flux type motor control device capable of changing the magnetic flux from the permanent magnet of the rotor toward the stator according to the current applied to the motor by a magnetic barrier provided on the rotor. ,
A calculation timing generator that determines a plurality of gain calculation timings used for feedback control of the applied current in a path from the current operating point indicated by the applied current to the target operating point indicated by the target current.
A gain calculation unit that calculates the gain at the calculation timing,
A motor control device including an inverter that applies a current obtained by the feedback control using the gain to the motor.

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