JP6635289B2 - Wireless receiving device and received data restoring device - Google Patents

Description

  The present invention relates to a wireless receiving device and a received data restoring device, and more particularly, to a wireless receiving device and a received data restoring device used in a wireless communication system that performs multi-carrier transmission by a multiple-input and multiple-output (MIMO) method.

  In a wireless communication system that performs multicarrier transmission, the carrier frequencies generated by the local oscillation circuits of both the transmitter and the receiver are ideally designed to be the same, but are different in an actual environment. Similarly, in the transmitter and the receiver, the sampling frequency of the D / A converter and the A / D converter is different in an actual environment. That is, the received signal processed on the receiver side includes these frequency offsets.

  In wireless communication using the OFDM (Orthogonal Frequency Division Multiplexing) method, orthogonality between subcarriers is disrupted due to a frequency offset of a received signal, thereby causing inter-carrier interference. Therefore, conventionally, a technique for correcting a frequency offset in an OFDM receiving apparatus has been proposed.

  For example, WO 2007/091320 (Patent Literature 1) discloses that a phase rotation amount of a symbol is estimated by using a GI (guard interval) method, and the symbol rotation amount is determined according to a carrier frequency offset amount obtained from the phase rotation amount. Discloses that an oscillation frequency of an oscillation signal output to an RF unit is analog-corrected.

  Also, Japanese Patent Application Laid-Open No. 2000-224134 (Patent Document 2) discloses that digitally correcting a symbol phase error caused by a carrier frequency error and a guard interval period.

  Further, Japanese Patent Application Laid-Open No. 2006-108763 (Patent Document 3) discloses that a phase change of an OFDM symbol caused by a shift of a sampling frequency between a transmitter and a receiver is digitally corrected.

International Publication No. 2007/091320 JP 2000-224134 A JP 2006-108763 A

  In a wireless communication system employing a so-called SISO (Single Input Single Output) system in which a transmitting antenna and a receiving antenna are in a one-to-one relationship, interference between subcarriers can be reduced even if the frequency offset is regarded as a fixed value and is roughly corrected. Reduced. For example, in a device that generates a carrier frequency in the 5 GHz band, it is known that sufficient restoration accuracy can be obtained by correcting the carrier frequency to about 5 ppm.

  On the other hand, in a wireless communication system that employs the MIMO method, since complex channel estimation using a multiplexed stream and highly accurate MIMO decoding are required, sufficient restoration accuracy cannot be obtained by conventional frequency offset correction. In particular, when the number of streams is large, the frequency offset situation and its correction accuracy affect MIMO decoding.

  The present invention has been made in order to solve the above-described problems, and an object of the present invention is to provide a radio receiving apparatus capable of increasing the restoration rate of received data even when the number of streams is large, and And to provide a received data restoring device.

  A wireless receiving apparatus according to an aspect of the present invention is a receiving apparatus used in a wireless communication system that performs multicarrier transmission according to a MIMO scheme, and includes a plurality of receiving antennas for receiving transmission signals of a plurality of streams, and receiving signals at each receiving antenna. Demodulation means for demodulating the obtained RF band signal and outputting a baseband signal, and digital processing means for restoring received data by digitally processing the baseband transmission signal in stream units. . The digital processing means includes an extracting means, an estimating means, a correcting means, and a converting means. The extracting means removes the signal of the redundant part for each symbol constituting the transmission signal and extracts the signal of the data part. The estimating means estimates the amount of phase rotation of the data part in time units by linearly approximating the amount of change in the phase rotation of the entire symbol calculated using the signal of the redundant part for each symbol. The correction means individually corrects the signal of the data part for each signal based on the amount of phase rotation in time units estimated by the estimation means, and outputs the signal of the data part within one symbol and over a plurality of symbols in packet units. Is corrected. The conversion means converts the data signal corrected by the correction means into a subcarrier signal.

  Preferably, in each symbol, the correction means calculates the phase shift compensation amount of each signal of the data part by adding the phase rotation amount of the previous symbol to the phase rotation amount in time units. It is desirable that the correction unit corrects the signal of the data part by applying a reverse rotation to each signal of the data part by the calculated phase shift compensation amount.

  Preferably, the digital processing means further includes means for calculating a change amount of the phase rotation of the entire symbol as an average value of the rotation amount calculated using the signal of the redundant portion for each of the plurality of streams.

  Preferably, the digital processing means further includes a MIMO processing means, a linear approximation means, and a subcarrier correction means. The MIMO processing unit separates the subcarrier signal obtained from the conversion unit for each stream, and converts it into a separated subcarrier signal. The linear approximation unit linearly approximates the amount of change in phase rotation based on pilot subcarriers included in the separated subcarrier signal for each frequency domain symbol, and calculates a linear expression. The subcarrier correction means calculates a phase shift of each data subcarrier included in the separated subcarrier signal for each frequency domain symbol based on the linear expression calculated by the linear approximation means, The carriers are rotated in reverse.

  It is desirable that the linear approximation unit calculates a linear expression for each frequency domain symbol using an average value of a plurality of streams for each pilot subcarrier.

  A reception data restoration device according to another aspect of the present invention is a device for restoring reception data by digitally processing a transmission signal of a baseband in a stream unit, comprising: an extraction unit, an estimation unit, and a correction unit. And conversion means. The extracting means removes the signal of the redundant part for each symbol constituting the transmission signal and extracts the signal of the data part. The estimating means estimates the amount of phase rotation of the data part in time units by linearly approximating the amount of change in the phase rotation of the entire symbol calculated using the signal of the redundant part for each symbol. The correction means individually corrects the signal of the data part for each signal based on the amount of phase rotation in time units estimated by the estimation means, and outputs the signal of the data part within one symbol and over a plurality of symbols in packet units. Is corrected. The conversion means converts the data signal corrected by the correction means into a subcarrier signal.

  According to the present invention, the restoration rate of received data can be increased even when the number of streams is large.

FIG. 2 is a diagram showing an outline of a MIMO-OFDM communication system according to Embodiment 1 of the present invention. FIG. 3 is a diagram for explaining an influence on a received signal when a carrier frequency is different from a carrier frequency of a MIMO-OFDM transmitting apparatus in the MIMO-OFDM receiving apparatus according to Embodiment 1 of the present invention. FIG. 2 is a block diagram showing a circuit configuration of a MIMO-OFDM receiver according to Embodiment 1 of the present invention. FIG. 4 is a block diagram illustrating a functional configuration of a restoration processing circuit illustrated in FIG. 3. FIG. 3 is a diagram illustrating a data structure of an OFDM signal output from a synchronization unit according to the first embodiment of the present invention. FIG. 6 is a diagram for describing a method of calculating a change amount of the phase rotation of the entire symbol according to the first embodiment of the present invention. FIG. 4 is a diagram for explaining a phase shift of a signal of a data part calculated for each symbol in the first embodiment of the present invention. FIG. 4 is a diagram illustrating a method of correcting a signal of a data portion performed for each symbol in the first embodiment of the present invention. 7 is a graph showing a change in phase rotation calculated by analysis for a certain packet in the first embodiment of the present invention. 5 is a flowchart illustrating a received data restoration process according to the first embodiment of the present invention. FIG. 7 is a diagram illustrating a difference in output of the demapping unit between a case where there is no offset of a carrier frequency and a case where there is a carrier frequency offset. FIG. 9 is a block diagram showing a functional configuration of a restoration processing circuit in a MIMO-OFDM receiving apparatus according to Embodiment 2 of the present invention. FIG. 9 is a diagram for explaining an influence on a received signal when a sampling frequency is different from a sampling frequency of a MIMO-OFDM transmitting apparatus in a MIMO-OFDM receiving apparatus according to Embodiment 2 of the present invention. 9 is a flowchart illustrating a frequency domain correction process executed in the second embodiment of the present invention. FIG. 15 is a diagram schematically illustrating a process of step S34 in FIG. 14. FIG. 14 is a diagram illustrating an output example of a demapping unit when a sampling frequency offset exists.

  Embodiments of the present invention will be described in detail with reference to the drawings. In the drawings, the same or corresponding portions have the same reference characters allotted, and description thereof will not be repeated.

<First Embodiment>
First, an outline of the MIMO-OFDM receiving apparatus according to the present embodiment will be described.

  Referring to FIG. 1, a MIMO-OFDM receiver (hereinafter abbreviated as “OFDM receiver”) 10 receives a signal wirelessly transmitted from a MIMO-OFDM transmitter (hereinafter abbreviated as “OFDM transmitter”) 90. . Each of the OFDM transmitter 90 and the OFDM receiver 10 has a plurality of antennas, and performs data transmission of video data and the like by the MIMO method. The wireless communication system according to the present embodiment employs, for example, the 8 × 8 MIMO scheme, and performs stream transmission by allocating antennas in subcarrier units.

  In this case, the OFDM transmitting apparatus 90 includes eight antennas 91 (91a to 91h), and the OFDM receiving apparatus 10 includes eight antennas 11 (11a to 11h). The OFDM transmitting apparatus 90 divides transmission data into a plurality of streams, and simultaneously transmits different streams from the antennas 91a to 91h. The antennas 11a to 11h of the OFDM receiver 10 simultaneously receive the signal sequence of the stream transmitted from each antenna 91 of the OFDM transmitter 90. The OFDM receiving apparatus 10 demodulates a signal received by each antenna 11, and separates and extracts mixed stream data from the demodulated received signal.

  The OFDM receiving apparatus 10 and the OFDM transmitting apparatus 90 employ a quadrature amplitude modulation scheme. For example, any one of 256 QAM, 64 QAM, 16 QAM, and QPSK is selectively or fixedly used.

  The OFDM receiver 10 restores data by digitally processing the received signal after demodulation. Therefore, if the received signal input to the digital circuit has distortion, correct data cannot be extracted. This will be described with reference to FIG. In FIG. 2, data transmission in the case where the number of antennas (the number of streams) is one-to-one is illustrated for ease of understanding.

Referring to FIG. 2, an RF band signal (hereinafter, referred to as “RF signal”) r (t) is transmitted from antenna 91 of OFDM transmitting apparatus 90. RF signal r (t) is generated by modulating a carrier CW ₁ at baseband of the signal x (t). Specifically, RF signal r (t) obtained by multiplying the signal x (t) to a carrier CW _1.

The OFDM receiver 10 demodulates the RF signal r (t) received by the antenna 11 and extracts a baseband received signal y (t). Received signal y (t) is obtained by multiplying the carrier CW ₂ into an RF signal r (t).

Ideally, the carrier frequency fc ₁ on the transmitting side and the carrier frequency fc ₂ on the receiving side have the same value fc. Therefore, the carrier CW ₁ on the transmitting side and the carrier CW ₂ on the receiving side are:
It is expressed as

  In this case, the reception signal y (t) is the same as the transmission signal x (t) as represented by the following equation (1).

However, in a real environment, the carrier frequency fc ₁ on the transmitter side and the carrier frequency fc ₂ on the receiver side do not have the same value fc, and the carrier CW ₁ on the transmitter side and the carrier CW ₂ on the receiver side are:
It is expressed as

In this case, the received signal y (t) does not become the same as the transmitted signal x (t), and as shown by the following equation (2), the received signal y (t) includes the carrier frequency fc ₁ on the transmitter side. the difference between the carrier frequency fc ₂ of the receiver side, i.e. will ride error offset Delta] f. That is, the phase of the received signal y (t) is rotated.

Therefore, OFDM receiver 10 according to the present embodiment has a function of compensating for phase rotation caused by carrier frequency offset Δf. Hereinafter, the configuration and operation of the OFDM receiver 10 having such a function will be described in detail.

(About configuration)
FIG. 3 is a block diagram schematically showing a circuit configuration of OFDM receiving apparatus 10 according to the present embodiment. FIG. 3 representatively shows a configuration necessary for restoring received data 1 and 2 from RF signals received by two antennas 11a and 11b among eight antennas 11a to 11h, respectively. ing.

  Referring to FIG. 3, OFDM receiving apparatus 10 includes a low noise amplifier (LNA) 12 (12a, 12b,...) For amplifying an RF signal received by each receiving antenna 11, and a local oscillation circuit 14 for generating a carrier. And a quadrature demodulator 13 (13a, 13b,...) For demodulating an RF signal based on a carrier and extracting an analog OFDM signal. In the present embodiment, only one local oscillation circuit 14 is provided in the analog RF circuit in order to make the frequency fluctuation occurring in each channel uniform. Similarly, in the OFDM transmitting apparatus 90, only one local oscillation circuit is provided. The local oscillation circuit 14 includes a crystal oscillator (OSC) 21, a phase locked loop (PLL) 22, and a voltage controlled oscillator (VCO) 23.

  The OFDM receiver 10 further includes a pair of A / D converters (ADCs) 15 (15a, 15b,...) For converting an analog OFDM signal into a digital OFDM signal, and digitally processing the converted digital OFDM signal. And a restoration processing circuit 30 (30a, 30b,...) For extracting received data. The restoration processing circuits 30 for the number of antennas 11 (the number of streams) are mounted on the digital reception circuit 16 as digital processing means. The digital receiving circuit 16 includes one oscillation circuit 24 that supplies a sampling clock to each A / D converter 15.

  The detailed functional configuration of the restoration processing circuit 30 is shown in FIG. Referring to FIG. 4, each restoration processing circuit 30 includes a synchronization unit 31, a GI removal unit 32, a phase rotation compensation unit 33, an FFT unit 34, a MIMO processing unit 35, a demapping unit 36, An interleaving section 37, a Viterbi decoding section 38, and a descrambling section 39 are included.

  The synchronization unit 31 receives a digital OFDM signal (transmission signal) from the A / D converter 15. Synchronizing section 31 synchronizes data between OFDM transmitting apparatus 90 and OFDM receiving apparatus 10 to determine the position of a symbol included in a digital OFDM signal. As shown in FIG. 5, the OFDM signal includes a preamble 51 and a plurality (for example, 20) of symbols 52. Each symbol 52 includes a GI (guard interval) section 61 as a redundant section and a data section 62. The signal of the GI section 61 is a redundant signal generated by copying the signal of the rear end 620 of the data section 62.

  The GI removing section 32 removes the GI section 61 from the symbol 52 and extracts a signal (valid OFDM signal) of the data section 62. The FFT unit 34 converts the effective OFDM signal into a data string by Fourier transform. That is, the effective OFDM signal of each symbol 52 is converted into a subcarrier signal, and channel equalization is performed. The functions of the GI removing unit 32 and the FFT unit 34 may be realized by a common circuit.

  The MIMO processing unit 35 performs MIMO detection and MIMO equalization. Specifically, based on the subcarrier signal obtained from the FFT unit 34, the propagation path is estimated by known MIMO detection, and a weight matrix is calculated. Further, the data signal is multiplied by the inverse matrix of the calculated weight matrix, and only necessary signals (transmission signals) are separated and extracted. That is, the MIMO processing unit 35 separates the subcarrier signal for each stream. This separated subcarrier signal is herein referred to as “separated subcarrier signal”. Note that the MIMO processing unit 35 is arranged across a plurality of restoration processing circuits 30 included in the digital reception circuit 16.

  The demapping unit 36 converts the arrangement of signal points on the complex plane, that is, the phase and amplitude information of the separated subcarrier signal into a digital bit string. The deinterleave unit 37 corrects data errors that occur during communication. The Viterbi decoding unit 38 decodes data encoded at the time of transmission based on the Viterbi algorithm. The descrambling unit 39 restores the data string that was randomly converted at the time of transmission, and extracts the received data. The received data extracted by the descrambling unit 39 is output from each restoration processing circuit 30.

  In the present embodiment, as described above, the restoration processing circuit 30 includes the phase rotation compensation unit 33. The phase rotation compensation unit 33 is arranged between the synchronization unit 31 and the FFT unit 34, and compensates for the phase rotation of the effective OFDM signal in the time domain. That is, the received signal y (t) shown in FIG. 2 is corrected.

  FIG. 11 is a diagram showing a difference between the output of the demapping unit 36 when there is no carrier frequency offset Δf and when there is a carrier frequency offset Δf.

  FIG. 11A shows an ideal output example when there is no offset Δf, and FIG. 11B shows an output example when there is an offset Δf. FIG. 11 plots the data values of all the separated subcarrier signals when one packet of data is QPSK-transmitted by the 8 × 8 MIMO method. Reception 1 to reception 8 represent transmission data of streams received by the antennas 11a to 11h, respectively. If one packet includes, for example, 20 symbols, as the transmission time series, reception 1 to reception 8 simultaneously receive symbol 1 data, symbol 2 data,...

  In the ideal output example of FIG. 11A, the plot value is constant for any transmission data, but if there is a carrier frequency offset Δf, it is affected by the phase rotation of the effective OFDM signal. The plot value changes as shown in the output example of FIG. If the data is restored with the output value as shown in FIG. 11B, correct received data cannot be extracted. In the output example of FIG. 11B, the plot points vary due to the influence of noise of the analog RF circuit.

  In the present embodiment, as the crystal oscillator 21 of the local oscillation circuit 14, a commercially available crystal oscillator having a relatively high performance is mounted, and the accuracy of the carrier frequency is improved to about 0.5 ppm. . When the frequency offset Δf in the analog RF circuit is increased to about 0.5 ppm, a slight frequency fluctuation due to the stability of the device becomes apparent.

  Such a small frequency fluctuation is a fluctuation smaller than the minimum step that can be adjusted in the analog RF circuit and does not cause any problem in the SISO system, but affects the data restoration rate in the MIMO system. For example, in the 8 × 8 MIMO system, it is required to correct the frequency offset to about 1/10 (0.05 ppm) of the frequency offset Δf improved by the analog RF circuit.

  The phase rotation compensator 33 of the present embodiment performs the following processing in order to reduce the influence of such a slight frequency fluctuation between the transmitter and the receiver (hereinafter also referred to as “dynamic frequency offset”).

First, the phase rotation compensator 33 calculates, for each symbol 52, the amount of phase rotation change θ _T of the entire symbol 52 using the signal of the GI unit 61. This calculation method will be described with reference to FIG.

As shown in FIG. 6, in each symbol 52, the GI section 61 includes guard interval data for T _GI points, and the data section 62 includes valid data for T _sym points. Assume as an example, the total number of points in the data portion 62 is _{160, T GI} point is 32 _points, and _{T sym} point is 128 points. Since the same data as the guard interval data is also included in the T _GI point at the rear end portion 620 of the valid data for the T _sym point, there is a certain data y _GI (t) in the GI section 61 and a T _{sym based on the} data. This is the same as the data y _data (t) after the point. Therefore, by comparing the data y _GI (t) with the data y _data (t), the change amount θ _T of the phase rotation of the effective OFDM signal in units of T _sym points (that is, the whole) for each symbol 52 is _calculated. It can be calculated.

Specifically, assuming that the comparison data y _GI (t) of the GI section 61 is s (t), the comparison data y _data (t) of the data section separated by T _SYM points is affected by the phase rotation. It is represented by the following equation (3).

The comparison data y _GI (t) and y _data (t) are complex numbers. When the comparison data y _GI (t) is multiplied by its complex conjugate y _GI (t) ^* , the signal d (t) after the multiplication is The rotation of the signal on the complex plane is obtained by the following equation (4).

The change amount θ _T of the phase rotation per T _sym for each symbol 52 is obtained as an average value of the rotation amounts d (t) of the streams 1 to 8. That is, the change amount θ _T of the phase rotation of one symbol as a whole is expressed by the following equation (5). From this equation (5), the carrier frequency offset Δf can be expressed as θ _T / 2πT _sym .

Since the guard interval data is data inserted to suppress the influence of interference between the symbols 52 due to multipath fading, data at the end of the symbol 52, that is, data near the boundary between consecutive symbols 52, is Poor. Therefore, in both the GI section 61 and the rear end section 620 of the data section 62, only the data at the center is used as comparison data. The data at the center is data excluding data at one point or more at both ends of each part, for example, data excluding data at eight points at both ends. In this case, the data at points 9 to 24 of the GI section 61 becomes the comparison data y _GI (t) of the GI section 61, respectively, and the data at points 137 to 152 at the rear end 620 of the data section 62 correspond to the data section 62, respectively. _Is the comparison data y _data (t). The data used for the comparison may be one point of data for both the GI section 61 and the rear end 620 of the data section 62.

Next, the phase rotation compensator 33 linearly approximates the change amount θ _T of the phase rotation of the entire symbol 52 to thereby obtain the phase shift θ (t) ′ of each signal y (t) of the data unit 62 shown in FIG. Is calculated. As described above, when the carrier frequency offset in the analog RF circuit is improved to about 0.5 ppm, the change in the frequency offset within one symbol 52 becomes gradual, so the phase rotation amount per t within one symbol 52 Can be regarded as θ _T / T _sym (T _sym = 128). Therefore, within one symbol 52, the phase shift θ (t) ′ of each signal y (t) of the data section 62 is expressed by the following equation (6).

  When attention is focused on only one symbol 52, each received signal y (t) of the data section 62 is reversely rotated by a phase shift θ (t) ′ to obtain a corrected received signal (hereinafter, “correction signal”). Y) (t) is obtained. That is, the received signals y (1) to y (128) of the data unit 62 are individually corrected for each signal as follows, for example.

  However, in the case of the MIMO wireless communication as in the present embodiment, it is not sufficient to simply correct the received signal y (t) using only the phase shift θ (t) ′ in the symbol 52. This is because the dynamic frequency offset affects over a plurality of symbols 52.

  Therefore, based on the phase shift θ (t) ′ for each signal calculated in each symbol 52, the phase rotation compensator 33 of the present embodiment calculates the reception signal y (t) within one symbol 52 and over a plurality of symbols 52. Is to be corrected. That is, the correction of the received signal y (t) of the next symbol 52 is performed following the correction of the received signal y (t) of the previous symbol 52 in units of packets.

In this case, the n-th phase shift is used for correction of the received signal _y n (t) of the symbol 52 theta _n (t) is, n-th time unit of a symbol 52 of the phase rotation amount θ _Tn / _{T sym,} it A value obtained by adding the amount of phase rotation of the symbol 52 before (first to (n-1) th) symbols. Such a phase shift θ _n (t) obtained by adding the phase rotation amount of the previous symbol 52 is herein referred to as “phase shift compensation amount θ _n (t)”.

Such tracking correction of the received signal y (t) will be specifically described with reference to FIG. The amounts of change in the phase rotation of the whole symbols (per 128 t) of the symbols 1, 2, 3,... Calculated by the GI method are θ _T1 , θ _T2 , θ _T3,. In this case, in each packet, the phase shift compensation amount θ ₁ (t) of the first symbol 1 is equal to the phase rotation amount (θ _T1 / T _sym ) · t per 1 t of the symbol 1 as in the above equation (6). It is calculated based on only That is, the phase shift compensation amount θ ₁ (t) of the symbol 1 is
It is expressed as Therefore, the correction of each data of the effective OFDM signal of the first symbol 1 in a certain packet is performed only within the symbol 1.

On the other hand, the phase shift compensation amount θ ₂ (t) of the symbol 2 following the symbol 1 is calculated by adding the phase rotation amount ((θ _T2 / T _sym ) · t) per t of the symbol 2 to the phase of the symbol 1 preceding the symbol 2. It is calculated by adding the amount of rotation ((θ _T1 / T _sym ) · 128). Therefore, the phase shift compensation amount θ ₂ (t) of the symbol 2 is
The phase shift θ ₂ (t) ′ in the symbol 2 is calculated by multiplying the phase shift compensation amount θ ₁ (128) of the previous symbol 1 as a whole.

Similarly, the phase shift compensation amount θ ₃ (t) of the symbol 3 next to the symbol 2 is calculated by adding the phase rotation amount ((θ _T3 / T _sym ) · t) per t of the symbol 3 to the symbol 1 before the symbol 1 2 ((θ _T1 / T _sym ) · 128) and (θ _T2 / T _sym ) · 128)). Therefore, the phase shift compensation amount θ ₃ (t) of the symbol 3 is
And is calculated by multiplying the phase shift compensation amount θ ₂ (128) of the entire previous symbol 2 by the phase shift θ ₃ (t) ′ in the symbol 3.

As described above, when correcting data (received signal) of t = 1 to T _sym in the symbol n, the phase shift compensation amount θn (t) is expressed by the following equation (7), and the correction signal y of each data is obtained. _n (t) _} is represented by the following equation (8). The correction signal y _n (t) _{} of} each data obtained in this way ideally corresponds to or approximates the original transmission signal x _n (t) transmitted from the OFDM transmitter 90.

Note that the calculation formula of Expression (7) is synonymous with the following Expression (9) using the carrier frequency offset Δf _n in the symbol n.

  In the graph of FIG. 9, the change of the phase rotation calculated by analysis in a certain packet is shown along the time axis. From the change in the slope of this graph, it can be seen that the carrier frequency offset Δf dynamically changes within one symbol 52 and over a plurality of symbols 52.

  As described above, according to the present embodiment, the signals in the data section are individually corrected using the phase shift compensation amount θ (t) within one symbol and over a plurality of symbols calculated for each symbol. Thereby, the phase rotation of a plurality of symbols is systematically compensated for each packet with reference to the first symbol 1. Therefore, it is possible to reduce distortion of the received signal due to slight fluctuation of the carrier frequency. As a result, the restoration rate of the MIMO-OFDM signal can be improved.

Since the average value of the rotation amounts d (t) of a plurality of streams is used as the phase rotation change amount θ _T of the entire symbol 52, the phase rotation change amount is calculated for each stream, and the received signal of each stream is calculated. The circuit scale can be reduced as compared with the correction.

(About operation)
The operation of OFDM receiving apparatus 10 according to the present embodiment will be described.

  FIG. 10 is a flowchart illustrating a received data restoration process performed by the digital receiving circuit 16 of the OFDM receiving apparatus 10 according to Embodiment 1 of the present invention.

  Referring to FIG. 10, synchronization section 31 determines the position of a symbol included in the received OFDM signal by timing synchronization (step S (hereinafter abbreviated as “S”) 2). Specifically, a plurality of symbols 52 included in the OFDM signal are sequentially cut out.

  The phase rotation compensator 33 extracts, for each symbol 52, the signal (part) of the GI part 61 and the signal (part) of the rear end 620 of the data part 62 that is the copy source (S4).

From the two signals extracted in S4, the phase rotation compensator 33 calculates the total phase rotation change amount θ _Tn for each symbol 52 based on the above-described equations (4) and (5) (S5). ). Further, for each symbol 52, a phase shift θ _n (t) ′ in units of time within the symbol 52 is estimated (calculated) by linearly approximating the change amount θ _Tn of the entire phase rotation (S6).

Thereafter, the phase rotation compensator 33 calculates the phase shift compensation amount θ _n (t) between the symbols 52 in accordance with the above equation (7) in order from the first symbol 52 (S7). That is, by multiplying the phase shift θ _n (t) ′ in the symbol 52 calculated in S6 by the phase shift compensation amount θ _n−1 (T _sym ) of the _immediately preceding symbol 52, the data portion 62 The phase shift compensation amount θ _n (t) within each symbol and between symbols is calculated for each signal.

  The GI remover 32 removes the GI part 61 in the symbol 52 and extracts only the effective OFDM signal of the data part 62 in parallel or in series with these processes of the phase rotation compensator 33 (S8). .

Subsequently, the phase rotation compensator 33 adds, for each symbol 52, the phase shift compensation amount θ _n (t) calculated in S7 based on the equation (8) to each signal of the data unit 62 extracted in S8. The effective OFDM signal is corrected by applying reverse rotation (S10). As a result, the signal of the data section 62 is corrected for tracking within one symbol and over a plurality of symbols according to the amount of phase rotation in units of packets and in units of time.

  Next, the FFT unit 34 performs a Fourier transform on the corrected signal of the data unit 62, thereby converting the effective OFDM signal into a subcarrier signal (subcarrier signal before stream separation) and performing channel equalization ( S12).

  After estimating the propagation path by MIMO detection (S14), the MIMO processing unit 35 performs MIMO equalization (S16). That is, only a necessary signal (transmission signal) is separated and extracted from the reception signal.

  The demapping unit 36 converts the signal point arrangement on the complex plane of the separated subcarrier signal output from the MIMO processing unit 35 into a digital bit sequence (S18).

  Thereafter, post-processing is performed by the deinterleaving unit 37, the Viterbi decoding unit 38, and the descrambling unit 39 (S20), and the received data is output.

  As described above, in the OFDM receiving apparatus 10 according to the present embodiment, the influence of the dynamic carrier frequency offset is reduced by the phase rotation compensator 33, so that the restoration rate of the finally extracted received data is increased. Can be.

<Embodiment 2>
The OFDM receiving apparatus according to Embodiment 2 of the present invention has a function of compensating the phase rotation of the received signal not only in the time domain but also in the frequency domain. Hereinafter, the OFDM receiving apparatus according to the present embodiment will be described in detail only with respect to differences from the first embodiment.

  FIG. 12 is a block diagram illustrating a circuit configuration example of the restoration processing circuit 30A included in the digital reception circuit 16 of the OFDM reception apparatus 10 according to the present embodiment. The restoration processing circuit 30A includes another phase rotation compensator 40 in addition to the configuration shown in FIG. The phase rotation compensating unit 40 is arranged between the MIMO processing unit 35 and the demapping unit 36. The phase rotation compensating unit 40 converts the phase rotation of the received signal that cannot be (or cannot be) processed by the time domain correction by the phase rotation compensating unit 33 into a frequency. Compensate in the area. That is, the phase rotation compensator 40 compensates at least the phase rotation of the received signal affected by the offset of the sampling frequency between the transmitter and the receiver in the frequency domain.

  Prior to the detailed description of the function of the phase rotation compensator 40, the sampling frequency offset and its influence will be described. As shown in FIG. 3, the OFDM receiver 10 includes an oscillation circuit 24 that supplies a sampling clock to the A / D converter 15. The OFDM transmitting apparatus 90 (FIG. 1) also includes an oscillation circuit (not shown) that supplies a sampling clock when D / A-converting a signal after IFFT (inverse Fourier transform). Unless both oscillation circuits are very high-performance, an offset occurs in the sampling frequency. The sampling frequency offset affects the received signal after FFT.

  Specifically, the presence of a sampling frequency offset has two effects: i) the length of the cut-out OFDM symbol expands and contracts; and ii) the cut-out position of the OFDM symbol shifts toward the latter half of the packet.

  The effect of i) causes inter-carrier interference as in the case of the carrier frequency offset Δf. The effect increases as the value obtained by dividing the offset frequency by the baseband bandwidth increases. However, since the baseband bandwidth is generally larger than the subcarrier frequency interval, the influence of the sampling frequency offset is smaller than the effect of the carrier frequency offset and can be ignored.

  On the other hand, the effect of ii) is particularly significant when a long packet is transmitted in a system that performs channel equalization using a preamble at the beginning of a packet, such as the IEEE 802.11 standard. This is because the shift of the cutout position increases as the number of symbols increases.

  As shown in FIG. 13, when a time shift occurs by Δt from the start point of the original symbol 52x in a certain symbol 52y, a time shift of the FFT occurs, and a difference Δt occurs in the symbol length. That is, the received signal Y (f) after the FFT undergoes a phase rotation from the original transmitted signal X (f) as follows.

  In this case, a phase rotation is applied to each separated subcarrier signal obtained by the MIMO processing. Then, an error occurs in the arrangement of signal points on the complex plane. If there is a sampling frequency offset (Δt), an erroneous phase rotation is applied to each stream (receptions 1 to 8). FIG. 16 shows a typical example of plot values of the separated subcarrier signals when there is a sampling frequency offset. As shown in FIG. 16, all streams of receptions 1 to 8 are affected by the frequency offset, so that erroneous phase rotation is applied to all streams.

  The phase rotation of the separated subcarrier signal obtained after the MIMO processing also occurs when there is a residual offset of the carrier frequency. This is because, when the carrier frequency offset that has not been completely corrected in the time domain remains, a shift of the convergence point due to the residual phase difference appears in the signal point arrangement on the complex plane.

  Therefore, there is a limit to improving the restoration rate of received data only by correcting the carrier frequency shown in the first embodiment. Therefore, in the present embodiment, the phase rotation compensator 40 digitally corrects the sampling frequency offset and the residual offset of the carrier frequency.

  The phase rotation compensator 40 linearly approximates (linearly approximates) the amount of change in phase rotation for each symbol 52y in the frequency domain based on the pilot subcarrier that is a known signal embedded in each symbol. The data subcarrier is corrected based on the data subcarrier. Note that the symbol 52y in the frequency domain is also referred to as a “frequency domain symbol” in distinction from the symbol 52 in the time domain. A specific processing example of the phase rotation compensator 40 will be described with reference to the flowchart of FIG.

  FIG. 14 is a flowchart showing the frequency domain correction processing executed in the present embodiment. This correction process is inserted between the MIMO equalization process (S16) and the demapping process (S18) in the received data restoration process shown in FIG.

  Referring to FIG. 14, phase rotation compensation section 40 divides the separated subcarrier signal output from MIMO processing section 35 into a data subcarrier (data signal) and a pilot subcarrier (pilot signal) (S32). ). Here, for example, it is assumed that the number of separated subcarrier signals output from MIMO processing section 35 is 114 per symbol 52y, and is divided into 108 data subcarriers and 6 pilot subcarriers. As an example, pilot signals are embedded in the 6, 34, 48, 67, 81, and 109th subcarriers. Note that the number of pilot subcarriers differs depending on the transmission bandwidth.

Next, the phase rotation compensator 40 calculates the amount (average) of the phase rotation by linear approximation for each frequency domain symbol 52y from the six pilot subcarriers of each stream (S34). Specifically, first, an average of streams 1 to 8 is obtained for each pilot subcarrier. That is, the symbol k (= 1, 2, 3, 4, 5, 6) is a pilot number, the symbol n (= 1, 2, 3, 4, 5, 6, 7, 8) is a stream number, and the symbol P _n ( Assuming that k) is a received pilot signal, the average P _AVG (k) of the k-th received pilot signal is represented by the following equation (10).

Next, an error is calculated by comparing the stream average of the k-th received pilot signal with a known pilot signal. In other words, as shown in the following equation (11), the stream average of the k-th received pilot signal is multiplied by the complex conjugate P _REF (k) ^{* of the} known pilot signal, and the phase-rotated signal P _d ( k) is calculated.

In this case, the average phase rotation change amount θ (k) is expressed as the following equation (12) as the argument of the calculated signal P _d (k).

  As shown in FIG. 15, the phase rotation compensator 40 linearly approximates the amount of phase rotation change in symbol units from the phase rotation amounts (average) of the six pilot signals. The slope of the linear expression that is linearly approximated represents the influence of the sampling frequency offset Δt in a certain symbol.

  When the linear expression is calculated for each symbol in this manner, the phase rotation compensator 40 calculates the phase shift θ (f) of the data subcarrier y (f) for each frequency domain symbol 52y in each stream. (S35). Symbol f (= 1, 2,... 108) indicates a data subcarrier number. Thereafter, the phase rotation compensator 40 corrects the data subcarrier y (f) in each stream according to the calculated phase shift θ (f) (S36). Specifically, reverse rotation is performed for all data subcarriers y (f) by the calculated phase shift θ (f) according to the following equation (13).

  In this way, the phase rotation compensator 40 creates a linear expression common to all streams for each frequency domain symbol 52y, and then calculates the phase difference (phase shift) between the linear expression and each data subcarrier. Then, within one symbol, the tracking correction of the data subcarrier is performed based on the calculated phase difference. Therefore, the optimal phase reverse rotation is performed for each frequency domain symbol 52y, so that the dynamically changing sampling frequency offset and carrier frequency residual offset can be appropriately digitally corrected.

  Also, in the frequency domain, since the average value of the rotation amounts of a plurality of streams is used as the phase rotation change amount θ (k) of the symbol 52y, the phase rotation change amount is calculated for each stream to receive each stream. The circuit scale can be reduced as compared with the case where the signal is corrected.

  As described above, in the present embodiment, since the phase rotation of the received signal is compensated not only in the time domain but also in the frequency domain, the change in the plot value of the demapping unit can be suppressed to the maximum. Therefore, the received data can be correctly extracted.

  In each embodiment, an example in which the number of streams is eight has been described. However, even when the number of streams is larger (for example, 16 × 16 MIMO scheme), deterioration of received data can be minimized. .

  Further, in each embodiment, the OFDM receiver 10 has been described as an example. However, a transmission signal to be received is a symbol formed of a redundant part and a data part as a transmission unit, and a signal of the redundant part is used in a time domain. As long as the apparatus can calculate the amount of change in the phase rotation of the entire symbol, a wireless receiver of a multicarrier transmission scheme other than the OFDM scheme may be used.

  Further, of the OFDM receiver 10 shown in each embodiment, only the function of the digital receiving circuit 16 may be provided as a received data restoring device for restoring received data. Such a received data restoring device restores received data by digitally processing a baseband transmission signal on a stream basis.

  The embodiments disclosed this time are to be considered in all respects as illustrative and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the description above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims.

  Reference Signs List 10 OFDM receiver (wireless receiver), 11 (11a to 11h) receive antenna, 13 quadrature demodulator, 14 local oscillator circuit, 15 A / D converter, 16 digital receiver circuit (received data recovery device), 21 crystal oscillator, Reference Signs List 22 phase synchronization circuit, 23 voltage controlled oscillator, 24 oscillation circuit, 30, 30A restoration processing circuit, 31 synchronization section, 32 GI removal section, 33, 40 phase rotation compensation section, 34 FFT section, 35 MIMO processing section, 36 demapping Unit, 37 deinterleave unit, 38 Viterbi decoding unit, 39 descramble unit, 90 OFDM transmitter, 91 (91a to 91h) transmission antenna.

Claims (6)

A receiving device used in a wireless communication system that performs multicarrier transmission by a MIMO method,
A plurality of receiving antennas for receiving a plurality of streams of transmission signals;
Demodulating means for demodulating an RF band signal received at each of the receiving antennas and outputting a base band signal;
Digital processing means for restoring received data by digitally processing the baseband transmission signal in units of the stream,
The digital processing means,
Extracting means for removing a signal of a redundant part and extracting a signal of a data part for each symbol constituting the transmission signal;
Estimating means for estimating a time-based phase rotation amount of the data portion by linearly approximating a change amount of a phase rotation of the entire symbol calculated using the signal of the redundant portion for each symbol,
The signal of the data section is individually corrected for each signal based on the phase rotation amount in time unit estimated by the estimating means, and the signal of the data section is divided into one symbol and a plurality of symbols in packet units. Correction means for performing tracking correction;
A conversion unit that converts the signal of the data portion corrected by the correction unit into a subcarrier signal,
The correction means calculates, in each symbol, a phase shift compensation amount of each signal of the data portion by adding a phase rotation amount of a previous symbol to the phase rotation amount in time units, and calculates the calculated phase shift amount. A wireless receiving apparatus for correcting a signal of the data part by applying a reverse rotation to each signal of the data part by a compensation amount . The phase shift compensation amount θn (t) for the symbol n is
The wireless receiving device according to claim 1 , wherein the value is calculated as: The digital processing unit further includes a unit that calculates a change amount of a phase rotation of the entire symbol as an average value of a rotation amount calculated using a signal of the redundant unit for each of the plurality of streams. 3. The wireless receiving device according to 1 or 2 . The digital processing means,
MIMO processing means for separating a subcarrier signal obtained from the conversion means for each stream and converting the separated subcarrier signal into a separated subcarrier signal;
For each frequency domain symbol, linear approximation means for linearly approximating the amount of change in phase rotation based on pilot subcarriers included in the separated subcarrier signal, and calculating a linear expression,
Based on the linear equation calculated by the linear approximation unit, for each of the frequency domain symbols, calculate a phase shift of each data subcarrier included in the separated subcarrier signal, and calculate the phase shift and the data subcarrier. The radio receiving apparatus according to any one of claims 1 to 3 , further comprising: a subcarrier correction unit configured to reversely rotate the subcarriers. The radio receiving apparatus according to claim 4 , wherein the linear approximation unit calculates a linear expression for each of the frequency domain symbols using an average value of the plurality of streams for each of the pilot subcarriers. An apparatus for restoring received data by digitally processing a transmission signal of a baseband in a stream unit,
Extracting means for removing a signal of a redundant part and extracting a signal of a data part for each symbol constituting the transmission signal;
Estimating means for estimating a time-based phase rotation amount of the data portion by linearly approximating a change amount of a phase rotation of the entire symbol calculated using the signal of the redundant portion for each symbol,
The signal of the data section is individually corrected for each signal based on the phase rotation amount in time unit estimated by the estimating means, and the signal of the data section is divided into one symbol and a plurality of symbols in packet units. Correction means for performing tracking correction;
A conversion unit that converts the signal of the data portion corrected by the correction unit into a subcarrier signal,
The correction means calculates, in each symbol, a phase shift compensation amount of each signal of the data portion by adding a phase rotation amount of a previous symbol to the phase rotation amount in time units, and calculates the calculated phase shift amount. A received data restoring device for correcting a signal of the data part by applying reverse rotation to each signal of the data part by a compensation amount .