JP6624729B2 - Insulated bidirectional DC / DC converter and control method therefor - Google Patents

Description

  The present invention relates to an isolated bidirectional DC / DC converter.

  As electrical insulation and voltage matching means, an insulated bidirectional DC / DC converter using a small and lightweight high-frequency transformer (hereinafter simply referred to as a transformer) has attracted attention. The bidirectional DC / DC converter is characterized by a circuit configuration in which two single-phase full-bridge circuits (also called H-bridge circuits) are connected via a transformer 102, and is suitable for high-power applications (Non-Patent Document 1). ~ 3). The bidirectional DC / DC converter is also called a DAB (Dual-Active-Bridge) converter because of its circuit configuration. The DAB converter generates a square wave voltage having a duty ratio of 50% in the primary winding and the secondary winding of the transformer by simultaneously switching the diagonal switches of each bridge circuit. The transmission power can be adjusted by controlling the phase difference between the bridge circuits. In this case, each bridge can perform a zero-voltage switching (ZVS) operation, and high-efficiency power transmission can be realized. However, in a low output region, there is a problem that switching loss (snubber loss) associated with incomplete ZVS operation increases and conversion efficiency decreases (Non-Patent Documents 3 and 4).

  There has also been proposed a combination of pulse-width-modulation (PWM) control with a DAB converter (Non-patent Documents 5 and 6). It is intended to adjust the transmission power while reducing the value. This is intended for a small-capacity DAB converter, which is an imperfect ZVS operation in a low output region, and is a concept in which controllability is prioritized over loss reduction.

  There is an intermittent operation as a power adjustment technique that emphasizes loss reduction in a low output region. This is to provide a power transmission suspension period to adjust transmission power on average, and is also called a burst mode (Non-Patent Documents 7 and 8). Intermittent operation itself has been known for a long time. For example, intermittent operation is applied to an induction heating device or a corona discharge treatment device having a resonance type inverter to realize a stable operation in a low output region and a reduction in loss (Non-Patent Documents 9 and 10). In such an intermittent operation, since the resonance frequency is used as a basic cycle, power adjustment is discrete, and is also called pulse density modulation (PDM: Pulse-Density-Modulation). During the power transmission period, the operation is performed in the output region where the ZVS operation is possible, so that the snubber loss can be significantly reduced. Further, since no iron loss occurs during the power transmission suspension period, a significant improvement in efficiency in the low output region can be expected.

RWD Doncker, DM Divan, MH Kheraluwala: “A three-phase soft-switched high-power-density dc / dc converter for high-power applications,” IEEE Trans. Ind. Appl., Vol. 27, No. 1, pp . 63-73, (1991-1) MH Kheraluwala, RW Gascoigne, DM Divan, ED Baumann: “Performance characterization of a high-power dual active bridge dc-todc converter,” IEEE Trans. Ind. Appl., Vol. 28, No. 6, pp. 1294-1301 , (1992-11) S. Inoue, H. Akagi: “Operating voltage and loss analysis of a bidirectional isolated dc / dc converter,” IEEJ Trans. Ind. Appl., Vol. 127, No. 2, pp. 189-197, (2007-5 ) T. Yamagishi, H. Akagi: “A 750-V, 100-kW, 20-kHz bidirectional isolated dc / dc converter using SiC-MOSFET / SBD modules,” IEEJ Trans. Ind. Appl., Vol. 134, No. 5, pp. 544-553, (2014-5) A. K. Jain, R. A. Ayyanar: “PWM control of dual active bridge: comprehensive analysis and experimental verification,” IEEE Trans. Power Electron., Vol. 26, No. 4, pp. 1215-1227, (2011-4) G. G. Oggier, G. O. Garcia, A. R. Oliva: “Switching control strategy to minimize dual active bridge converter losses,” IEEE Trans. Power Electron., Vol. 24, No. 7, pp. 1826-1838, (2009-7) A. Rodriguez, A. Vazquez, DG Lamar, MM Hernando, J. Sebastian: “Different purpose design strategies and techniques to improve the performance of a dual active bridge with phase-shift control,” IEEE Trans. Power Electron., Vol. 30, No. 2, pp. 790-804, (2015-2) G. G. Oggier, M. Ordonez: “High-Efficiency DAB Converter Using Switching Sequences and Burst Mode,” IEEE Trans. Power Electron., Vol. 31, No. 3, pp. 2069-2082, (2016-3) H. Fujita, H. Akagi: “Pulse-density-modulated power control of a 4 kW, 450 kHz voltage-source inverter for induction melting applications,” IEEE Trans. Ind. Applicat., Vol. 32, No. 2, pp. .279-286, Mar / Apr 1996 H. Fujita, H. Akagi: “Control and performance of a pulse-density modulated series-resonant inverter for corona discharge processes,” IEEE Trans. Ind. Applicat., Vol. 35, No. 3, pp. 621-627, May / Jun 1999

  The inventor has come to recognize the following problem when applying intermittent operation to a DAB converter having a large capacity of several hundred kW.

  Non-Patent Documents 7 and 8 are directed to a small-capacity DAB converter of about 1 kW. If the knowledge obtained therefrom is applied as it is to a large-capacity converter of several tens of kW to several hundreds of kW, high power at a certain transmission power is obtained. Efficiency may be obtained, but efficiency at other transmit powers will be significantly reduced.

  In the combination of the DAB converter and the intermittent operation, there are a plurality of degrees of freedom in adjusting the transmission power. However, the prior arts (for example, Non-Patent Documents 7 and 8) do not effectively utilize the degrees of freedom, and particularly have a low degree of freedom. There is room for further improvement in efficiency at output.

  The present invention has been made in view of such a problem, and one of exemplary purposes of one embodiment of the present invention is to provide an isolated bidirectional DC / DC converter capable of obtaining high efficiency in a wide range of transmission power.

One embodiment of the present invention relates to an isolated bidirectional DC / DC converter. The bidirectional DC / DC converter includes a transformer, a first full bridge circuit connected to a primary winding of the transformer, a second full bridge circuit connected to a secondary winding of the transformer, and a first full bridge circuit. And a controller for controlling the second full bridge circuit. The controller performs the continuous operation of switching the first full-bridge circuit and the second full-bridge circuit with the phase difference δ in the first region where the transmission power is higher than the predetermined threshold value P _TH to change the phase difference δ. By adjusting the transmission power in the second region where the transmission power is lower than the threshold value P _TH , the phase difference δ _FIX is fixed, m is a real number that is a natural number multiple of 0.5, n is a real number, and δ _{FIX is} By switching the first full-bridge circuit and the second full-bridge circuit for a period of + 2πm, and performing an intermittent operation of stopping the switching of the first full-bridge circuit and the second full-bridge circuit for a period of 2πn, thereby changing n. Adjust the transmission power. Threshold P _TH, when P ₁ a transmission power which provides the maximum efficiency (maximum efficiency point) in continuous _operation, the transmission power that gives the minimum loss in continuous operation (minimum loss point) was P _2, P ₂ ≦ P _TH ≦ P ₁ × 1.1 is satisfied.

The inventor has obtained the knowledge that, when the continuous operation of the phase difference control is performed, if the transmission power falls below the maximum efficiency point, or if the transmission power falls below the minimum loss point, the efficiency starts to significantly decrease. Was. By defining the threshold value P _TH so as to satisfy P ₂ ≦ P _TH ≦ P ₁ × 1.1, a high output region exceeding several tens to several hundreds kW to a low output region on the order of several kW or less. High efficiency can be obtained in a wide power range including

The phase difference [delta] _FIX used in the second region, the phase difference [delta] ₁ corresponding to the transmission power P _1, may be included between the phase difference [delta] ₂ corresponding to the transmission power P _2.

The phase difference δ _FIX used in the second region may be a phase difference δ corresponding to the transmission power P _TH .

The threshold value P _TH may be substantially equal to P _2, and the phase difference δ used in the second region may be substantially equal to the phase difference δ ₁ corresponding to P ₁ . This can further reduce losses and improve efficiency.

  According to an aspect of the present invention, the efficiency of the isolated bidirectional DC / DC converter can be improved.

FIG. 2 is a circuit diagram of a bidirectional DC / DC converter according to the embodiment. FIG. 2 is an equivalent circuit diagram of primary-side conversion of the bidirectional DC / DC converter of FIG. 1. FIG. 3 is an operation waveform diagram of the equivalent circuit diagram of FIG. 2 in a second mode. FIG. 3 is a circuit diagram of a bidirectional DC / DC converter used in an experiment. It is a figure showing the constant used for the experiment. It is an operation | movement waveform diagram at the time of continuous operation of rated output P = 100kW. It is an operation | movement waveform diagram at the time of the continuous driving | operation of output power P = 45kW. It is an operation | movement waveform diagram at the time of intermittent driving | operation at output power _Pint = 10kW. It is an operation | movement waveform diagram at the time of intermittent driving | operation at output power _Pint = 10kW. FIGS. 10A and 10B are diagrams showing the relationship between the output power P and the loss P _loss . FIGS. 11A and 11B are diagrams showing the relationship between the output power P and the efficiency η. In _{P con = 41kW (δ = 8.8} °) and _{P con = 19kW (δ = 4} °), it is a diagram showing the relationship between the output power _P int _(1~18kW) and n. Output power _P int and (1~18kW) is a diagram showing the relationship between voltage ripple? Vp-p.

  Hereinafter, the present invention will be described based on preferred embodiments with reference to the drawings. The same or equivalent components, members, and processes shown in each drawing are denoted by the same reference numerals, and the repeated description will be omitted as appropriate. In addition, the embodiments do not limit the invention but are exemplifications, and all features and combinations thereof described in the embodiments are not necessarily essential to the invention.

  In this specification, “the state in which the member A is connected to the member B” refers to the case where the member A and the member B are physically directly connected to each other. Indirect connection via another member that does not substantially affect the basic connection state or impair the function or effect provided by the combination thereof. Similarly, “the state in which the member C is provided between the member A and the member B” means that the member A and the member C or the member B and the member C are directly connected, Indirect connection via another member that does not substantially affect the basic connection state or impair the function or effect provided by the combination thereof.

  In this specification, electric signals such as voltage signals and current signals, or symbols attached to circuit elements such as resistors and capacitors denote the respective voltage values, current values, or resistance values and capacitance values as necessary. Shall be represented.

(Operating principle)
FIG. 1 is a circuit diagram of the bidirectional DC / DC converter according to the embodiment. The bidirectional DC / DC converter 100 includes a transformer 102, a first full bridge circuit 104, a second full bridge circuit 106, and a controller 110.

The transformer 102 has a primary winding W1 and a secondary winding W2. The turn ratio between the primary winding W1 and the secondary winding W2 is N: 1. The AC terminal of the first full bridge circuit 104 is connected to the primary winding W1 of the transformer 102 via the inductor _La1 , and the AC terminal of the second full bridge circuit 106 is connected to the primary winding W2 of the transformer 102 via the inductor _La2. Connected to next winding W2.

  The first full bridge circuit 104 includes first to fourth switches SW11 to SW14. A snubber capacitor C1x is provided in parallel with each switch SW1x (x = 1, 2, 3, 4). Similarly, the second full bridge circuit 106 includes a first switch SW21 to a fourth switch SW24, and a snubber capacitor C2x is provided in parallel with each switch SW2x. As the switch SW, a transistor such as a MOSFET (Metal Oxide Semiconductor Field Effect Transistor) or an IGBT (Insulated Gate Bipolar Transistor) can be used. The type of the transistor element may be selected based on the required withstand voltage and the rated capacity. A freewheel diode (flywheel diode) is required in parallel with the switch SW so that the cathode is on the high potential side. When a MOSFET is used, its body diode can be used as a freewheel diode.

The capacitor C1 is connected to the DC terminal side of the first full bridge circuit 104, and the capacitor C2 is connected to the DC terminal side of the second full bridge circuit 106. The bidirectional DC / DC converter 100 is capable of transmitting power bidirectionally. However, in the following, for convenience of description, the power supply 200 is connected to the first full bridge circuit 104 as an input, and the second full bridge circuit It is assumed that the load 202 is connected with the output on the 106 side. The voltage of the power supply 200 is E ₁ and the voltage generated at the load 202 is E ₂ . When operating in the reverse direction, the input and output may be switched in the following description.

The controller 110 controls the switches SW11 to SW14 of the first full bridge circuit 104 and the switches SW21 to SW24 of the second full bridge circuit 106. Specifically, with respect to the first full bridge circuit 104, the controller 110 determines whether the diagonally arranged pairs SW11 and SW14 are on at a predetermined frequency (switching frequency f _sw ) and the diagonally arranged pair SW11 and SW14. And the state in which the pair SW12 and SW13 are ON are alternately repeated at a duty ratio of 50%. At this time, the voltage _vac1 of the AC terminal of the first full bridge circuit 104 becomes a rectangular wave having a switching frequency f _SW and a duty ratio of 50%. The same applies to the second full-bridge circuit 106. At the switching frequency f _sw , the controller 110 switches the diagonally arranged pair SW21 and SW24 to the ON state, and sets the diagonally arranged paired SW22 and SW23 to another pair. The ON state and the ON state are alternately repeated at a duty ratio of 50%. At this time, the voltage _{v ac2} the AC terminal of the second full bridge circuit 106 is also a switching frequency _{f SW,} a duty ratio of 50% of the rectangular wave.

FIG. 2 is an equivalent circuit diagram of primary-side conversion of the bidirectional DC / DC converter 100 of FIG. Wiring resistance 1, excitation inductance of the transformer 102, transformer 102 and the iron loss of the external inductor _{L a1, L a2,} ignoring copper loss, transformer 102 and inductor _{L a1, L a2} as single inductor L expressed. This inductor L represents a combined inductance of the primary-side converted inductance (L _a1 + N ² _{La 2} ) of the inductors La ₁ and _{La 2 in} FIG. 1 and the leakage inductance 1 of the transformer 102.
L = (L _a1 + N ² L _a2 ) +1
In other words, the value of the inductance L so that the appropriate value _{L opt,} inductor _{L a1, L a2} and transformer 102 of the external is designed. In general, it is possible to l is the transformer 102 is designed to be smaller, designing _{L a1, L a2} as appropriate _{L opt} is obtained. Note that when the transformer 102 can be designed so that L _opt = 1, the external inductors La ₁ and _{La 2} can be omitted.

In FIG. 1, the voltages E ₁ and E ₂ are constant, the switch SW is regarded as an ideal switch, and there is no switching delay and no dead time. At this time, the first full bridge circuit 104 and the second full bridge circuit 106 are represented as ideal voltage sources 204 and 206 that generate a rectangular wave AC voltage having a switching frequency f _SW and a duty ratio of 50%, respectively. The voltage source 204 applies an AC rectangular voltage having an amplitude _{vac1 to} one end of the inductor L. The voltage source 206 applies, to the other end of the inductor L, an AC rectangular voltage having an amplitude N × _vac2 . N is the turns ratio of the transformer 102.

Returning to FIG. 1, the control of the transmission power will be described. The controller 110 can switch between two operation modes. The first mode is selected in a first region where the transmission power (output power) P is higher than a predetermined threshold P _TH , and the second mode is selected in a second region where the transmission power P is lower than the threshold P _TH. Is done. Hereinafter, the operation of each mode will be described.

(1st mode)
In the first mode, the first full bridge circuit 104 and the second full bridge circuit 106 are switched by the phase difference δ. That is, the AC voltage _{v ac2} of the AC voltage _{v ac1} the primary side of the transformer 102 and the secondary side, the phase difference becomes a rectangular wave of [delta]. In the first mode, continuous operation is performed, and the transmission power is adjusted by changing the phase difference δ. The transmission power P _con in the first mode is given by equation (1).

(2nd mode)
In the second mode, an intermittent operation is performed. In the second mode, the phase difference δ between the first full bridge circuit 104 and the second full bridge circuit 106 is fixed to a certain value δ _FIX . When one cycle of switching is 2π, the parameters m and n are real numbers, and the first full-bridge circuit 104 and the second full-bridge circuit 106 are switched for a period of δ _FIX + 2πm. Switching between the full bridge circuit 104 and the second full bridge circuit 106 is stopped. FIG. 3 is an operation waveform diagram of the equivalent circuit diagram of FIG. 2 in the second mode. Here, E <NE ₂ is made up. Since the transmission period is δ _FIX + 2πm and the rest period is 2πn, one cycle of the intermittent operation is 2π (m + n) + δ _FIX .

The parameter m needs to be a natural number multiple of 0.5 (m = 0.5, 1.0, 1.5...) To prevent magnetic saturation. When m is a non-integer (m = 0.5, 1.5, 2.5,...), the waveform of the transmission period is separated by half waves. For example, if m = 0.5, a positive half wave is generated during a certain transmission period, and a negative half wave is generated during the next transmission period. In the second mode, the transmission power is adjusted by changing the parameter n. The transmission power P _int in the second mode is represented by Expression (2). n does not need to be a natural number multiple of 0.5, and may be any real number.

The boundary (threshold P _TH ) between the first region operating in the first mode and the second region operating in the second mode is defined as follows. Equation (3) holds when the transmission power (also referred to as the maximum efficiency point) that gives the maximum efficiency in continuous operation is P ₁ and the transmission power (the minimum loss point) that gives the minimum loss in continuous operation is P ₂ .
P ₂ ≦ P _TH ≦ P ₁ × 1.1 (3)

The fixed value δ _FIX of the phase difference δ used in the second mode is a phase difference δ ₁ corresponding to the maximum efficiency point P ₁ during continuous operation and a phase difference δ ₁ corresponding to the minimum loss point P ₂ during continuous operation. It included in the range between [delta] _2.

  The above is the configuration and operation principle of the bidirectional DC / DC converter 100. According to the bidirectional DC / DC converter 100, high efficiency can be obtained in a wide power range from a high output region exceeding several tens to several hundred kW to a low output region of several kW order or less.

That is, when combining the DAB converter and the intermittent operation, there are a plurality of degrees of freedom in adjusting the transmission power. Specifically, which mode is used in which power region (threshold P _TH for switching the mode), the phase difference δ used in the second mode, and the like exist as design parameters, and combinations thereof are various. Provides freedom. According to the present embodiment, by appropriately selecting the threshold value P _TH , it is possible to suppress a decrease in efficiency particularly at the time of low output and improve the efficiency.

Hereinafter, experiments performed by the present inventors to verify the usefulness of the present invention will be described. FIG. 4 is a circuit diagram of the bidirectional DC / DC converter used in the experiment. Four 1.2 kV, 400 A SiC-MOSFET modules were used as the switches SW of the transformer 102 and the first full bridge circuit 104. This module does not include an SBD (Schottky Barrier Diode), and utilizes a parasitic pn diode between the source and drain of the SiC-MOSFET as a freewheeling diode. In this experimental system, the DC terminal of the second full bridge circuit 106 (the output terminal of the bidirectional DC / DC converter 100) is connected to the DC terminal of the first full bridge circuit 104 (the input terminal of the bidirectional DC / DC converter 100). By connecting, the output power P is regenerated to the DC power supply E side (input side). As a result, the DC power supply E on the input side supplies the power loss P _loss corresponding to the DC / DC converter, so that the 100 kW rated operation is possible even with the DC power supply in the university laboratory. Further, the loss P _loss of the bidirectional DC / DC converter 100 can be directly measured with high accuracy as the output power of the DC power supply E. By separately measuring the output power P and the power loss P _loss , , The power conversion efficiency η of the DC / DC converter can be calculated with high accuracy by equation (4).
η = P / (P + P _loss ) (4)
In this specification, since the power loss of only the main circuit is discussed, the power loss of the controller 110 (that is, the gate drive circuit and the control circuit) in FIG. 1 is not included.

ただし、Ｉ_ｓｗ１は第１フルブリッジ回路１０４のスイッチング時のインダクタ電流である。ダイオードの導通損失が発生するのはＴ_ｄ＞Ｔ_１の場合であり、第１フルブリッジ回路１０４側のダイオードの導通損失Ｐ_{ｄｉｏｄｅ１}は、式（６）となる。

ただし、Ｖ_ｆは寄生ｐｎダイオードの順方向電圧降下である。 When a MOSFET is used for the switch element SW, synchronous rectification can be employed. As a result, there is no need for a free wheel diode connected in anti-parallel to the MOSFET. The forward voltage of the parasitic pn diode of the SiC-MOSFET is higher than that of the Si-MOSFET. However, due to the adoption of synchronous rectification, the conduction period of the parasitic pn diode becomes shorter than the dead time (set to 0.6 s in the experiment). When transmitting electric power from the first full bridge circuit 104 to the second full-bridge circuit 106, after the dead time start of the first full bridge circuit 104, the time T ₁ required until the parasitic pn diode is conducting, the formula (5) Is represented by

Here, I _sw1 is an inductor current at the time of switching of the first full bridge circuit 104. The conduction loss of the diode occurs when T _d > T ₁ , and the conduction loss P _diode1 of the diode on the side of the first full bridge circuit 104 is represented by Expression (6).

Here, _Vf is a forward voltage drop of the parasitic pn diode.

ただし、Ｉ_ｓｗ２は第２フルブリッジ回路１０６のスイッチング時のインダクタ電流であり、Ｐ_{ｄｉｏｄｅ２}は第２フルブリッジ回路１０６側で生じるダイオードの導通損失である。 Equations (7) and (8) are similarly obtained for the second full bridge circuit 106.

Here, I _sw2 is an inductor current at the time of switching of the second full bridge circuit 106, and P _diode2 is a conduction loss of the diode generated on the second full bridge circuit 106 side.

FIG. 5 is a diagram showing constants used in the experiment. Using these constants, the conduction loss of the parasitic pn diode during rated operation (100 kW) was calculated from equations (5) and (7). Here, _Vf was set to 3.1V. As a result, the conduction loss is about 45 W, and the conduction loss due to the parasitic pn diode can be practically ignored. Since the ZVS operation is performed and no reverse recovery current flows through the parasitic pn diode, only the conduction loss is considered.

  Next, the results of the experiment will be described. In this experiment, power transmission from the first full bridge circuit 104 to the second full bridge circuit 106 was performed. In the following discussion, it is assumed that power is transmitted from the first full bridge circuit 104 to the second full bridge circuit 106. FIG. 6 is an operation waveform diagram at the time of continuous operation with the rated output P = 100 kW. FIG. 7 is an operation waveform diagram at the time of continuous operation with output power P = 45 kW. At outputs of 100 kW and 45 kW, the phase differences were δ = 27.4 ° and δ = 11 °.

FIGS. 8 and 9 are operation waveform diagrams at the time of intermittent operation at output power P _int = 10 kW. In FIG. 8, 9, have different values [delta] _FIX retardation [delta], in particular in FIG. _8, using the _{P con = 41kW δ FIX (=} 11 °), Fig. _{9, P con} Δ _FIX (= 3.3 °) where = 19 kW is used.

FIGS. 10A and 10B are diagrams showing the relationship between the output power P and the loss P _loss . FIGS. 11A and 11B are diagrams showing the relationship between the output power P and the efficiency η. FIGS. 10B and 11B are enlarged views of the range of 0 kW to 40 kW in FIGS. 10A and 11A, respectively. Here, equation (4) was used to calculate the conversion efficiency η. In the continuous operation, the maximum efficiency η _max = 98.8% was reached at the output power P = 41 kW. That is, in the case of the circuit constant used in the experiment, the maximum efficiency point P ₁ is 41 kW. The conversion efficiency was 98% at the time of rated operation with the output power P = 100 kW. At the time of continuous operation, there is a region where the power loss P _loss increases when the power P is further reduced after the output power P = 19 kW. That is, when the circuit constants used in the experiment, the minimum loss point _{P 2} is 19 kW. In this region, the snubber loss due to the imperfect ZVS operation is dominant.

In the intermittent operation, at an actually measured output power of 3 kW or more, η = 98.0% when P _con = 41 kW, and 97% when P _con = 19 kW. _Furthermore, when comparing the P con = 41kW and _P con = 19 kW, towards _P con = _41kW at full power regions measured is compared with the _P con = 19 kW, it can be seen that further reduce power loss.

  Referring to FIGS. 11 (a) and 11 (b). Comparing the conversion efficiency between continuous operation and intermittent operation, it is possible to greatly improve efficiency by applying intermittent operation to a low output region (19 kW or less) where snubber loss is dominant in continuous operation (conventional control). You can see that. This is because the snubber loss can be reduced and the iron loss during the idle period can be reduced to zero.

On the other hand, when the difference between P _int and P _con is equal to or smaller than a certain value, that is, when the cycle of the intermittent operation is shorter than a certain value, the efficiency of the intermittent operation is lower than that of the continuous operation. As shown in FIGS. 8 and 9, in the intermittent operation, the phase advance bridge at the start of transmission and the phase delay bridge at the end of transmission perform a hard switching operation. In order to control the inductor current during the idle period to zero, a hard switching operation occurs in principle. Therefore, when the cycle of the intermittent operation is shortened, the influence of the snubber loss caused by the hard switching operation increases, and as a result, the efficiency of the intermittent operation deteriorates.

The following findings are obtained by summarizing the threshold value P _TH and the phase difference δ _FIX based on these experimental results.

(First setting method)
Figure 11 (b), the most efficient that good in a wide power range, at higher than the minimum loss point _P 2 = 19 kW region, is operated in the first mode, the lower the minimum loss point _P 2 = 19 kW region , P _con = 41 kW, and operated in the second mode using the phase difference δ _FIX . That is, the threshold value P _TH is substantially equal to P _2, and a value substantially equal to the phase difference δ ₁ corresponding to the maximum efficiency point P ₁ is used as the phase difference δ used in the second mode. , High efficiency can be maintained.

(Second setting method)
Referring also to FIG. 11 (b), at higher than the minimum loss point _P 2 = 19 kW region, is operated in the first mode, the lower the minimum loss point _P 2 = 19 kW _regions, a phase difference provided the P con = 19kW δ _{Even when the FIX} is used to operate in the second mode, a high efficiency of 97% or more is maintained in a wide power range. That is, the threshold value P _TH is substantially equal to P _2, and a value substantially equal to the phase difference δ ₂ corresponding to the minimum loss point P ₂ may be used as the phase difference δ used in the second mode. .

(Third setting method)
Even in an intermediate state between the first setting method and the second setting method, an intermediate efficiency between them can be obtained. Therefore, the threshold value P _TH is substantially equal to P _2, and high efficiency can be maintained by using a value between the phase differences δ ₁ and δ ₂ as the phase difference δ used in the second mode. it can.

(Fourth setting method)
Referring to FIG. 11B, even when the intermittent operation is performed in a range exceeding 19 kW using the phase difference δ ₁ that gives P _con = 41 kW, a high efficiency of about 98% is maintained. Similarly, even when using phase difference [delta] ₂ to give P _{con =} 19 kW, even when performing the intermittent operation in a range in excess of 19 kW, as high as 97% efficiency is maintained. This tendency is maintained up to the point where P ₁ = 41 kW is exceeded (a value about 10% higher than P ₁ according to the study of the present inventors). This is the basis for determining the threshold value P _{TH based} on the above-described equation (3).

When combined with the knowledge in the third setting method, the threshold value P _TH is defined so as to satisfy Expression (3), and the phase difference δ used in the second mode is defined as the phase difference δ between the phase differences δ ₁ and δ ₂ . By using the value of, high efficiency can be maintained.

  Next, the parameter m will be discussed. Regarding how to determine the transmission period and the idle period in the intermittent operation of the second mode, there is a degree of freedom under the constraint of preventing magnetic saturation of the high-frequency transformer. The transmission period determination method of Non-Patent Document 8 does not have the concept of m in this specification, and attempts to cancel the magnetic flux of the high-frequency transformer with two positive and negative pulse periods. As a result, a total of six hard switching operations occur in two pulse periods, resulting in a decrease in conversion efficiency. On the other hand, if m = 1 is selected, the hard switching operation can be suppressed to twice.

  When the transmission period determination method of this specification is adopted, the minimum value of m in consideration of prevention of magnetic saturation is 0.5. However, when the transmission power is the same, the conversion efficiency is improved by setting m to be large from the viewpoint of reducing the snubber loss. On the other hand, when m is increased, the voltage ripple described below becomes a problem. As described above, considering the trade-off between the improvement of the conversion efficiency and the reduction of the voltage ripple, it is preferable to set m = 1.

Next, the voltage ripple caused by the intermittent operation will be discussed. In the intermittent operation, power transmission is divided into a transmission period and a pause period, so that voltage ripple occurring in the DC capacitor Cdc becomes a problem. In idle period _{nT sw (= n / f sw} ), when the DC capacitor Cdc is assumed to be supplied to a load a constant current _{I 2,} respectively voltage ripple Delta] v _p-p and the current _{I 2} is generated in the _{C dc,} formula (9 ) And (10).

FIG. 12 is a diagram showing the relationship between the output power P _int (1 to 18 kW) and n when P _con = 41 kW (δ = 8.8 °) and P _con = 19 kW (δ = 4 °). FIG. 13 is a diagram illustrating a relationship between the output power P _int (1 to 18 kW) and the voltage ripple Δv _pp . Each constant uses the same value in FIG. The turns ratio of the transformer is 1: 1 (that is, N = 1), and E ₁ = E ₂ = 750 _Vdc .

From the result of the theoretical analysis, even when the electrostatic constant of the DC capacitor is set to 1.1 ms (400 μs), the voltage ripple caused by the intermittent operation is not a practical problem. The reason for this is that the intermittent operation is applied only to the low output region. Specifically, in the output region of 1 to 18 kW, the ripple Δv _pp when P _con = 41 kW is 6.6 V or less (0.88% with respect to 750 V _dc ), and P _con = 19 kW. In this case, the ripple Δv _pp becomes 3.0 V (0.4% of 750 V _dc ) or less.

  In the embodiment, the DAB including the single-phase full bridge circuits 104 and 106 has been described. However, the present invention is not limited thereto, and the present invention is applicable to a DAB including a three-phase full bridge circuit and a three-phase transformer.

  Although the present invention has been described using specific terms based on the embodiments, the embodiments are merely illustrative of the principles and applications of the present invention, and the embodiments are defined in the claims. Many modifications and changes in arrangement may be made without departing from the spirit of the present invention.

100 bidirectional DC / DC converter, 102 transformer, W1 primary winding, W2 secondary winding, L1 leakage inductance, 104 first full bridge circuit, 106 second full bridge circuit, 110 controller.

Claims (6)

With a transformer,
A first full bridge circuit connected to a primary winding of the transformer;
A second full bridge circuit connected to a secondary winding of the transformer,
A controller for controlling the first full bridge circuit and the second full bridge circuit;
With
The controller is
Performing a continuous operation of switching the first full-bridge circuit and the second full-bridge circuit with a phase difference δ in a first region where the transmission power is higher than a predetermined threshold value P _TH to change the phase difference δ; Adjust the transmission power by
In a second region where the transmission power is lower than the threshold value P _TH , the phase difference δ is fixed, m is a real number that is a natural number multiple of 0.5, n is a real number, and the first full period is δ + 2πm. Switching the bridge circuit and the second full bridge circuit, performing intermittent operation of stopping the switching of the first full bridge circuit and the second full bridge circuit for 2πn, and adjusting the transmission power by changing n And
The threshold value _{P TH} is, _{P 1} the transmission power which provides the maximum efficiency in continuous operation, when the transmission power that gives the minimum loss in continuous operation was _{P 2,} the _{P 2 ≦ P TH ≦ P 1} × 1.1 An insulated bidirectional DC / DC converter characterized by satisfying. The phase difference [delta] used in the second region, claim wherein the phase difference [delta] ₁ corresponding to the transmission power P _1, to be included between the phase difference [delta] ₂ corresponding to the transmission power P ₂ 1 2. The bidirectional DC / DC converter according to 1. 2. The bidirectional DC / DC converter according to claim 1, wherein the phase difference δ used in the second area is a phase difference δ corresponding to transmission power P _TH . The threshold value P _TH is P ₂ and substantially equal to the phase difference δ to be used in the second region, according to claim 1, characterized in that substantially equal to ₁ and the phase difference δ corresponding to P ₁ 2. The bidirectional DC / DC converter according to 1.   5. The bidirectional DC / DC converter according to claim 1, wherein m = 1. A method of controlling an isolated bidirectional DC / DC converter, comprising:
The bidirectional DC / DC converter includes:
With a transformer,
A first full bridge circuit connected to a primary winding of the transformer;
A second full bridge circuit connected to a secondary winding of the transformer,
A controller for controlling the first full bridge circuit and the second full bridge circuit;
With
The control method includes:
When the transmission power that gives the maximum efficiency in continuous operation is P ₁ and the transmission power that gives the minimum loss in continuous operation is P ₂ , a threshold value P _TH that satisfies P ₂ ≦ P _TH ≦ P ₁ × 1.1 is defined. Steps to
Performing a continuous operation of switching the first full-bridge circuit and the second full-bridge circuit with a phase difference δ in a first region where the transmission power is higher than the threshold value P _TH to change the phase difference δ; Adjusting the transmission power by:
In a second region where the transmission power is lower than the threshold value P _TH , the phase difference δ is fixed, m is a real number that is a natural number multiple of 0.5, n is a real number, and the first full period is δ + 2πm. Switching the bridge circuit and the second full bridge circuit, performing intermittent operation for stopping the switching of the first full bridge circuit and the second full bridge circuit for 2πn, and adjusting the transmission power by changing n Steps to
A control method comprising:

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