JP6156266B2 - Control device - Google Patents

Description

  The present invention relates to a control device for a switched reluctance motor.

  As a control device for a switched reluctance motor, a device that controls a current flowing in a coil of a switched reluctance motor (hereinafter referred to as an SR motor) to a command current using a hysteresis comparator as known in Patent Document 1 below. ing. Specifically, in this apparatus, a zero voltage is used in addition to the positive DC voltage “VDC” and the negative DC voltage “−VDC” as the applied voltage of the SR motor coil. As a result, the number of switching operations is reduced, and switching loss caused by switching of the inverter device is reduced.

Japanese Patent No. 3255167

  Here, in the technique described in Patent Document 1, the applied voltage of the coil is switched from zero voltage to a positive DC voltage, or from zero voltage to negative in order to control the current flowing in the coil to be a command current. It can be switched to the DC voltage. For this reason, the interlinkage magnetic flux waveform of the coil includes a lot of distortion and harmonic components. As a result, there is a concern that the effect of reducing the harmonic iron loss and noise of the SR motor cannot be obtained sufficiently.

  Therefore, a method of controlling the flux linkage waveform by adjusting the voltage applied to the coil instead of controlling the current flowing through the coil can be considered. In this case, since the coil includes a resistance component in addition to the inductance, a voltage drop occurs due to the resistance component, and there is a concern that a desired voltage is not applied to the inductance of the coil. In particular, when the temperature of the coil increases, the resistance component increases and the influence increases. For this reason, we are anxious about the reduction effect of the harmonic iron loss and noise of SR motor by adjustment of the applied voltage of a coil.

  The present invention has been made to solve the above-described problem, and an object of the present invention is to provide a control device that can suitably adjust the flux linkage generated in the coil of the SR motor.

  The present invention provides a voltage output in a switched reluctance motor having a stator having a coil (22) and a rotor that is rotatably supported by the stator and rotates when a voltage is applied to the coil. A control device (30) for controlling the motor by adjusting an output voltage output from the circuit (20) to the coil, the coil having an inductance (L) and a resistance component (R) The rotation angle detection means (36) for detecting the rotation angle of the motor, and the linkage flux generated in the coil by applying a voltage to the inductance in one cycle of the rotation angle has a predetermined waveform. As described above, a command voltage which is a command value of the output voltage of the voltage output circuit is set based on the torque command value for commanding the output torque of the motor and the rotation angle. Command voltage setting means (37), current detection means (32) for detecting a coil current flowing in the coil, and suppression of a change in voltage applied to the inductance caused by current flowing in the resistance component A voltage correction means (38-41) for correcting the command voltage based on a rotation speed of the motor calculated from the rotation angle, the torque command value, and a detected value of the coil current; and the voltage correction means. And a voltage control means (30c) for controlling the output voltage of the voltage output circuit based on the command voltage corrected by the above.

  Since the coil has a resistance component in addition to the inductance, when a current flows through the coil, a voltage drop occurs in the resistance component, and a voltage obtained by subtracting the voltage drop due to the resistance component from the output voltage is applied to the inductance. For this reason, in order to set the interlinkage magnetic flux of the coil to a predetermined value, it is necessary to correct the command voltage in consideration of a voltage drop due to this resistance component. Since the resistance component of the coil changes depending on the coil temperature, it is necessary to consider the change in the resistance component of the coil when the command voltage is corrected.

  The current flowing through the coil is inversely proportional to the magnitude of the resistance component. That is, the influence of the voltage drop due to the resistance component can be corrected based on the magnitude of the current flowing through the coil. In the configuration in which the output voltage output to the coil is adjusted, the inductance of the coil changes depending on the rotation angle of the rotor, so that the current waveform flowing through the coil depends on the rotation state (rotation speed, torque) of the rotor. It changes with. Therefore, by correcting the command voltage based on the rotation state of the rotor and the detected value of the current flowing through the coil, the influence of changes in the resistance component of the coil is suppressed, and the interlinkage magnetic flux of the coil is suitably adjusted. Made possible.

The block diagram of the control system of SR motor concerning 1st Embodiment. The figure which shows the setting method of the command voltage concerning the embodiment. The figure which illustrates the time-dependent change of the electric current which flows into the coil concerning the embodiment. The figure which shows the control method of the command current concerning a prior art. 5 is a flowchart showing a procedure of modulated wave generation processing according to the first embodiment. The flowchart which shows the procedure of the voltage control process concerning the embodiment. The figure which shows the closed circuit formed at the time of the positive voltage application concerning the embodiment. The figure which shows the closed circuit formed at the time of the negative voltage application concerning the embodiment. 6 is a timing chart showing an example of pulse width modulation according to the embodiment. The figure which shows the resistance component which arises in a coil. The functional block diagram of the modulated wave production | generation part concerning 1st Embodiment. The functional block diagram of the correction value calculation part concerning the embodiment. The functional block diagram of the correction value calculation part concerning 2nd Embodiment.

(First embodiment)
Hereinafter, a first embodiment in which a control device according to the present invention is applied to a switched reluctance motor (hereinafter referred to as an SR motor) as an in-vehicle main machine will be described with reference to the drawings.

  As shown in FIG. 1, the high voltage battery 10 is a secondary battery whose terminal voltage is, for example, 100 V or more (288 V). As the high voltage battery 10, for example, a lithium ion secondary battery or a nickel hydride secondary battery can be used.

  A power conversion circuit 20 is connected to the high voltage battery 10 via a smoothing capacitor 12. A motor generator as an in-vehicle main machine is connected to the power conversion circuit 20. The motor generator is an SR motor. The SR motor has a stator having a coil 22 and a rotor (rotor) that is instructed to be rotatable by the stator (stator) and rotates when a voltage is applied to the coil 22. Specifically, in the present embodiment, a three-phase SR motor including a U-phase coil 22u, a V-phase coil 22v, and a W-phase coil 22w is used as the SR motor.

  The power conversion circuit 20 includes a U-phase upper arm switching element Sup and a U-phase lower arm diode Dun connected in series, a U-phase upper arm diode Dup and a U-phase lower arm switching element Sun connected in series, and a V-phase upper arm switching element. Svp and V-phase lower arm diode Dvn in series connection, V-phase upper arm diode Dvp and V-phase lower arm switching element Svn in series, W-phase upper arm switching element Swp and W-phase lower arm diode Dwn in series connection , And a serial connection body of a W-phase upper arm diode Dwp and a W-phase lower arm switching element Swn. Here, in this embodiment, IGBT is used as the U to W phase upper arm switching elements Sup, Svp, Swp and the U to W phase lower arm switching elements Sun, Svn, Swn. In the present embodiment, the U to W phase upper arm diodes Dup, Dvp, and Dwp correspond to “upper arm rectifier elements”, and the U to W phase lower arm diodes Dun, Dvn, and Dwn become “lower arm rectifier elements”. Equivalent to.

  Specifically, the connection point between the U-phase upper arm switching element Sup and the U-phase lower arm diode Dun and the connection point between the U-phase upper arm diode Dup and the U-phase lower arm switching element Sun are connected by the U-phase coil 22u. Yes. The emitter of U-phase upper arm switching element Sup and the cathode of U-phase lower arm diode Dun are connected to each other, and the collector of U-phase upper arm switching element Sup is connected to the positive terminal of high-voltage battery 10. The anode of the U-phase lower arm diode Dun is connected to the negative terminal of the high voltage battery 10. On the other hand, the anode of the U-phase upper arm diode Dup and the collector of the U-phase lower arm switching element Sun are connected to each other, and the cathode of the U-phase upper arm diode Dup is connected to the positive terminal of the high-voltage battery 10. The emitter of the U-phase lower arm switching element Sun is connected to the negative terminal of the high voltage battery 10.

  The connection mode of the switching elements Svp, Svn, Swp, Swn and the diodes Dvp, Dvn, Dwp, Dwn constituting the V phase and the W phase is the same as that of the U phase. For this reason, in this embodiment, the detailed description of the connection aspect about V phase and W phase is abbreviate | omitted.

  The control device 30 includes a central processing unit (CPU) and a memory (not shown), and executes various programs stored in the memory by the central processing unit, whereby the control amount (output torque) of the SR motor is set to its command value (hereinafter, referred to as “control value”). Control to command torque Trq *). The control device 30 detects a U-phase current sensor 32u that detects a current flowing through the U-phase coil 22u, a V-phase current sensor 32v that detects a current flowing through the V-phase coil 22v, and a current that flows through the W-phase coil 22w. The detection value of the W-phase current sensor 32w is input. Further, the control device 30 includes a voltage sensor 34 that detects a voltage between terminals of the smoothing capacitor 12 (an output voltage of the high-voltage battery 10 and an input voltage of the power conversion circuit 20), and a rotation angle (electrical angle) of the rotor of the SR motor. A detection value of a rotation angle sensor 36 (for example, a resolver) for detecting θe) is input. Based on the detection values of these various sensors, the control device 30 controls the upper arm switching element S ¥ p (¥ = u, v, etc.) constituting the power conversion circuit 20 to control the output torque of the SR motor to the command torque Trq *. The operation signals g \ p, g \ n are output to w) and the lower arm switching element S \ n to operate these switching elements S \ p, S \ n.

  Specifically, the control device 30 includes an electrical angular velocity calculation unit 30a, a modulated wave generation unit 30b, and an operation signal generation unit 30c. The electrical angular velocity calculation unit 30a calculates the electrical angular velocity ω as a time differential value of the electrical angle θe detected by the rotation angle sensor 36. Also, the modulation wave generation unit 30b generates the modulation wave α ¥ (¥ = u, v, w) of each phase based on the command torque Trq *, the electrical angle θe, and the electrical angular velocity ω. The operation signal generation unit 30c is configured to operate the upper arm switching element S ¥ p and the lower arm switching element S ¥ n based on the modulation wave α ¥ output from the modulation wave generation unit 30b. \ N is generated. Here, the modulated waves for each of the U to W phases are generated as waveforms that are shifted from each other by 120 ° in electrical angle θe.

  The command torque Trq * is input to the control device 30 from, for example, a control device higher than the control device 30 (for example, a control device that supervises vehicle travel control). In addition, each phase of the SR motor is independent, and each of the processes related to each phase in the control device 30 is basically the same process.

  Subsequently, a command voltage setting method according to the present embodiment will be described with reference to FIG. Here, FIG. 2A shows the transition of the command value (hereinafter referred to as command voltage V ¥ *) of the applied voltage of the coil 22 ¥ (¥ = u, v, w), and FIG. The transition of the interlinkage magnetic flux ψ \ of 22 \ is shown. FIG. 2 shows the transition when the electrical angular velocity ω is constant.

  As shown in the figure, in this embodiment, the command voltage V ¥ * is gradually changed in both the rising period (time t1 to t2, t3 to t4) and the falling period (time t2 to t3) of the command voltage V ¥ *. In addition, the command voltage V ¥ * is set so that the waveform drawn by the command voltage V ¥ * is continuous in one cycle (time t1 to t5) of the electrical angle θe. Here, the rising period is a period during which the command voltage V ¥ * increases (in other words, a period during which the change rate of the command voltage V ¥ * is higher than 0), and the falling period is the command voltage. This is a period during which V ¥ * falls (in other words, a period during which the change rate of the command voltage V ¥ * is lower than 0).

  In particular, in the present embodiment, in the energization instruction period (time t1 to t4) that is included in one cycle of the electrical angle θe and instructs energization of the coil 22 ¥, the first command voltage V ¥ * is drawn. After the sine waveform (the waveform drawn by the command voltage V ¥ * from time t1 to tA, which corresponds to the “first waveform”), the first axis L1 defining “0” of the command voltage V ¥ * and A second sine waveform that is point-symmetric with the first waveform with respect to the intersection γ of the second axis L2 that is orthogonal to the first axis L1 and passes through the center of the energization instruction period (command by time tA to t4) The command voltage V ¥ * is set so that the waveform is drawn by the voltage V ¥ * and corresponds to the “second waveform”. Here, the area surrounded by the first sine waveform and the first axis L1 is equal to the area surrounded by the second sine waveform and the first axis L1.

  The above setting of the command voltage V ¥ * is made for the purpose of reducing the harmonic iron loss, vibration and noise of the SR motor. In other words, by reducing the change in the interlinkage magnetic flux ψ \ of the coil 22 \, the harmonic iron loss, vibration and noise of the SR motor can be reduced. Here, from the relationship that the time integral value of the applied voltage of the coil 22 \ not affected by the time change of the inductance L becomes the linkage flux ψ \ of the coil 22 \, the rising period and the rising of the applied voltage of the coil 22 \ By gradually changing the voltage applied to the coil 22 ¥ during the falling period, the change of the linkage flux ψ ¥ can be moderated.

  Specifically, at time t1 to t2, the command voltage V ¥ * is gradually increased from 0 to the first voltage V1 that is the upper limit value, so that the applied voltage V ¥ r of the coil 22 ¥ is also increased from 0 to the upper limit value. Rises gradually. As a result, the interlinkage magnetic flux ψ \, which is an integral value of the applied voltage V ¥ r, is also gradually increased from 0, which is the timing value when the command voltage V ¥ * is 0, to the first magnetic flux amount ψ1, which is a positive value. To rise.

  At subsequent times t2 to tA, the applied voltage V ¥ r gradually decreases from the first voltage V1 to 0 by gradually decreasing the command voltage V ¥ * from the first voltage V1 to 0. As a result, the interlinkage magnetic flux ψ \ gradually rises to the second magnetic flux amount ψ2 that is the upper limit value while lowering the rate of increase (rate of increase).

  At subsequent times tA to t3, the applied voltage V ¥ r gradually decreases from 0 to the second voltage V2 by gradually decreasing the command voltage V ¥ * from 0 to the second voltage V2. Here, the second voltage V2 is a lower limit value and a negative value of the command voltage V ¥ *. In particular, in the present embodiment, the absolute value of the second voltage V2 and the absolute value of the first voltage V1 are set to the same value. As a result of the setting of the command voltage V ¥ *, the interlinkage magnetic flux ψ ¥ gradually decreases from the second magnetic flux amount ψ2 to the first magnetic flux amount ψ1.

  At subsequent times t3 to t4, the applied voltage V ¥ r is gradually increased from the second voltage V2 to 0 by gradually increasing the command voltage V ¥ * from the second voltage V2 to 0. As a result, the interlinkage magnetic flux ψ \ gradually decreases to 0 while lowering its descending speed (decreasing rate).

  In particular, by setting the command voltage V ¥ * so that the waveform drawn by the command voltage V ¥ * is a sine waveform (ideal sine wave), as shown in FIG. The waveform drawn by the interlinkage magnetic flux ψ \ in the period can be made sinusoidal, and the effect of reducing harmonic iron loss or the like can be increased.

  It should be noted that the current flowing through the coil 22 ¥ changes smoothly in a range between 0 and the upper limit value in the change of the command voltage V ¥ * in the electrical angle θe1 period. However, the current flows in a manner corresponding to the command voltage V ¥ *. This situation is shown in FIG. 3A shows the transition of the inductance L of the coil 22 ¥, FIG. 3B shows the transition of the command voltage V ¥ *, and FIG. 3C flows to the coil 22 ¥. The transition of the coil current i ¥ is shown.

  As shown in the figure, the change with time in one cycle of the command voltage V ¥ * may be different from the change with time in the one cycle of the coil current i ¥. This is because, for example, the inductance L is different or the electrical angular velocity ω is different according to the electrical angle θe. In FIG. 3, when the same applied voltage V ¥ r corresponding to the same command voltage V ¥ * is applied to the coil 22 ¥, the coil current i ¥ at the electrical angles θ1 to θ2 is represented by the electrical angles θ3 to θ4. The phenomenon of becoming larger than the coil current i ¥ in FIG. This phenomenon occurs because the inductance L at the electrical angles θ1 to θ2 is smaller than the inductance L at the electrical angles θ3 to θ4.

  Such a technical idea of gradually changing the applied voltage of the coil 22 \ so as to moderate the change of the linkage flux ψ \ is different from the technique described in Patent Document 1 (hereinafter, conventional technique). Here, FIG. 4 shows the transition of the interlinkage magnetic flux ψ \ and the like according to the prior art. Specifically, FIG. 4A shows the command value of the current flowing through the coil 22 ¥ (hereinafter referred to as command current i ¥ *) and the current detected by the $ phase current sensor 32 ¥ (hereinafter referred to as detected current i ¥ r). FIG. 4B shows the transition of the applied voltage V ¥ r of the coil 22 ¥, and FIG. 4C shows the transition of the flux linkage ψ ¥ of the coil 22 ¥.

  As shown in the figure, in the prior art, for the purpose of reducing torque ripple of the SR motor, a rectangular wave current is set as the command current i ¥ *, and the detection current i ¥ r is controlled to the command current i ¥ *. ing. In such a control method, as shown in FIG. 4 (b), the applied voltage V \ r of the coil 22 \ is the result. Therefore, the times t1 to t2 when both the upper arm switching element S ¥ p and the lower arm switching element S ¥ n are turned on, and both the upper arm switching element S ¥ p and the lower arm switching element S ¥ n are turned off. At the time t3 to t4 when the operation is performed, the applied voltage V ¥ r becomes discontinuous and changes abruptly. As a result, as shown in FIG. 4C, the interlinkage magnetic flux ψ \ of the coil 22 \ also changes steeply.

  Specifically, from time t1 to t2, the applied voltage V ¥ r steeply rises to 0 after steeply rising so that the detected current i ¥ r becomes the command current i ¥ * by setting the command current i ¥ *. To fall. Along with this, the detection current i \ r rises to a predetermined current value in a short time. The interlinkage magnetic flux ψ \ is an integral value of the applied voltage V \ r, and thus rises to a predetermined magnetic flux in a short time.

  At subsequent times t2 to t3, since the detected current i ¥ r becomes substantially equal to the command current i ¥ *, the applied voltage V ¥ r changes at a value that keeps the detected current i ¥ r constant. In this case, the applied voltage V ¥ r gradually increases from 0 to a predetermined positive voltage. Corresponding to the increase in the applied voltage V ¥ r, the flux linkage ψ ¥ also gradually increases.

  At subsequent times t3 to t4, the command current i ¥ * is set to zero. As a result, the applied voltage V ¥ r steeply rises to zero and then steeply rises to 0 so that the detection current i ¥ r becomes the command current i ¥ *. As a result, the detection current i \ r drops to 0 in a short time. The interlinkage magnetic flux ψ \ takes an upper limit at the timing when the applied voltage V \ r becomes 0, and then falls to 0 in a short time.

  There is a concern that the harmonic iron loss, vibration, and noise of the SR motor are not sufficiently reduced due to the change in the interlinkage magnetic flux ψ \ described above.

  In order to solve such a problem, in this embodiment, as described above, the command voltage V ¥ * is set while being gradually changed without setting the command current i ¥ *. Here, in the present embodiment, the upper arm switching element S ¥ p and the lower arm switching element S ¥ n are turned on and off by pulse width modulation. For this reason, the modulated wave α ¥ normalized by dividing the command voltage V ¥ * by the DC voltage Vdc output from the high voltage battery 10 is set as the command voltage V ¥ *.

  FIG. 5 shows a procedure of modulated wave generation processing according to the present embodiment. This process is repeatedly executed, for example, at a predetermined cycle by the modulated wave generation unit 30b included in the control device 30. In the present embodiment, a series of processing shown in FIG. 5 constitutes “command voltage setting means”.

  In this series of processing, first, in step S10, the command torque Trq *, the electrical angle θe, and the electrical angular velocity ω are acquired.

  In the subsequent step S12, based on the acquired command torque Trq *, electrical angle θe, and electrical angular velocity ω, the command modulation rate α0 and the ON phase θon that is the electrical angle θe that instructs the start of energization of the U-phase coil 22u (the previous phase). 2 is set, and an energization instruction width θw (time t1 to t4 in FIG. 2), which is the width of the electrical angle θe that indicates the energization continuation period from the ON phase θon. Here, the command modulation rate α0 is a value obtained by dividing the amplitude of the command voltage V ¥ * set as a sine waveform by the DC voltage Vdc output from the high-voltage battery 10. The period from when the electrical angle θe becomes the ON phase θon to when the energization instruction width θw elapses is the energization instruction period. Note that the command modulation rate α0, the ON phase θon, and the energization instruction width θw are related to the command torque Trq * and the electrical angular velocity ω, and are a map or formula that defines the command modulation rate α0, the ON phase θon, and the energization instruction width θw. Can be set using.

  In a succeeding step S14, it is determined whether or not the current electrical angle θe is the energization instruction period. Specifically, if a value obtained by subtracting the ON phase θon from the current electrical angle θe is defined as a numerator and a value having the energization instruction width θw as a denominator is defined as a determination parameter, the determination parameter is “0” or more and “ It is determined whether it is 1 ”or less.

  If a negative determination is made in step S14, it is determined that the current electrical angle θe is not the energization instruction period, and the process proceeds to step S16. In step S16, the modulation wave α ¥ is set to “0”. On the other hand, if an affirmative determination is made in step S14, it is determined that the current electrical angle θe is the energization instruction period, and the process proceeds to step S18. In step S18, a modulation wave α ¥ is generated by multiplying the command modulation factor α0 by a sine function having the determination parameter and the multiplication value of “360 °” as an independent variable. In addition, when the process of step S16, S18 is completed, this series of processes is once complete | finished.

  Incidentally, the value obtained by multiplying the modulation wave α ¥ generated by the processing shown in FIG. 5 by the DC voltage Vdc output from the high voltage battery 10 is the command voltage V ¥ * shown in FIG. It becomes.

  Next, FIG. 6 shows a procedure of voltage control processing according to the present embodiment. This process is repeatedly executed, for example, at a predetermined cycle by the operation signal generation unit 30c included in the control device 30. In the present embodiment, the series of processing shown in FIG. 6 constitutes “voltage control means”.

  In this series of processing, first, in step S20, it is determined whether or not the modulated wave α ¥ output from the modulated wave generation unit 30b is equal to or greater than the carrier signal Cs. In the present embodiment, a triangular wave signal is used as the carrier signal Cs. In the present embodiment, the carrier signal Cs has a minimum value set to “−1” and a maximum value set to “1”.

  When an affirmative determination is made in step S20, the process proceeds to step S22, and both the upper arm switching element S ¥ p and the lower arm switching element S ¥ n are turned on. As a result, as illustrated in FIG. 7 for the U phase, a current flows through the closed circuit including the high voltage battery 10, the U phase upper arm switching element Sup, the U phase coil 22u, and the U phase lower arm switching element Sun. A positive voltage “Vdc” is applied to the coil 22u.

  On the other hand, if a negative determination is made in step S20, the process proceeds to step S24, and both the upper arm switching element S ¥ p and the lower arm switching element S ¥ n are turned off. Thereby, as illustrated in FIG. 8 for the U phase, a current flows through a closed circuit including the high voltage battery 10, the U phase lower arm diode Dun, the U phase coil 22u, and the U phase upper arm diode Dup, and the U phase coil 22u A negative voltage “−Vdc” is applied to In addition, when the process of step S22 and S24 is completed, this series of processes is once complete | finished.

  According to the above-described operation processing of the upper arm switching element S ¥ p and the lower arm switching element S ¥ n, as shown in FIG. 9, the switching is performed by pulse width modulation based on the magnitude comparison of the modulated wave α ¥ and the carrier signal Cs. The elements S \ p and S \ n can be turned on / off. As a result, the average applied voltage of the coil 22 ¥ in each cycle of the carrier signal Cs can be controlled to the command voltage V ¥ *. 9A shows the transition of the modulation wave α ¥ and the carrier signal Cs, and FIG. 9B shows the transition of the applied voltage V ¥ r and the command voltage V ¥ * of the coil 22 ¥.

  Here, in order to change the linkage flux ψ ¥ to a desired change, the command voltage V ¥ * and the applied voltage V ¥ r (that is, applied to the inductance L) contributing to the linkage flux generated in the coil 22 are applied. Voltage) must match. However, as shown in FIG. 10, since the coil 22 has a resistance component R in addition to the inductance L, when a current flows through the coil 22, a voltage drop VR is generated by the resistance component R (VR = i ¥ · R). Due to this voltage drop VR, the applied voltage V ¥ r becomes smaller than the command voltage V ¥ *.

  If the resistance component R is constant regardless of temperature or the like, the voltage drop VR can be calculated by the product of the constant value and the detected value i ¥ r of the coil current flowing through the coil 22, The command voltage V ¥ * can be corrected based on the calculated value. However, when a current flows through the coil 22, the coil 22 generates heat, and the resistance component R increases with the heat generation. That is, the value of the resistance component R varies depending on the operation of the three-phase SR motor, and the influence of the voltage drop VR due to the resistance component R also changes.

  Therefore, in the present embodiment, the value of the command voltage V ¥ * is corrected so as to suppress the influence of the resistance component R accompanying the temperature change. Note that the actual correction target is the value of the modulated wave α ¥ normalized by dividing the command voltage V ¥ * by the DC voltage Vdc.

  Specifically, the value of the resistance component R at a predetermined temperature (for example, 30 degrees) is set as the reference value R *, and the coil current estimated to flow through the coil 22 when the value of the resistance component R is the reference value R *. Is compared with the detected current i \ r that is the detected value of the coil current that is actually flowing through the coil 22. Then, a change in the resistance component R is calculated based on a comparison between the detected current i ¥ r and the estimated current i ¥ ^, and a voltage drop change ΔVR is calculated based on the calculated value. Then, by correcting the modulation wave α ¥ based on the voltage drop change amount ΔVR, the command voltage V ¥ * can be corrected so as to suppress the influence of the voltage drop VR change.

  The coil current i ¥ flows when a voltage is applied to the coil 22 during the energization instruction period (θon to θoff) included in one cycle of the electrical angle θe. Since the energization instruction period is set according to the command torque Trq * and the electrical angular speed ω, the energization instruction period changes when at least one of the command torque Trq * and the electrical angular speed ω changes. Therefore, the detected current i \ r and the estimated current i \ ^ are calculated by calculating the deviation of each feature amount (for example, effective value) in one cycle of the electrical angle θe for the detected current i \ r and the estimated current i \ ^. It is set as the structure which compares with ^.

  FIG. 11 shows a functional block diagram of the modulated wave generator 30b. The modulation wave generation unit 30b includes a basic modulation wave generation unit 37, a correction value calculation unit 38, a first feature value calculation unit 39, a second feature value calculation unit 40, and an addition unit 41. The basic modulation wave generation unit 37 performs the control shown in FIG. 5 to calculate and output a modulation wave α ¥ so that the interlinkage magnetic flux of the coil 22 ¥ has a predetermined waveform. The correction value calculation unit 38, the first feature value calculation unit 39, the second feature value calculation unit 40, and the addition unit 41 constitute voltage correction means.

  Based on the command torque Trq * and the electrical angular velocity ω, the first feature amount calculation unit 39 calculates Ie *, which is an effective value of the estimated current i ¥ ^, as the first feature amount. As shown in FIG. 3C, the coil current i ¥ changes in waveform according to the applied voltage V ¥ r, the ON phase θon, and the energization instruction width θw. The applied voltage V ¥ r, the ON phase θon, and the energization instruction width θw can be calculated using a map associated with the command torque Trq * and the electrical angular velocity ω, and the estimated current i ¥ ^ and its effective value Ie * are the same. And a map associated with the command torque Trq * and the electrical angular velocity ω.

  Based on the detected current i \ r in one cycle of the electrical angle θe, the second feature quantity calculator 40 calculates Ie, which is the effective value of the detected current i \ r, as the second feature quantity.

  The correction value calculation unit 38 calculates the correction value αad of the modulated wave α ¥ based on the first feature value Ie *, the second feature value Ie, and the detection current i ¥ r. Adder 41 corrects by adding correction value αad to modulated wave α ¥. Then, the corrected modulated wave α ¥ is output to the operation signal generator 30c.

  FIG. 12 is a functional block diagram showing details of the correction value calculation unit 38. The deviation calculating unit 42 calculates a deviation ΔIe between the first feature value Ie * (effective value of the estimated current i ¥ ^) and the second feature value Ie (effective value of the detected current i ¥ r). The PI calculation unit 43 calculates the change amount ΔR of the resistance component by performing a proportional-integral calculation on the deviation ΔIe calculated by the deviation calculation unit 42. The multiplication unit 44 calculates a change amount ΔVR of a voltage drop accompanying a change in the resistance component R as a product of the detection current i ¥ r and the change amount ΔR of the resistance component.

  The multiplication unit 45 calculates a reference value VR * of a voltage drop due to the resistance component R as a product of the detection current i ¥ r and the reference value R * of the resistance component R. The adder 46 calculates the correction value Vad as the sum of the voltage drop variation ΔVR calculated by the multiplier 44 and the voltage drop reference value VR * calculated by the multiplier 45. The division unit 47 divides the voltage correction value Vad by the DC voltage Vdc to normalize the modulation wave correction value αad, and outputs the result to the addition unit 41.

  Hereinafter, effects in the present embodiment will be described.

  Since the coil 22 has a resistance component R in addition to the inductance L, a voltage drop VR occurs when a current flows through the resistance component R, and a voltage obtained by subtracting the voltage drop VR due to the resistance component R from the output voltage is the inductance L of the coil 22. Will be applied. For this reason, in order to set the interlinkage magnetic flux of the coil 22 to a predetermined value, it is necessary to correct the command voltage V ¥ * in consideration of the voltage drop VR due to the resistance component R. When the command voltage V ¥ * is corrected, the resistance component R of the coil 22 changes depending on the temperature of the coil 22, and therefore, it is necessary to consider the change of the resistance component R of the coil 22.

  The current flowing through the coil 22 is inversely proportional to the magnitude of the resistance component R. That is, the influence of the voltage drop VR due to the resistance component R can be corrected based on the magnitude of the current flowing through the coil 22. In the configuration in which the output voltage output to the coil 22 is adjusted, since the inductance L of the coil 22 varies depending on the electrical angle θe of the rotor, the current waveform flowing through the coil 22 changes to the motor rotation state (rotation speed, torque). It changes according to the course. Therefore, by correcting the command voltage V ¥ * based on the rotation state of the rotor and the detected value of the current flowing in the coil 22, the influence due to the change in the resistance component R of the coil 22 is suppressed, and the interlinkage magnetic flux generated in the coil 22. Can be suitably adjusted.

  The current waveform flowing through the coil 22 can be estimated based on the rotation state of the motor (the command torque Trq * and the electrical angular velocity ω of the rotor). When the resistance component R varies, the amplitude of the current waveform that actually flows through the coil 22 differs from the amplitude of the current waveform that is estimated based on the rotation state of the motor. That is, based on a comparison between the estimated current i ¥ ^ that is an estimated value estimated based on the rotation state of the motor and the detected current i ¥ r that is a detected value of the current that actually flows through the coil 22, the resistance component R It is possible to calculate the correction value Vad of the command voltage V ¥ * in consideration of the change of.

  In the present embodiment, in the comparison of the current waveforms of the estimated current i \ ^ and the detected current i \ r, the feature amounts in one cycle of the electrical angle θe are calculated, and the feature amounts are compared. With this configuration, errors caused by the phase difference between the estimated current i \ ^ and the detected current i \ r can be suppressed.

  As the first feature value Ie *, an effective value of the estimated current i ¥ ^ in one cycle of the electrical angle θe is calculated, and as the second feature value Ie, the effective value of the detected current i ¥ r in one cycle of the electrical angle θe. It was set as the structure which calculates a value. The command voltage V ¥ * is corrected based on the deviation ΔIe between the effective values. According to this configuration, even if instantaneous noise occurs in the detection current i \ r, the influence of noise can be suppressed by using the effective value over one period.

  By changing and setting the command voltage V ¥ * gradually in at least one of the rising period and the falling period of the command voltage V ¥ *, the change of the linkage flux generated in the coil 22 is gradually changed. By gradually changing the interlinkage magnetic flux, the high frequency component of the interlinkage magnetic flux can be suppressed, and the effect of reducing the harmonic iron loss and noise of the SR motor can be suppressed. In the present embodiment, the flux linkage is made sinusoidal by setting the command voltage to be sinusoidal.

(Second Embodiment)
In the above embodiment, the reference value R * of the resistance component R of the coil 22 is set, and ΔR that is the amount of change with respect to the reference value R * is calculated. In the second embodiment, this is changed and the current resistance component R is directly calculated.

  FIG. 13 is a functional block diagram of the correction value calculation unit 38a in the present embodiment. The deviation calculating unit 50 calculates a deviation ΔIe between the first feature quantity Ie * and the second feature quantity Ie. The PI control unit 51 calculates the resistance component R by performing proportional-integral control on the deviation ΔIe. The integrating unit 52 calculates the command voltage correction value Vad by integrating the resistance component R calculated by the PI control unit 51 and the detected current i ¥ r. The division unit 53 divides the voltage correction value Vad by the DC voltage Vdc to normalize the modulation wave correction value αad, and outputs the result to the addition unit 41.

(Other embodiments)
As the first feature value and the second feature value, a peak value or an average value of the coil current i ¥ in one cycle of the electrical angle θe may be used.

  -It is good also as a structure which compares the estimated value of a coil current, and the detected value of a coil current, without calculating a 1st feature-value and a 2nd feature-value. That is, at a predetermined timing, the estimated value of the coil current and the detected value of the coil current may be compared, and the command voltage V ¥ * may be corrected based on the deviation.

  In the above embodiment, the modulation wave α ¥ standardized by the DC voltage Vdc is corrected. However, it may be changed to directly correct the command voltage V ¥ *.

  In the above embodiment, the sine wave command voltage V ¥ * is output. However, this may be changed. In this case, the command voltage V ¥ * may be gradually changed and set in at least one of the rising period and the falling period of the command voltage V ¥ *. For example, instead of the sinusoidal command voltage V ¥ *, a triangular wave or trapezoidal command voltage may be output.

  DESCRIPTION OF SYMBOLS 20 ... Power conversion circuit, 22 ... Coil, L ... Inductance, R ... Resistance component, 30 ... Control apparatus, 30c ... Operation signal production | generation part (voltage control means), 32 ... ¥ phase current sensor (current detection means), 36 ... Rotation angle sensor (rotation angle detection means), 37 ... basic modulation wave generation section (command voltage setting means), 38 ... correction value calculation section (voltage correction means), 39 ... first feature quantity calculation section (voltage correction means), 40: second feature amount calculation unit (voltage correction unit), 41 ... addition unit (voltage correction unit).

Claims (5)

A voltage output circuit (20) in a switched reluctance motor having a stator having a coil (22) and a rotor rotatably supported by the stator and rotating when a voltage is applied to the coil. A control device (30) for controlling the motor by adjusting an output voltage output from the coil to the coil,
The coil has an inductance (L) and a resistance component (R),
Rotation angle detection means (36) for detecting the rotation angle of the motor;
A torque command value for instructing an output torque of the motor so that a linkage magnetic flux generated in the coil by applying a voltage to the inductance in one cycle of the rotation angle has a predetermined waveform; and Command voltage setting means (37) for setting a command voltage which is a command value of the output voltage of the voltage output circuit based on the rotation angle;
Current detection means (32) for detecting a coil current flowing in the coil;
In order to suppress a change in voltage applied to the inductance caused by a current flowing through the resistance component, the rotational speed of the motor calculated from the rotation angle, the torque command value, and the detected value of the coil current Voltage correction means (38-41) for correcting the command voltage based on
Voltage control means (30c) for controlling the output voltage of the voltage output circuit based on the command voltage corrected by the voltage correction means;
A control device comprising:   The voltage correction means calculates a correction value of the command voltage based on a comparison between the estimated value of the coil current determined according to the torque command value and the rotation speed and a detected value of the coil current, The control device according to claim 1, wherein the command voltage is corrected by adding a correction value to the command voltage. The voltage correction means includes
First feature amount calculating means (39) for calculating a first feature amount representing an estimated value of the coil current in one cycle of the rotation angle based on the torque command value and the rotation speed;
Second feature amount calculating means (40) for calculating a second feature amount representing a detected value of the coil current in one cycle of the rotation angle based on the detected value of the coil current;
With
The control device according to claim 2, wherein a correction value of the command voltage is calculated based on a deviation between the first feature value and the second feature value. The first feature amount calculating means calculates an effective value of an estimated value of the coil current in one cycle of the rotation angle as the first feature amount,
The control device according to claim 3, wherein the second feature amount calculation unit calculates an effective value of the detected value of the coil current in one cycle of the rotation angle as the second feature amount.   5. The command voltage setting unit according to claim 1, wherein the command voltage is set by gradually changing the command voltage in at least one of a rising period and a falling period of the command voltage. 6. Control device.

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