JP6110650B2 - Wraparound canceller and relay device - Google Patents

Description

  The present invention relates to a relay station or a relay apparatus in digital broadcasting or digital transmission using an OFDM (Orthogonal Frequency Division Multiplexing) system, and in particular, between a transmitting and receiving antenna of a broadcast wave relay station in SFN (Single Frequency Network). The present invention relates to a sneak canceller for removing a sneak current of radio waves in the network.

  The sneak canceller cancels the sneak wave component generated by the coupling between the transmitting and receiving antennas included in the received signal in the SFN broadcast wave relay station that retransmits at the same frequency as the frequency of the received signal, and retransmits only the upper station signal. It is a device for.

  The sneak occurs when a part of the radio wave radiated from the transmitting antenna passes through the sneak path and is received by the receiving antenna that receives the higher-order local wave, and is the same as the sneak path within the sneak canceller. If a circuit with transmission characteristics is realized, a replica signal of a sneak wave can be generated.

  The sneak canceller is a device that cancels sneak by subtracting a sneak wave replica signal generated inside the device from a received signal, and extracts only the upper station signal (see, for example, Patent Documents 1 to 6).

  On the other hand, as digital transmission using the OFDM system, multimedia broadcasting using the VHF band has been operated and studied. Comparing multimedia broadcasting using the VHF band and terrestrial digital broadcasting using the UHF band, although transmission parameters and the like are generally the same, multimedia broadcasting has a characteristic that the transmission bandwidth is wider than that of terrestrial digital broadcasting. is there. In particular, this tendency becomes remarkable at the time of concatenated transmission in which a plurality of segments are transmitted from the same point without a guard band.

JP-A-11-355160 JP 2000-341238 A JP 2000-349734 A JP 2001-94528 A JP 2000-295195 A Japanese Patent No. 4132578

  In such multimedia broadcasting, the transmission device adds phase correction (phase rotation correction) to each segment signal at the time of concatenated transmission in which a plurality of segment signals are transmitted without a guard band. This is because the transmitting device collectively performs IFFT (Inverse Fast Fourier Transform) on a plurality of segment signals without applying phase correction, transmits an OFDM signal, and the receiving device transmits a desired segment signal. If you select, perform FFT (Fast Fourier Transform), and perform demodulation, the center frequency of multiple segment signals differs from the center frequency of the selected segment signal, so phase rotation is performed on the received signal to be demodulated. This is because discontinuity due to the above occurs and correct demodulation cannot be performed.

  For this reason, assuming a case where a conventional wraparound canceller is applied to multimedia broadcasting, the wraparound canceller adds phase correction to each segment signal at the time of concatenated transmission in which the transmitter transmits a plurality of segment signals without a guard band. It is necessary to consider that.

  Therefore, the present invention has been made to solve the above-described problems, and its purpose is to eliminate wraparound due to coupling between the transmitting and receiving antennas when receiving a broadcast wave in which a plurality of unit transmission waves are connected, An object of the present invention is to provide a sneak canceller that equalizes distortion caused by multipath and a relay device that relays a higher-order station wave favorably and stably using the same.

To achieve the above object, the wraparound canceller according to claim 1 receives an OFDM signal of a broadcast wave that is concatenated and transmitted by performing phase rotation correction on a segment basis by a transmitting side , and a wraparound canceller included in the received OFDM signal. FBF (Feed Back Filter) that generates a replica signal, FFF (Feed Forward Filter) that equalizes distortion due to multipath of a signal after canceling a wraparound replica signal from the received OFDM signal, and the FBF and FFF An effective symbol period in which the filter coefficient control unit extracts an OFDM signal having an effective symbol period from the OFDM signal equalized by the FFF The O extracted by the extraction unit and the effective symbol period extraction unit FFT (Fast Fourier Transform) performed on the DM signal and converted into a carrier symbol, a channel response calculator for calculating a channel response of the carrier symbol converted by the FFT unit, and a channel response calculator An equalization error calculation unit that calculates an equalization error based on a channel response; an FFF coefficient calculation unit that calculates a filter coefficient used in the FFF from the equalization error calculated by the equalization error calculation unit; Based on the channel response calculated by the channel response calculation unit, a cancellation residual calculation unit for calculating a cancellation residual, and a filter coefficient used in the FBF from the cancellation residual calculated by the cancellation residual calculation unit and a FBF coefficient calculation unit for calculating a, the channel response calculation section, amplitude and phase are known Pas A pilot signal generation unit that generates a lot signal, a phase compensation unit that performs the same phase rotation correction as the transmission side on the pilot signal generated by the pilot signal generation unit, and compensates the phase of the pilot signal; and the FFT A division unit that divides a predetermined pilot signal of the carrier symbols converted by the unit by a pilot signal whose phase is compensated by the phase compensation unit, and the division result by the division unit is interpolated in the symbol direction and the carrier direction. An interpolation unit that outputs the division result after interpolation as the channel response.

The wraparound canceller according to claim 2 is the wraparound canceller according to claim 1 , wherein the channel response calculation unit includes a compensated pilot signal generation unit instead of the pilot signal generation unit and the phase compensation unit, The completed pilot signal generation unit generates a pilot signal having a known amplitude, the phase of which is the same as that of the transmission side and having the known phase, and the division unit converts the carrier symbol converted by the FFT unit. The predetermined pilot signal is divided by the pilot signal generated by the compensated pilot signal generation unit.

A wraparound canceller according to claim 3 is the wraparound canceller according to claim 1, wherein a new channel response calculation unit in place of the channel response calculation unit generates a pilot signal having a known amplitude and phase. A first phase compensation unit that performs the same phase rotation correction on the pilot signal generated by the pilot signal generation unit as the transmission side and compensates for the phase of the pilot signal, and the carrier converted by the FFT unit. A first division unit that divides a predetermined pilot signal of symbols by a pilot signal whose phase is compensated by the first phase compensation unit, and a division result obtained by the first division unit as a symbol direction and a carrier direction And an interpolation unit that outputs the interpolated channel response and a carrier converted by the FFT unit. The symbol is divided by the channel response interpolated by the interpolation unit, and a second division unit for channel equalization, and the Euclidean distance in the signal space from the carrier symbol channel equalized by the second division unit is the largest. A determination unit that determines a small known transmission symbol as an estimation value of a transmission symbol, and an estimation value of the transmission symbol that is subjected to the same phase rotation correction as that of the transmission side on the estimation value of the transmission symbol determined by the determination unit wherein the second phase compensating portion for compensating the phase, the converted carrier symbols by the FFT unit, divided by the estimate of the transmitted symbol whose phase is compensated by the second phase compensating section, the division result of And a third division unit that outputs a channel response.

The wraparound canceller according to claim 4 is the wraparound canceller according to claim 3 , wherein the new channel response calculation unit includes a new determination unit instead of the determination unit and the second phase compensation unit, The new determination unit performs the same phase rotation correction as that of the transmission side on the known transmission symbol having the smallest Euclidean distance in the signal space from the carrier symbol channel-equalized by the second division unit. The estimated value is determined as an estimated value of the transmission symbol, and the third division unit divides the carrier symbol converted by the FFT unit by the estimated value of the transmission symbol determined by the new determination unit, The division result is output as the channel response.

The wraparound canceller according to claim 5 is the wraparound canceller according to claim 4 , wherein the new channel response calculation unit is a compensated pilot signal generation unit instead of the pilot signal generation unit and the first phase compensation unit. And the compensated pilot signal generation unit generates a pilot signal having a known amplitude and having a known phase subjected to the same phase rotation correction as that of the transmission side, and the first division unit includes A predetermined pilot signal of the carrier symbols converted by the FFT unit is divided by the pilot signal generated by the compensated pilot signal generation unit.

Furthermore, a relay apparatus according to a sixth aspect uses the wraparound canceller according to any one of the first to fifth aspects.

  As described above, according to the present invention, when receiving a broadcast wave in which a plurality of unit transmission waves are connected, a wraparound canceller that eliminates wraparound due to coupling between transmitting and receiving antennas and equalizes distortion due to multipath, and By using this, it is possible to provide a relay device that relays the upper station wave favorably and stably.

It is a block diagram which shows the structure of the 1st (Example 1) wraparound canceller by embodiment of this invention. It is a block diagram which shows the structure of the 2nd (Example 2) wraparound canceller by embodiment of this invention. FIG. 3 is a block diagram illustrating a first configuration of a channel estimation unit in the first embodiment. It is a block diagram which shows the 2nd structure of the channel estimation part in Example 1. FIG. It is a block diagram which shows the 1st structure of the channel estimation part in Example 2. FIG. It is a block diagram which shows the 2nd structure of the channel estimation part in Example 2. FIG. It is a block diagram which shows the 3rd structure of the channel estimation part in Example 2. FIG. It is a block diagram which shows the 4th structure of the channel estimation part in Example 2. FIG. It is a figure which shows arrangement | positioning of SP. It is a block diagram which shows the 1st structure of an LPF part. It is a block diagram which shows the 2nd structure of an LPF part. It is a block diagram which shows the structure of the relay apparatus using the wraparound canceller of Example 1,2.

  Hereinafter, embodiments for carrying out the present invention will be described in detail with reference to the drawings. The wraparound canceller according to the embodiment of the present invention receives a broadcast wave in which a plurality of unit transmission waves are connected by phase correction on a segment basis, removes a wraparound replica signal from the signal after orthogonal demodulation, A device that equalizes distortion caused by multipath by an FFF (Feed Forward Filter), a filter coefficient used by an FBF (Feed Back Filter) that generates a wraparound replica signal, and a filter that generates a filter coefficient used by the FFF The coefficient control unit is characterized in that the channel response is estimated in consideration of the phase correction component applied in segment units by the transmission side. The filter coefficient control unit of the wraparound canceller according to the embodiment of the present invention generates a channel response from which the multipath distortion component is removed from the channel response estimated in consideration of the phase correction component, and generates an FBF filter coefficient. Further, the filter coefficient control unit extracts a low-frequency component from the channel response estimated in consideration of the phase correction component, generates a channel response including only the propagation path characteristic, and generates an FFF filter coefficient.

[Wraparound canceller of Example 1]
First, the wraparound canceller according to the first embodiment will be described. FIG. 1 is a block diagram illustrating a configuration of the wraparound canceller according to the first embodiment. The wraparound canceller 1 includes a frequency converter 11, an A / D (analog / digital) converter 12, an orthogonal demodulator 13, a subtractor 14, an FFF 15, an FBF 16, an orthogonal modulator 17, and a D / A (digital / analog) conversion. A unit 18, a frequency conversion unit 19, and a filter coefficient control unit 20 are provided.

  When the wraparound canceller 1 receives a broadcast wave in which a phase correction is performed on a transmission side and a plurality of unit transmission waves are connected, the frequency conversion unit 11 converts the received signal into an IF band signal and outputs it To do. The IF signal output from the frequency converter 11 is input to the A / D converter 12. The A / D conversion unit 12 converts the analog IF signal input from the frequency conversion unit 11 into a digital signal using a sampling clock supplied from a synchronous reproduction unit (not shown), and outputs the digital signal as a digital IF signal. The digital IF signal output from the A / D converter 12 is input to the quadrature demodulator 13. The quadrature demodulator 13 performs quadrature demodulation on the digital IF signal input from the A / D converter 12 and outputs it as an equivalent baseband signal. The equivalent baseband signal output from the orthogonal demodulation unit 13 is input to the subtraction unit 14.

  Note that the A / D converter 12 and the orthogonal demodulator 13 may be configured in reverse order. In this case, the quadrature demodulator 13 performs quadrature demodulation on the analog IF signal and outputs an equivalent baseband signal of I and Q, and the two A / D converters 12 receive the I and Q input from the quadrature demodulator 13. Are converted into digital signals using the equivalent baseband signal.

  The subtractor 14 subtracts the wraparound replica signal input from the FBF 16 from the equivalent baseband signal input from the quadrature demodulator 13 to output an equivalent baseband signal with the wraparound canceled. The equivalent baseband signal output from the subtracting unit 14 is input to the FFF 15. The FFF 15 performs the filtering process on the equivalent baseband signal input from the subtracting unit 14 using the filter coefficient of the FFF 15 input from the filter coefficient control unit 20, thereby equalizing the equivalent baseband with multipath distortion. Outputs a band signal. The equivalent baseband signal output from the FFF 15 is divided into three and input to the quadrature modulation unit 17, the filter coefficient control unit 20, and the FBF 16, respectively. The FBF 16 performs a filtering process on the equivalent baseband signal input from the FFF 15 using the filter coefficient of the FBF 16 input from the filter coefficient control unit 20 to output a wraparound replica signal. The wraparound replica signal output from the FBF 16 is input to the subtracting unit 14.

  The quadrature modulation unit 17 performs quadrature modulation on the equivalent baseband signal input from the FFF 15 and outputs it as a digital IF signal. The digital IF signal output from the quadrature modulation unit 17 is input to the D / A conversion unit 18. The D / A converter 18 converts the digital IF signal input from the quadrature modulator 17 into an analog signal and outputs the analog signal. The IF signal output from the D / A converter 18 is input to the frequency converter 19. The frequency conversion unit 19 converts the IF signal input from the D / A conversion unit 18 into an RF band transmission signal and outputs the signal to the outside.

  The filter coefficient control unit 20 receives the equivalent baseband signal output from the FFF 15 and generates and outputs the filter coefficient of the FFF 15 and the filter coefficient of the FBF 16. The filter coefficient of the FFF 15 output from the filter coefficient control unit 20 is input to the FFF 15, and the filter coefficient of the FBF 16 is input to the FBF 16.

  The filter coefficient control unit 20 includes an effective symbol period extraction unit 21, an FFT unit 22, a frequency characteristic calculation unit (channel response calculation unit) 23, a cancellation residual calculation unit 29, an IFFT unit 30, a multiplication unit 31, an addition unit 32, a delay A unit 33, an equalization error calculation unit 34, an IFFT unit 35, a multiplication unit 36, an addition unit 37, a delay unit 38, and a convolution operation unit 39. The IFFT unit 30, the multiplication unit 31, the addition unit 32, and the delay unit 33 constitute an FBF coefficient generation unit, and the IFFT unit 35, the multiplication unit 36, the addition unit 37, the delay unit 38, and the convolution operation unit 39 constitute an FFF coefficient generation unit. Configure.

  The effective symbol period extraction unit 21 extracts and outputs a signal in a period corresponding to an effective symbol period in one OFDM transmission symbol period from the equivalent baseband signal input from the FFF 15. The OFDM signal in the time domain of the effective symbol period output from the effective symbol period extraction unit 21 is input to the FFT unit 22. The FFT unit 22 performs FFT on the OFDM signal in the time domain of the effective symbol period input from the effective symbol period extraction unit 21, converts the OFDM signal into a carrier symbol that is a frequency domain signal, and outputs the carrier symbol. The carrier symbol output from the FFT unit 22 is input to the frequency characteristic calculation unit 23.

  The frequency characteristic calculation unit 23 estimates a channel response in consideration of the phase correction component performed in units of segments by the transmission side from the carrier symbol input from the FFT unit 22, and removes a multipath distortion component from the estimated channel response. The generated channel response is generated and output, and the channel response including only the propagation path characteristics is generated and output from the estimated channel response. The channel response from which the multipath distortion component output from the frequency characteristic calculation unit 23 is removed is input to the cancellation residual calculation unit 29, and the channel response including only the propagation path characteristic is input to the equalization error calculation unit 34.

  The frequency characteristic calculation unit 23 includes a phase compensation unit 24, a channel estimation unit 25, an LPF (Low Pass Filter) unit 26, a division unit 27, and an FFT window position correction unit 28. The phase compensation unit 24 multiplies (complex multiplication) the carrier symbol input from the FFT unit 22 by a phase rotation correction amount obtained by reversing the sign of the phase correction component applied in segment units by the transmission side. To compensate. That is, the phase of the carrier symbol is compensated by applying a phase correction opposite to that on the transmission side. The carrier symbol output from the phase compensation unit 24 is input to the channel estimation unit 25.

  The channel estimation unit 25 estimates a channel response from the carrier symbol input from the phase compensation unit 24 and outputs it. As a result, a channel response is estimated in consideration of the phase correction component applied in segment units by the transmission side. The channel response output from the channel estimation unit 25 is divided into two, one being input to the LPF unit 26 and the other being input to the division unit 27.

  The LPF unit 26 extracts a low frequency component from the channel response input from the channel estimation unit 25 and outputs it. The low-frequency component of the channel response output from the LPF unit 26 is divided into two, one being input to the division unit 27 and the other being input to the FFT window position correction unit 28. The division unit 27 divides the channel response input from the channel estimation unit 25 by the low frequency component of the channel response input from the LPF unit 26 for each subcarrier of the OFDM signal, thereby multipath distortion from the channel response. Remove components and output. The channel response from which the multipath distortion component output from the division unit 27 is removed is input to the cancellation residual calculation unit 29.

  The low-frequency component of the channel response output from the LPF unit 26 indicates a channel response including only the multipath component in the delay time range that can be equalized by the main wave and the FFF 15, and the multipath distortion output from the division unit 27. The channel response from which the component is removed indicates a channel response that includes only the main wave and the sneak wave component in the delay time range in which the replica can be generated by the FBF 16. Details will be described later.

  The FFT window position correction unit 28 performs phase rotation due to a shift between the effective symbol period and the effective symbol period extracted by the effective symbol period extraction unit 21 included in the low-frequency component of the channel response input from the LPF unit 26. The component is corrected to output a channel response in which the phase rotation component due to the shift of the FFT window position is corrected and includes only the propagation path characteristic. A channel response in which the phase rotation component due to the shift of the FFT window position output from the FFT window position correction unit 28 is corrected and includes only the propagation path characteristic is input to the equalization error calculation unit 34.

  The cancellation residual calculation unit 29 is a difference between the actual wraparound channel characteristic and the channel characteristic realized by the FBF 16 from the channel response from which the multipath distortion component input from the division unit 27 is removed. The cancellation residual (frequency domain signal) is calculated and output. The wraparound cancellation residual output from the cancellation residual calculation unit 29 is input to the IFFT unit 30. The IFFT unit 30 performs IFFT on the wraparound cancellation residual input from the cancellation residual calculation unit 29, converts it to a change in the impulse response of the wraparound propagation path, and outputs it. A change in the impulse response of the sneak path output from the IFFT unit 30 is input to the multiplication unit 31. The multiplier 31 multiplies the change in the impulse response of the sneak path input from the IFFT unit 30 by a predetermined constant and outputs the result. The output signal of the multiplication unit 31 is input to the addition unit 32. The adder 32 adds the change of the impulse response of the sneak path input from the multiplier 31 to the filter coefficient of the FBF 16 before the unit coefficient update time input from the delay unit 33, and outputs the result as the filter coefficient of the FBF 16 To do. The filter coefficients of the FBF 16 output from the adder 32 are divided into two, one being input to the FBF 16 and the other being input to the delay unit 33.

  The equalization error calculation unit 34 inputs a channel response that is input from the FFT window position correction unit 28 and that is a channel response in which the phase rotation component due to the shift of the FFT window position is corrected and includes only the propagation path characteristic, An equalization error that is a difference between the equalization process to be actually performed and the equalization process realized by the FFF 15 is calculated and output. The equalization error output from the equalization error calculation unit 34 is input to the IFFT unit 35. The IFFT unit 35 performs IFFT on the equalization error input from the equalization error calculation unit 34, converts it into a change in the impulse response of the equalization error, and outputs it. A change in the impulse response of the equalization error output from the IFFT unit 35 is input to the multiplication unit 36. The multiplier 36 multiplies the change in the impulse response of the equalization error input from the IFFT unit 35 by a predetermined constant and outputs the result. The output signal of the multiplication unit 36 is input to the addition unit 37. The adder 37 adds the change in the impulse response of the equalization error input from the multiplier 36 to the filter coefficient of the FFF 15 before the unit coefficient update time input from the delay unit 38, and outputs the result as the filter coefficient of the FFF 15 To do. The filter coefficients of the FFF 15 output from the adder 37 are divided into two, one being input to the convolution calculator 39 and the other being input to the delay unit 38.

  The convolution operation unit 39 includes a multiplication unit 40 and a digital SAW (Surface Acoustic Wave filter) filter coefficient storage unit 41. The digital SAW filter coefficient storage unit 41 stores a filter coefficient of a digital filter having a stopband attenuation equivalent to that of an analog SAW filter, and stores a predetermined filter coefficient having a predetermined filter length. . The multiplier 40 performs a convolution operation on the filter coefficient of the FFF 15 input from the adder 37 and a predetermined filter coefficient input from the digital SAW filter coefficient storage 41 and outputs the result. The filter coefficient of the FFF 15 output from the multiplier 40 is input to the FFF 15.

  As described above, according to the wraparound canceller 1 of the first embodiment illustrated in FIG. 1, the phase compensation unit 24 of the frequency characteristic calculation unit 23 included in the filter coefficient control unit 20 is configured to carry the equivalent baseband signal carrier from the FFF 15. The symbol is multiplied by the phase rotation correction amount obtained by reversing the sign of the phase correction component applied in segment units by the transmission side, that is, the phase of the carrier symbol is compensated by adding the phase correction opposite to that of the transmission side. I tried to do it. The channel estimation unit 25 estimates the channel response from the carrier symbol from the phase compensation unit 24. As a result, it is possible to estimate the channel response in consideration of the phase correction component applied in segment units by the transmission side.

  Then, the LPF unit 26 extracts a low frequency component from the estimated channel response, and the division unit 27 removes the multipath distortion component by dividing the estimated channel response by the extracted low frequency component. The channel response is generated, and the components from the cancellation residual calculation unit 29 to the delay unit 33 generate the filter coefficient of the FBF 16 from the channel response from which the multipath distortion component is removed. Further, the FFT window position correction unit 28 calculates a phase rotation component due to a shift between the effective symbol period and the effective symbol period extracted by the effective symbol period extraction unit 21 included in the extracted low frequency component of the channel response. Correcting and generating a channel response in which the phase rotation component due to the FFT window position shift is corrected and including only the propagation path characteristic, and the components from the equalization error calculator 34 to the convolution calculator 39 The FFF15 filter coefficients are generated from the channel response including only the propagation path characteristics.

  As a result, the wraparound canceller 1 generates the FBF 16 from the channel response in consideration of the phase correction component when the broadcast wave in which the phase correction is performed in the segment unit and a plurality of unit transmission waves are connected is received from the transmission side. The filter processing is performed using the filter coefficient of the FBF 16 to generate a wraparound replica signal, the subtracting unit 14 removes the wraparound replica signal from the signal obtained by orthogonal demodulation of the received signal, and the FFF 15 performs phase correction. Multipath distortion is equalized by performing filter processing on the signal from the subtractor 14 using the filter coefficient of the FFF 15 generated from the channel response considering the component. Therefore, when receiving a broadcast wave in which a plurality of unit transmission waves are connected, it is possible to remove the wraparound due to coupling between the transmitting and receiving antennas and equalize the distortion due to multipath.

[Sneak canceller of Example 2]
Next, a wraparound canceller according to the second embodiment will be described. FIG. 2 is a block diagram illustrating a configuration of the wraparound canceller according to the second embodiment. The sneak canceller 2 includes a frequency conversion unit 11, an A / D conversion unit 12, an orthogonal demodulation unit 13, a subtraction unit 14, an FFF 15, an FBF 16, an orthogonal modulation unit 17, a D / A conversion unit 18, a frequency conversion unit 19, and a filter coefficient. A control unit 42 is provided. When comparing the wraparound canceller 1 of the first embodiment shown in FIG. 1 and the wraparound canceller 2 of the second embodiment, the wraparound canceller 2 of the second embodiment includes a filter coefficient control unit 20 provided in the wraparound canceller 1 of the first embodiment, and Are different in that they include different filter coefficient control units 42. Since the components other than the filter coefficient control unit 42 of the wraparound canceller 2 are the same as those of the wraparound canceller 1 of the first embodiment, description thereof will be omitted.

  The filter coefficient control unit 42 receives the equivalent baseband signal from the FFF 15 and generates and outputs the filter coefficient of the FFF 15 and the filter coefficient of the FBF 16. The filter coefficient of the FFF 15 output from the filter coefficient control unit 42 is input to the FFF 15, and the filter coefficient of the FBF 16 is input to the FBF 16.

  The filter coefficient control unit 42 includes an effective symbol period extraction unit 21, an FFT unit 22, a frequency characteristic calculation unit 43, a cancellation residual calculation unit 29, an IFFT unit 30, a multiplication unit 31, an addition unit 32, a delay unit 33, and an equalization error. A calculation unit 34, an IFFT unit 35, a multiplication unit 36, an addition unit 37, a delay unit 38, and a convolution operation unit 39 are provided. The frequency characteristic calculation unit 43 includes a channel estimation unit 44, an LPF unit 26, a division unit 27, and an FFT window position. A correction unit 28 is provided. When the filter coefficient control unit 20 of the first embodiment shown in FIG. 1 and the filter coefficient control unit 42 of the second embodiment are compared, the frequency characteristic calculation unit 43 in the filter coefficient control unit 42 of the second embodiment is similar to that of the first embodiment. A difference is that a channel estimation unit 44 is provided instead of the phase compensation unit 24 and the channel estimation unit 25 provided in the frequency characteristic calculation unit 23 in the filter coefficient control unit 20, and the phase compensation unit 24 is not provided. Since components other than the channel estimation unit 44 of the filter coefficient control unit 42 are the same as those of the filter coefficient control unit 20 of the first embodiment, description thereof will be omitted.

  The channel estimation unit 44 of the frequency characteristic calculation unit 43 uses an SP (Scattered Pilot) signal that takes into account the phase correction component applied in segment units by the transmission side from the carrier symbol input from the FFT unit 22. To estimate the channel response. In this case, the phase is compensated by multiplying the reference SP signal by the same phase rotation correction amount as the phase correction component applied in segment units by the transmission side. That is, the phase of the SP signal is compensated by applying the same phase correction to the SP signal as on the transmission side. As a result, it is possible to estimate the channel response in consideration of the phase correction component applied in segment units by the transmission side. The channel response output by the channel estimation unit 44 is divided into two, one being input to the division unit 27 and the other being input to the LPF unit 26.

  As described above, according to the wraparound canceller 2 of the second embodiment illustrated in FIG. 2, the channel estimation unit 44 of the frequency characteristic calculation unit 43 included in the filter coefficient control unit 42 uses the carrier symbol from the FFT unit 22 based on the carrier symbol. The channel response is estimated using an SP signal that takes into account the phase correction component applied in segment units by the transmission side.

  As a result, the wraparound canceller 2, like the wraparound canceller 1, receives a broadcast wave in which a phase correction is performed on a segment basis and a plurality of unit transmission waves are connected from the transmission side, the FBF 16 uses the phase correction component. Filter processing is performed using the filter coefficient of the FBF 16 generated from the channel response taken into consideration, thereby generating a wraparound replica signal, and the subtracting unit 14 removes the wraparound replica signal from the signal obtained by orthogonal demodulation of the received signal. The FFF 15 equalizes multipath distortion by performing filter processing on the signal from the subtracting unit 14 using the filter coefficient of the FFF 15 generated from the channel response considering the phase correction component. Therefore, when receiving a broadcast wave in which a plurality of unit transmission waves are connected, it is possible to remove the wraparound due to coupling between the transmitting and receiving antennas and equalize the distortion due to multipath.

  Hereinafter, the case where the operation of the wraparound canceller 1 of the first embodiment shown in FIG. 1 and the wraparound canceller 2 of the second embodiment shown in FIG. 2 are applied to terrestrial digital audio broadcasting will be described. However, the following is a description of the principle, and description of basic parts such as frequency conversion, A / D conversion, D / A conversion, quadrature modulation / demodulation, and transmission / reception unit is omitted, and synchronous reproduction is performed with sufficient accuracy. It shall be realized. Since these components are well-known techniques, description thereof is omitted.

[Phase Compensator: Example 1]
First, the phase compensation unit 24 provided in the wraparound canceller 1 of the first embodiment shown in FIG. 1 will be described in detail. The phase rotation correction amount of the phase correction component applied in segment units on the transmission side is a predetermined value, and the difference between the mode, the guard interval ratio, and the center frequency of the transmission signal and the center frequency of the segment Therefore, it is shown in ARIB STD-B29 “Transmission method of digital terrestrial audio broadcasting” Table 4.3-1. If the phase rotation compensation amount (phase rotation correction amount) described in this table is φ, the phase compensation unit 24 reverses the sign of the phase rotation correction amount φ with respect to the carrier symbol input from the FFT unit 22. The phase is compensated by performing phase rotation of the corrected amount (−φ).

とすると、位相補償部２４の入出力は次式で表すことができる。

Carrier symbols before and after phase compensation by the phase compensation unit 24 are respectively

Then, the input / output of the phase compensation unit 24 can be expressed by the following equation.

[Channel estimation unit: Example 1]
Next, the channel estimation unit 25 provided in the wraparound canceller 1 according to the first embodiment illustrated in FIG. 1 will be described in detail. FIG. 9 shows allocation of specific symbols to specific subcarriers in ISDB-T (Integrated Services Digital Broadcasting-Terrestrial) and DVB-T (Digital Video Broadcasting-Terrestrial), which are broadcasting systems for digital terrestrial television broadcasting. It is a figure which shows arrangement | positioning of SP currently used. In FIG. 9, SP is indicated by a black circle, and other carrier symbols such as data symbols are indicated by white circles. Hereinafter, regarding the SP arrangement, the interval in the subcarrier direction in consecutive symbols is N _f , and the interval in the symbol direction in the same subcarrier is N _t . Since the amplitude and phase of the SP are determined in advance, the wraparound canceller 1 on the reception side can generate the same signal. The same applies to the wraparound canceller 2 of the second embodiment.

(First configuration)
FIG. 3 is a block diagram illustrating a first configuration of the channel estimation unit 25 according to the first embodiment. The channel estimation unit 25-1 includes an SP signal (pilot signal) extraction unit 45, an SP signal (pilot signal) generation unit 46, a division unit 47, and an interpolation unit 48, and uses the SP shown in FIG. Estimate the channel response.

ただし、ｍｏｄは剰余を示す。以下、前記式（２）を満足するｉ，ｋをそれぞれｉ_ｐ，ｋ_ｐとする。 If the symbol number of the subcarrier assigned to the SP is i and the subcarrier number is k, the relationship between i and k can be expressed by the following equation.

However, mod indicates a remainder. In the following, i and k that satisfy the above equation (2) are assumed to be i _p and k _p , respectively.

  The SP signal extraction unit 45 receives the carrier symbol from the phase compensation unit 24, extracts the SP signal from the carrier symbol, and outputs the SP signal to the division unit 47. The SP signal generation unit 46 generates a reference SP signal having the same amplitude and phase as the SP signal at the time of signal generation on the transmission side, and outputs the reference SP signal to the division unit 47. The division unit 47 divides the SP signal extracted by the SP signal extraction unit 45 by the reference SP signal generated by the SP signal generation unit 46 and outputs a channel response to the interpolation unit 48. The interpolation unit 48 interpolates the channel response generated by the division unit 47 in the symbol direction and the carrier direction.

とし、ＳＰ信号生成部４６により生成される基準ＳＰ信号を

とすると、シンボル番号ｉ_ｐ、サブキャリヤ番号ｋ_ｐにおけるチャネル応答

は、次式で表される。

Here, the SP signal extracted by the SP signal extraction unit 45 is

And the reference SP signal generated by the SP signal generator 46 is

The channel response at symbol number i _p and subcarrier number k _p

Is expressed by the following equation.

  The channel estimation unit 25-1 uses the SP adopted in the ISDB-T method as the reference signal for calculating the channel response, but the present invention is not limited to this. . Another carrier symbol that has a known amplitude and phase and can be generated by the wraparound canceller 1 on the receiving side may be used as the reference signal. The same applies to the channel estimation unit 25-2 and the channel estimation units 44-1 to 44-4 described later.

(Second configuration)
FIG. 4 is a block diagram illustrating a second configuration of the channel estimation unit 25 according to the first embodiment. The channel estimation unit 25-2 includes a channel equalization unit 49, a determination unit 55, and a division unit 56. Hereinafter, the symbol number i is omitted.

The channel equalization unit 49 includes an SP signal (pilot signal) extraction unit 50, an SP signal (pilot signal) generation unit 51, a division unit 52, an interpolation unit 53, and a division unit 54. The SP signal extraction unit 50 receives the carrier symbol from the phase compensation unit 24, extracts the SP signal from the carrier symbol, and outputs the SP signal to the division unit 52. The SP signal generation unit 51 generates a reference SP signal having the same amplitude and phase as the SP signal at the time of signal generation on the transmission side, and outputs the reference SP signal to the division unit 52. The division unit 52 divides the SP signal extracted by the SP signal extraction unit 50 by the reference SP signal generated by the SP signal generation unit 51 and outputs the result to the interpolation unit 53. The interpolation unit 53 interpolates the channel response generated by the division unit 52 in the symbol direction and the carrier direction. The division unit 54 divides a carrier symbol (hereinafter referred to as a data symbol) Y _k other than the SP signal among the carrier symbols from the phase compensation unit 24 by the channel response F _k input from the interpolation unit 53. Channel equalization.

ここで、Ｄ_ｋは、サブキャリヤ番号ｋのサブキャリヤの周波数における送信シンボルを示し、Ｎ_ｋは雑音を示す。前記式（４）からわかるように、データシンボルＺ_ｋは、第２項により信号点Ｄ_ｋを中心に分散する。 If the data symbol after channel equalization by the channel equalization unit 49 is Z _k , Z _k is expressed by the following equation.

Here, D _k represents a transmission symbol at the frequency of the subcarrier of subcarrier number k, and N _k represents noise. As can be seen from the equation (4), the data symbol Z _k is distributed around the signal point D _k by the second term.

として出力する。

ここで、ｄｅｃ（ｙ）はしきい値判定の関数であり、信号空間においてｙに最も近い送信信号を返す。前記式（５）において、判定誤りがないと仮定すると

となる。 The determination unit 55 receives the carrier symbol after channel equalization from the division unit 54 and, as shown in the following equation (5), a known transmission having the smallest Euclidean distance in the signal space from the carrier symbol after equalization. Symbol, estimate of transmitted symbol

Output as.

Here, dec (y) is a threshold determination function, and returns a transmission signal closest to y in the signal space. Assuming that there is no determination error in the equation (5),

It becomes.

で除算し、チャネル応答Ｆ_ｋを求める。

The division unit 56 uses the carrier symbol Y _k from the phase compensation unit 24 as the estimated value of the transmission symbol from the determination unit 55 as shown in the following equation (6).

In division, and obtains the channel response _{F k.}

をＳＰと同様に基準信号として用いることで、チャネル応答Ｆ_ｋを求めることができる。 This gives an estimate of the transmitted symbol

Is used as a reference signal in the same manner as SP, so that the channel response F _k can be obtained.

[Channel estimation unit: Example 2]
Next, the channel estimation unit 44 provided in the wraparound canceller 2 according to the second embodiment illustrated in FIG. 2 will be described in detail.
(First configuration)
FIG. 5 is a block diagram illustrating a first configuration of the channel estimation unit 44 in the second embodiment. The channel estimation unit 44-1 includes an SP signal extraction unit 45, an SP signal generation unit 46, a division unit 47, an interpolation unit 48, and a phase compensation unit 57. A channel response is obtained using the SP shown in FIG. presume. When the channel estimation unit 25-1 of the first embodiment shown in FIG. 3 and the channel estimation unit 44-1 of the second embodiment are compared, the channel estimation unit 44-1 includes the SP signal generation unit 46 and the division unit 47. It is different from the channel estimation unit 25-1 in that a phase compensation unit 57 is provided in between. Since the SP signal extraction unit 45, the SP signal generation unit 46, the division unit 47, and the interpolation unit 48 have already been described, description thereof is omitted here.

  The phase compensator 57 performs the same process as the process in which the transmission side performs phase correction on a segment basis for the reference SP signal generated by the SP signal generator 46. That is, the phase compensation unit 57 compensates the phase by multiplying the reference SP signal input from the SP signal generation unit 46 by the phase rotation correction amount of the phase correction component applied in units of segments by the transmission side. That is, the phase of the reference SP signal is compensated by applying the same correction as the phase correction on the transmission side. The reference SP signal output from the phase compensation unit 57 is input to the division unit 47.

とすると、位相補償部５７の入出力は次式で表すことができる。

As described above, the phase rotation correction amount of the phase correction component that is applied in segment units on the transmission side is a predetermined value. This phase rotation correction amount is φ, and SP signals before and after phase compensation by the phase compensation unit 57 are respectively

Then, the input / output of the phase compensation unit 57 can be expressed by the following equation.

(Second configuration)
FIG. 6 is a block diagram illustrating a second configuration of the channel estimation unit 44 according to the second embodiment. The channel estimation unit 44-2 includes an SP signal extraction unit 45, a division unit 47, an interpolation unit 48, and a compensated SP signal (compensated pilot signal) generation unit 58, and uses the SP shown in FIG. Estimate the channel response. When the channel estimation unit 44-1 shown in FIG. 5 is compared with the channel estimation unit 44-2, the channel estimation unit 44-2 includes an SP signal generation unit 46 and a phase compensation unit included in the channel estimation unit 44-1. The difference is that a compensated SP signal generation unit 58 is provided instead of 57. Since the SP signal extraction unit 45, the division unit 47, and the interpolation unit 48 have already been described, description thereof is omitted here.

  The SP signal is a predetermined value, and as described above, the phase rotation correction amount of the phase correction component applied in units of segments on the transmission side is also a predetermined value. For this reason, the compensated SP signal generation unit 58 can generate, as a reference SP signal, a phase-compensated SP signal for a predetermined amount of phase rotation correction. Specifically, the compensated SP signal generation unit 58 generates, as a reference SP signal, an SP signal having a known amplitude and phase-compensated for the known phase by the same amount of phase rotation correction as the transmission side. To do. The phase-compensated reference SP signal output from the compensated SP signal generation unit 58 is input to the division unit 47. As a result, the phase compensation unit 57 is not required, so that phase compensation processing for all subcarriers or all SPs is unnecessary, and the circuit scale can be reduced. Furthermore, there is an advantage that it is not necessary to add a circuit to the conventional wraparound canceller, and it is only necessary to rotate the SP phase using a predetermined value.

(Third configuration)
FIG. 7 is a block diagram illustrating a third configuration of the channel estimation unit 44 in the second embodiment. The channel estimation unit 44-3 includes a channel equalization unit 59, a determination unit 55, a multiplication unit (phase compensation unit) 61, and a division unit 56. The channel equalization unit 59 includes an SP signal extraction unit 50, an SP signal generation unit 51, a phase compensation unit 60, a division unit 52, an interpolation unit 53, and a division unit 54. Comparing the channel estimation unit 25-2 of the first embodiment shown in FIG. 4 with the channel estimation unit 44-3 of the second embodiment, the channel estimation unit 44-3 has a relationship between the SP signal generation unit 51 and the division unit 52. It differs from the channel estimation unit 25-2 in that a phase compensation unit 60 is provided between them, and a multiplication unit 61 is provided between the determination unit 55 and the division unit 56. Since the SP signal extraction unit 50, the SP signal generation unit 51, the division unit 52, the interpolation unit 53, the division unit 54, the determination unit 55, and the division unit 56 have been described, description thereof is omitted here.

  Similarly to the phase compensation unit 57 shown in FIG. 5, the phase compensation unit 60 performs the same processing as the processing in which the transmission side performs phase correction on a segment basis for the reference SP signal generated by the SP signal generation unit 51. . The reference SP signal output from the phase compensation unit 60 is input to the division unit 52. Since the phase compensation unit 60 is the same as the phase compensation unit 57, the detailed description thereof is omitted.

  Multiplier 61 performs the same processing as the processing in which the transmission side performs phase correction on a segment basis for the estimated value of the transmission symbol from determination unit 55. That is, the multiplying unit 61 compensates the phase by multiplying the estimated value of the transmission symbol input from the determining unit 55 by the phase rotation correction amount of the phase correction component applied in units of segments by the transmitting side. That is, the phase of the estimated value of the transmission symbol is compensated by applying the same correction as the phase correction on the transmission side. The estimated value of the transmission symbol output from the multiplier 61 is input to the divider 56.

とすると、乗算部６１の入出力は次式で表すことができる。

As described above, the phase rotation correction amount of the phase correction component that is applied in segment units on the transmission side is a predetermined value. This phase rotation correction amount is φ, and the estimated values of transmission symbols before and after phase compensation by the multiplier 61 are respectively

Then, the input / output of the multiplication unit 61 can be expressed by the following equation.

(Fourth configuration)
FIG. 8 is a block diagram illustrating a fourth configuration of the channel estimation unit 44 in the second embodiment. The channel estimation unit 44-4 includes a channel equalization unit 59, a determination unit (compensated determination unit) 62, and a division unit 56. The channel equalization unit 59 includes an SP signal extraction unit 50, an SP signal generation unit 51, a phase compensation unit 60, a division unit 52, an interpolation unit 53, and a division unit 54. When the channel estimation unit 44-3 shown in FIG. 7 is compared with the channel estimation unit 44-4, the channel estimation unit 44-4 replaces the determination unit 55 and the multiplication unit 61 provided in the channel estimation unit 44-3. The difference is that a determination unit 62 is provided. Since each component of the channel equalization unit 59 and the division unit 56 have already been described, description thereof is omitted here.

  The symbol determination values in the determination units 55 and 62 are predetermined values, and the phase rotation correction amount is also a predetermined value as described above. Therefore, the determination unit 62 can generate a phase determination-completed symbol determination value for a predetermined phase rotation correction amount. Specifically, the determination unit 62 inputs the carrier symbol after channel equalization from the division unit 54, specifies a known transmission symbol having the smallest Euclidean distance on the signal space from the carrier symbol after equalization, A phase-compensated transmission symbol corresponding to the identified transmission symbol, that is, a transmission symbol that has been phase-compensated for the same phase rotation correction amount as that of the transmission side is determined and output as an estimated value of the transmission symbol. The estimated value of the phase-compensated transmission symbol output from the determination unit 62 is input to the division unit 56. This eliminates the need for the multiplication unit 61 that performs phase compensation, which eliminates the need for phase compensation processing for all subcarriers, and reduces the circuit scale. Further, there is an advantage that it is not necessary to add a circuit to the conventional wraparound canceller, and it is only necessary to rotate the phase of the estimated value of the transmission symbol using a predetermined value.

  In the channel estimation unit 44-4 shown in FIG. 8, the compensated SP signal generation unit 58 shown in FIG. 6 is used instead of the SP signal generation unit 51 and the phase compensation unit 60 provided in the channel equalization unit 59. You may make it prepare. This eliminates the need for the phase compensator 57 that performs phase compensation, thereby further reducing the circuit scale.

[Channel response delay time separation: Examples 1 and 2]
Next, the LPF unit 26 and the division unit 27 provided in the wraparound cancellers 1 and 2 of the first and second embodiments shown in FIGS. FIG. 10 is a block diagram showing a first configuration of the LPF unit 26. The LPF unit 26-1 extracts a low frequency component of the channel response input from the channel estimation units 25 and 44.

  FIG. 11 is a block diagram showing a second configuration of the LPF unit 26. The LPF unit 26-2 includes an IFFT unit 63, a window function unit 64, and an FFT unit 65. The IFFT unit 63 performs IFFT on the channel response input from the channel estimation units 25 and 44 and converts it into a delay profile that is a time domain expression. The window function unit 64 multiplies the delay profile converted by the IFFT unit 63 by a predetermined window function to the component in the delay time range that can be equalized by the FFT unit 65 in the subsequent stage. The FFT unit 65 performs FFT on the component in the delay time range multiplied by the window function unit 64 and converts it again into a channel response that is a frequency domain representation. As a result, the LPF unit 26-2 can extract the low-frequency component of the channel response in the same manner as the LPF unit 26-1 illustrated in FIG.

ここで、Ｆ_ｋは、チャネル推定部２５，４４から入力されるチャネル応答を示し、

は、サブキャリヤ番号ｋについて、チャネル応答の低域成分である、主波とＦＦＦ１５によって等化可能な遅延時間範囲のマルチパス成分のみが含まれるチャネル応答を示す。 Input / output of the LPF units 26-1 and 26-2 can be expressed by the following equation.

Here, F _k indicates a channel response input from the channel estimation units 25 and 44,

Indicates a channel response including only a multipath component in the delay time range that can be equalized by the main wave and the FFF 15, which is a low frequency component of the channel response, for the subcarrier number k.

で除算し、
ＯＦＤＭ信号のサブキャリヤ毎のサブキャリヤ番号ｋについて、マルチパス歪み成分を除去したチャネル応答である、主波とＦＢＦ１６によってそのレプリカを生成可能な遅延時間範囲の回り込み波成分のみが含まれるチャネル応答

を生成する。

As shown in the following equation, the division unit 27 converts the channel response F _k input from the channel estimation units 25 and 44 into the low-frequency component (mainly) of the channel response input from the LPF units 26-1 and 26-2. Channel response including only multipath components in the delay time range that can be equalized by wave and FFF15)

Divide by
For the subcarrier number k for each subcarrier of the OFDM signal, the channel response is a channel response from which a multipath distortion component is removed, and includes a main wave and a sneak wave component in a delay time range in which a replica can be generated by the FBF 16

Is generated.

と、
マルチパス歪み成分を除去したチャネル応答（主波とＦＢＦ１６によってそのレプリカを生成可能な遅延時間範囲の回り込み波成分のみが含まれるチャネル応答）

とに分離することができる。詳細については、特開２００４−３２０６７７号公報を参照されたい。 As described above, according to the LPF units 26-1 and 26-2 and the division unit 27, the channel response F _k estimated by the channel estimation units 25 and 44 can be equalized by the low frequency component (main wave and FFF 15). Channel response including only multipath components in a wide delay time range)

When,
Channel response from which multipath distortion components have been removed (channel response including only the sneak wave component in the delay time range in which the replica can be generated by the main wave and FBF 16)

And can be separated. For details, refer to Japanese Patent Application Laid-Open No. 2004-320677.

[Calculation of wraparound cancellation residual and update of filter coefficient of FBF: Examples 1 and 2]
Next, the details from the cancellation residual calculation unit 29 to the delay unit 33 provided in the wraparound cancellers 1 and 2 of the first and second embodiments shown in FIGS. The cancellation residual calculation unit 29 calculates a wraparound cancellation residual δW _k for the subcarrier number k as shown in the following equation.

When IFFT section 30 converts cancellation residual δW _k into a time domain signal by IFFT processing, as shown in the following equation, IFFT unit 30 obtains impulse response δw of the wraparound cancellation residual, which is a change in the impulse response of the wraparound propagation path. Can do.

ここで、μ_ｂは雑音抑圧のための適応係数を示す。 Assuming that the current time is i, an arbitrary discrete time is n, and the filter coefficient of the FBF 16 is w (i, n), the multiplication unit 31, the addition unit 32, and the delay unit 33 are as follows. The coefficient w (i + 1, n) is updated.

Here, μ _b represents an adaptation coefficient for noise suppression.

[Calculation of equalization error and update of filter coefficient of FFF: Examples 1 and 2]
Next, the equalization error calculation unit 34 to the delay unit 38 included in the wraparound cancellers 1 and 2 of the first and second embodiments illustrated in FIGS. The equalization error calculation unit 34 calculates an equalization error δH _k for the subcarrier number k as shown in the following equation.

When the equalization error δH _k is converted into a time domain signal by IFFT processing, the IFFT unit 35 obtains an equalization error impulse response δh, which is a change in the equalization error impulse response, as shown in the following equation. it can.

ここで、μ_ｆは雑音抑圧のための適応係数を示す。 Assuming that the current time is i, the arbitrary discrete time is n, and the filter coefficient of the FFF 15 is h (i, n), the multiplication unit 36, the addition unit 37, and the delay unit 38 are as shown in the following equation. The filter coefficient h (i + 1, n) of the FFF 15 output from the above is updated.

Here, μ _f represents an adaptation coefficient for noise suppression.

[Convolution operation unit: Examples 1 and 2]
Next, the convolution operation unit 39 provided in the wraparound cancellers 1 and 2 of the first and second embodiments shown in FIGS. The purpose of the processing of the convolution operation unit 39 is to prevent the bandwidth of the FFF 15 from exceeding any of the bandwidth of the wraparound loop and the cancellation loop.

ここで、ｌは非適応狭帯域フィルタｇ（ｎ）のフィルタ長を示す。 The convolution operation unit 39 is a non-adaptive filter, and a narrowband filter whose bandwidth does not exceed any of the wraparound loop and the cancellation loop is determined in advance as g (n), and the equation (16) As shown in the following equation, a convolution operation is performed on the filter coefficient h (n) of the FFF 15 generated by the above.

Here, l indicates the filter length of the non-adaptive narrowband filter g (n).

  In general, when a non-adaptive narrowband filter is connected in cascade to the FFF 15, the processing delay by the wraparound cancellers 1 and 2 becomes large. However, as described above, since the convolution operation unit 39 performs the convolution operation between the filter coefficients in the time domain, the processing delay by the wraparound cancellers 1 and 2 is not increased, and the bandwidth of the FFF 15 is canceled and canceled. It is possible not to exceed any bandwidth of the loop.

[Repeater]
Next, a relay device using the wraparound cancellers 1 and 2 of the first and second embodiments shown in FIGS. FIG. 12 is a block diagram showing the configuration of the relay device. The relay apparatus 101 includes a reception antenna 102, a reception filter 103, a reception conversion unit (reception unit) 104, sneak cancellers 1 and 2, a transmission conversion unit (transmission unit) 105, a PA (amplification unit) 106, a transmission filter 107, and a transmission. An antenna 108 is provided.

  The relay apparatus 101 of the broadcast wave relay station including the wraparound cancellers 1 and 2 according to the first and second embodiments receives the desired wave transmitted from the upper station by the reception antenna 102. The reception filter 103 receives the reception signal output from the reception antenna 102 via a feeder cable, and removes unnecessary signal components outside the frequency band of the desired wave. The reception conversion unit 104 receives the output signal of the reception filter 103, performs AGC amplification so that the output level becomes constant, and then performs frequency conversion to generate and output an IF signal. As the center frequency of this IF signal, 37.15 MHz is generally used. The IF signal output from the reception conversion unit 104 is input to the wraparound cancellers 1 and 2.

  The wraparound cancellers 1 and 2 receive the IF signal from the reception conversion unit 104. The frequency conversion unit 11 of the wraparound cancellers 1 and 2 performs frequency conversion on the IF signal input from the reception conversion unit 104, converts the IF signal into a second IF signal, and outputs the second IF signal. As the center frequency of the second IF signal, 512/63 (8.127689) MHz is generally used. The A / D converter 12 receives the second IF signal from the frequency converter 11, performs A / D conversion, and outputs a digital IF signal. The quadrature demodulator 13 receives the digital IF signal from the A / D converter 12, performs quadrature demodulation processing, converts it into an equivalent baseband signal, and outputs it. The subtractor 14 receives the equivalent baseband signal from the quadrature demodulator 13, synthesizes the wraparound replica signal from the FBF 16 in reverse phase, and outputs it to the FFF 15. The FFF 15 equalizes the multipath component and outputs it as an equivalent baseband signal that has been subjected to cancellation of wraparound and multipath equalization. The quadrature modulation unit 17 receives the equivalent baseband signal, performs quadrature modulation processing, converts it into a digital IF signal, and outputs it. The D / A converter 18 receives the digital IF signal from the quadrature modulator 17, converts it into a second IF signal, and outputs it. The frequency conversion unit 19 receives the second IF signal from the D / A conversion unit 18, converts it to an IF signal, and outputs it.

  The transmission conversion unit 105 receives the IF signal from the frequency conversion unit 19 of the wraparound cancellers 1 and 2, converts the frequency to the RF band, amplifies it to a certain level, and outputs it. The PA 106 receives an RF band signal from the transmission conversion unit 105, and amplifies the power to output a transmission signal having a desired output. The transmission filter 107 receives an RF band transmission signal from the PA 106 and removes unnecessary radiation components outside the band. The transmission signal from which unnecessary components outside the band are removed by the transmission filter 107 is supplied to the transmission antenna 108 via the feeder cable and radiated as radio waves. As described above, according to the relay apparatus 101 shown in FIG. 12, by using the wraparound cancellers 1 and 2, it is possible to relay the upper station wave favorably and stably.

1, 2 wraparound cancellers 11, 19 Frequency converter 12 A / D converter 13 Orthogonal demodulator 14 Subtractor 15 FFF
16 FBF
17 Quadrature modulation unit 18 D / A conversion unit 20, 42 Filter coefficient control unit 21 Effective symbol period extraction unit 22, 65 FFT unit 23, 43 Frequency characteristic calculation unit 24, 57, 60 Phase compensation unit 25, 44 Channel estimation unit 26 LPF unit 27, 47, 52, 54, 56 Division unit 28 FFT window position correction unit 29 Cancel residual calculation unit 30, 35, 63 IFFT unit 31, 36, 40, 61 Multiplication unit 32, 37 Addition unit 33, 38 Delay Unit 34 equalization error calculation unit 39 convolution operation unit 41 digital SAW filter coefficient storage unit 45, 50 SP signal extraction unit 46, 51 SP signal generation unit 48, 53 interpolation unit 49, 59 channel equalization unit 55, 62 determination unit 58 Compensated SP signal generation unit 64 Window function unit 101 Relay device 102 Reception antenna 103 Reception filter 104 Reception conversion unit 105 Transmission Conversion unit 106 PA
107 transmit filter 108 transmit antenna

Claims (6)

An FBF (Feed Back Filter) that receives a broadcast wave OFDM signal that has been transmitted in a concatenated manner with phase rotation correction performed by the transmission unit on a segment basis, and generates a wraparound replica signal included in the received OFDM signal; An FFF (Feed Forward Filter) that equalizes distortion due to multipath of a signal after canceling a wraparound replica signal from the OFDM signal, and a filter coefficient control unit that generates a filter coefficient for controlling the FBF and FFF In the wraparound canceller provided,
The filter coefficient control unit
An effective symbol period extraction unit that extracts an OFDM signal of an effective symbol period from the OFDM signal equalized by the FFF;
An FFT unit that performs FFT (Fast Fourier Transform) on the OFDM signal extracted by the effective symbol period extraction unit and converts the signal into a carrier symbol;
A channel response calculation unit for calculating a channel response of the carrier symbol converted by the FFT unit;
An equalization error calculation unit that calculates an equalization error based on the channel response calculated by the channel response calculation unit;
An FFF coefficient calculation unit for calculating a filter coefficient used in the FFF from the equalization error calculated by the equalization error calculation unit;
A cancellation residual calculation unit that calculates a cancellation residual based on the channel response calculated by the channel response calculation unit;
From the cancel residual calculated by said canceling residual calculating unit, and a FBF coefficient calculating unit that calculates a filter coefficient used by the FBF,
The channel response calculation unit
A pilot signal generator for generating a pilot signal of known amplitude and phase;
A phase compensation unit that performs the same phase rotation correction on the pilot signal generated by the pilot signal generation unit as the transmission side and compensates for the phase of the pilot signal;
A division unit that divides a predetermined pilot signal of the carrier symbols converted by the FFT unit by a pilot signal whose phase is compensated by the phase compensation unit;
An interpolation unit for interpolating a division result by the division unit in a symbol direction and a carrier direction, and outputting the division result after interpolation as the channel response;
A wraparound canceller characterized by comprising: In the wraparound canceller according to claim 1 ,
The channel response calculation unit includes a compensated pilot signal generation unit instead of the pilot signal generation unit and the phase compensation unit,
The compensated pilot signal generator is
A pilot signal having a known amplitude and having a known phase subjected to the same phase rotation correction as that of the transmitting side;
The division unit is
A wraparound canceller that divides a predetermined pilot signal of the carrier symbols converted by the FFT unit by a pilot signal generated by the compensated pilot signal generation unit. In the wraparound canceller according to claim 1,
A new channel response calculation unit replacing the channel response calculation unit ,
A pilot signal generator for generating a pilot signal of known amplitude and phase;
A first phase compensation unit that performs the same phase rotation correction on the pilot signal generated by the pilot signal generation unit as the transmission side and compensates the phase of the pilot signal;
A first division unit for dividing a predetermined pilot signal of the carrier symbols converted by the FFT unit by a pilot signal whose phase is compensated by the first phase compensation unit;
An interpolation unit for interpolating a division result by the first division unit in a symbol direction and a carrier direction, and outputting the interpolated channel response;
A second division unit for dividing the carrier symbol converted by the FFT unit by the channel response interpolated by the interpolation unit to equalize the channel;
A determination unit that determines a known transmission symbol having the smallest Euclidean distance in the signal space from the carrier symbol channel-equalized by the second division unit as an estimated value of the transmission symbol;
A second phase compensation unit that performs the same phase rotation correction as that on the transmission side to the estimated value of the transmission symbol determined by the determination unit, and compensates the phase of the estimated value of the transmission symbol;
The transformed carrier symbols by the FFT unit, divided by the estimate of the transmitted symbol whose phase is compensated by the second phase compensating portion, a third divider for outputting the division result as the channel response,
A wraparound canceller characterized by comprising: In the wraparound canceller according to claim 3 ,
The new channel response calculation unit includes a new determination unit instead of the determination unit and the second phase compensation unit,
The new determination unit is
From the carrier symbol channel-equalized by the second division unit, an estimated value obtained by performing the same phase rotation correction as that of the transmission side with respect to a known transmission symbol having the smallest Euclidean distance in the signal space is obtained as a transmission symbol. As an estimate of
The third division unit is
The transformed carrier symbols by the FFT unit, the divided by the estimate of the transmitted symbol determined by the new diagnostic unit, and outputs the division result as the channel response, wraparound, wherein the canceller. In the wraparound canceller according to claim 4 ,
The new channel response calculation unit includes a compensated pilot signal generation unit instead of the pilot signal generation unit and the first phase compensation unit,
The compensated pilot signal generator is
A pilot signal having a known amplitude and having a known phase subjected to the same phase rotation correction as that of the transmitting side;
The first division unit is:
A wraparound canceller that divides a predetermined pilot signal of the carrier symbols converted by the FFT unit by a pilot signal generated by the compensated pilot signal generation unit. A relay apparatus using the wraparound canceller according to any one of claims 1 to 5 .

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