JP6033744B2 - Bridgeless power supply circuit - Google Patents

Description

  The present invention relates to a bridgeless power supply circuit that outputs an DC voltage from a smoothing capacitor by repeatedly storing an electromagnetic energy in an inductor and releasing an electromagnetic energy from the inductor by controlling an AC power supply as an input and controlling a switching element.

  Patent Document 1 introduces a full-bridge power supply circuit in FIG. 8 (conventional example treated in the same document). This first conventional example is shown here as FIG. 5 and will be described below. Reference numeral 12 denotes a bridge rectifier circuit that full-wave rectifies an alternating current from the alternating current power source 1 using four diodes D11, D12, D13, and D14. The diodes D11 and D14 are turned on during the positive half cycle period of the AC power supply 1, and the diodes D12 and D13 are turned on during the negative half cycle period. Electromagnetic energy is accumulated in the inductor L11 when the switching element Q11 is on, and electromagnetic energy is released from the inductor L11 when the switching element Q11 is off, and a counter electromotive voltage due to kickback is generated at the connection point between the inductor L11 and the switching element Q11. (Boosting), and the resulting current charges the smoothing capacitor C11 via the rectifier diode D15, and a DC voltage is taken out to the load RL. The conventional power supply circuit has room for simplification of configuration in that it has a full-bridge rectifier circuit. In addition, in both the positive half cycle period and the negative half cycle period, a current flows through two diodes at the same time, resulting in a large power loss.

  As a countermeasure for solving the above-described problems of the full-bridge rectifier circuit, Patent Document 1 introduces a bridgeless PFC (power factor improvement) power supply circuit in FIG. This second conventional example is shown here as FIG. 6 and will be described below. Here, there is no bridge rectifier circuit composed of four diodes. In the positive half cycle period of the AC voltage, the switching element Q22 is always turned off, and the switching element Q21 performs a switching operation. When the switching element Q21 is on, a current flows through the inductor L21 and the switching element Q21, and electromagnetic energy is accumulated in the inductor L21. When the switching element Q21 is turned off, electromagnetic energy is released from the inductor L1, and a back electromotive voltage due to kickback is generated at the connection point between the inductor L21 and the switching element Q21 (step-up), and the resulting current is smoothed via the rectifier diode D21. Capacitor C22 is charged, and a DC voltage is taken out to load RL. The return current on the terminal 1N side flows through the body diode of the switching element Q22 and the inductor L22. At this time, the inductor L22 functions as a simple current path component not involved in the step-up operation. In addition to the power loss in the inductor L22, a loss also occurs in the body diode. The number of diodes to be used is as few as D21 and D22, but the inductor L21 is paired with the diode D21 and the inductor L22 is paired with the diode D22, and two inductors are required.

  As a measure for avoiding the return current from flowing through the body diode and the inductor, Patent Document 1 introduces a bridgeless power supply circuit provided with a return diode in FIG. This third conventional example is shown here as FIG. 7 and will be described below. As a return current path from the smoothing capacitor C22, two diodes D23 and D24 to which the cathode K is connected to the AC power supply 1 are added (bypassing the body diode and the inductor). When the switching element Q21 is turned off and the smoothing capacitor C22 is charged in the positive half cycle period of the AC voltage, the charging current is returned via the diode D24, and the body diode of the switching element Q22 and No return current flows through the inductor L22. Although the power loss in the inductor L22 and the body diode in the case of FIG. 6 is eliminated, a large power loss occurs because the current flows through the two diodes of the diode D21 and the return diode D24 simultaneously.

  The invention described in Patent Document 1 has been proposed with the aim of high efficiency and low loss against the background described above. Here, the invention of Patent Document 1 is shown in FIG. 8 as a fourth conventional example and will be described below.

  The first and second input terminals 1P and 1N to which the AC power source 1 is connected and the first and second circuit lines lp and ln are derived, and the first and second output terminals 2P from which a DC voltage is output. , 2N and a smoothing capacitor C1 connected between the first and second output terminals 2P, 2N. Further, the inductor L1, the first switching element Q1 connected between the first circuit line lp and one end of the inductor L1, and the second circuit line ln and the other end of the inductor L1 are connected. And a second switching element Q2. In addition, an anode is connected to one end of the inductor L1, a cathode is connected to the first output terminal 2P, an anode is connected to the other end of the inductor L1, and a cathode is connected to the first output terminal 2P. Is connected to the second diode D2, the cathode connected to the first circuit line lp, the third diode D3 having the anode connected to the second output terminal 2N, and the cathode connected to the second circuit line ln. And a fourth diode D4 having an anode connected to the second output terminal 2N. Further, during one half cycle period of the AC power supply voltage, the first switching element Q1 is always turned on, and the second switching element Q2 is switched to operate during the other half cycle period of the AC power supply voltage. The second switching element Q2 is always turned on, and the control circuits 3A and 3B are provided for switching the first switching element Q1.

  In the positive half cycle period in which the first switching element Q1 is always turned on, the first input terminal 1P → the first switching element Q1 → the inductor in the period in which the second switching element Q2 performing the switching operation is on. A current flows through a path of L1 → second switching element Q2 → second input terminal 1N, and electromagnetic energy is accumulated in the inductor L1. Next, when the second switching element Q2 is turned off, the electromagnetic energy accumulated in the inductor L1 is released, and the smoothing capacitor C1 is charged via the diode D2. The return current at this time flows to the second input terminal 1N via the diode D4.

  On the other hand, in the negative half cycle period in which the second switching element Q2 is always turned on, the second input terminal 1N → the second switching element Q2 in the period in which the first switching element Q1 performing the switching operation is on. The current flows through the path of the inductor L1, the first switching element Q1, and the first input terminal 1P, and electromagnetic energy is accumulated in the inductor L1. Next, when the first switching element Q1 is turned off, the electromagnetic energy accumulated in the inductor L1 is released, and the smoothing capacitor C1 is charged via the diode D1. The return current at this time flows to the first input terminal 1P via the diode D3.

  In the bridgeless power supply circuit of the fourth conventional example, neither the inductor L1 nor the body diode is a return path. The number of inductors used is only one, and since there are only two diodes (D2 and D4 or D1 and D3) that conduct simultaneously in each half cycle period, the number of parts is smaller than the prior art, The power loss will be less.

JP 2010-93989 A

  In the bridgeless power supply circuit of the fourth conventional example (FIG. 8), the number of diodes used is four, so there is a room for improvement in terms of reducing the number of components. In addition, since there are two diodes that conduct at the same time in each half cycle period, there is room for improvement in terms of reducing power loss.

  The present invention has been created in view of such circumstances, and an object of the present invention is to reduce the power loss and simplify the configuration by reducing the number of parts in the bridgeless power supply circuit.

  The present invention solves the above problems by taking the following measures.

The bridgeless power supply circuit according to the present invention includes:
AC power input terminal,
DC voltage output terminal,
A smoothing capacitor connected between the output terminals;
A first switching element and a second switching element in which one terminal of each current path is connected to each other, and the connection point is connected to one terminal of the smoothing capacitor;
A first unidirectional element connected between the other terminal of the smoothing capacitor and the other terminal of the current path of the first switching element, the current direction from the former to the latter;
A second unidirectional element connected between the other terminal of the smoothing capacitor and the other terminal of the current path of the second switching element, the current direction from the former to the latter;
An inductor inserted between one of the other terminal of the first switching element or the other terminal of the second switching element and the input terminal;
During the first half cycle period of the AC power supply voltage, the second switching element is always turned on and the first switching element is switched, and the second switching element is operated during the second half cycle period of the AC power supply voltage. And a control circuit for always switching on the first switching element and switching the second switching element.

  In the first half cycle period in which the second switching element is always turned on, in the on state of the first switching element, an alternating current is connected to the series circuit of the first and second switching elements and the inductor that are both in the on state. Current flows and electromagnetic energy is stored in the inductor. In this state, a short circuit of the bypass path (return path) with respect to the series circuit of the second and first switching elements is prevented by the rectifying action of the second unidirectional element. In the OFF state of the first switching element, the current accompanying the release of electromagnetic energy from the inductor flows through the path of the second switching element, the smoothing capacitor, and the first unidirectional element in the ON state, and charging the smoothing capacitor A DC voltage is output from the output terminals at both ends of the smoothing capacitor. At this time, the first unidirectional element is in a conductive state and forms a return path.

  On the other hand, in the second half cycle period in which the first switching element is always turned on, the series circuit of the first and second switching elements and the inductor that are both in the on state when the second switching element is on. AC current flows through the inductor, and electromagnetic energy is accumulated in the inductor. In this state, a short circuit of the bypass path (return path) with respect to the series circuit of the first and second switching elements is prevented by the rectifying action of the first unidirectional element. In the OFF state of the second switching element, the current accompanying the release of electromagnetic energy from the inductor flows through the path of the first switching element, the smoothing capacitor, and the second unidirectional element that are in the ON state. The capacitor is charged and a DC voltage is output from the output terminals at both ends of the smoothing capacitor. At this time, the second unidirectional element is in a conductive state and forms a return path.

  As described above, the backflow prevention action of the second unidirectional element enables the switching boosting operation by the first switching element in the first half cycle period, and the first switching element is in the off state. The unidirectional element conducts and secures a return current path from the smoothing capacitor.

  On the other hand, the backflow prevention action of the first unidirectional element enables the switching step-up operation by the second switching element in the second half cycle period, and the second unidirectional element in the off state of the second switching element. The element conducts and secures a return current path from the smoothing capacitor.

  As described above, both the first unidirectional element and the second unidirectional element have both functions of preventing backflow and current return. Since it has both functions of preventing backflow and current return, the number of constituent elements can be reduced.

  In the above-described configuration of the present invention, there is only one unidirectional element that is in a conductive state in the charging path for the smoothing capacitor in one half cycle period and consumes power, and the second conventional example (FIG. 6) or The inductor in the return path as in the third conventional example (FIG. 7) does not exist, and the body diode of the switching element as in the second conventional example is not used for the return path. There is no additional diode for bypassing, and no two diodes are inserted in the charging path for the smoothing capacitor as in the fourth conventional example (FIG. 8).

  The unidirectional element in the present invention may be a thyristor other than a diode, or a diode-connected transistor. In the case of a bipolar transistor, the one in which the collector and the base are short-circuited is a unidirectional element, and in the case of the MOSFET, the one in which the drain and the gate are short-circuited is a unidirectional element.

  According to the present invention, since the unidirectional element is used for both the rectification (backflow prevention) function and the return path for the bridgeless power supply circuit, the power loss is reduced and the number of components is reduced as compared with the conventional example. Simplification of the configuration can be realized.

The circuit diagram which shows the structure of the bridgeless power supply circuit in embodiment of this invention Current state explanatory diagram in the positive half cycle period of the bridgeless power supply circuit of the embodiment of the present invention Current state explanatory diagram in the negative half cycle period of the bridgeless power supply circuit of the embodiment of the present invention Timing chart showing the operation of the bridgeless power supply circuit of the embodiment of the present invention 1 is a circuit diagram showing a configuration of a bridgeless power supply circuit according to a first conventional example. The circuit diagram which shows the structure of the bridgeless power supply circuit of the 2nd prior art example Circuit diagram showing configuration of third conventional bridgeless power supply circuit 4 is a circuit diagram showing a configuration of a bridgeless power supply circuit of a fourth conventional example.

  Embodiments of a bridgeless power supply circuit according to the present invention will be described below in detail with reference to the drawings.

  1 is a circuit diagram showing a configuration of a bridgeless power supply circuit according to an embodiment of the present invention, FIG. 2 is an explanatory diagram of a current state in a positive half cycle period of the bridgeless power supply circuit of the embodiment, and FIG. 3 is a negative half cycle period. FIG. 4 is a timing chart showing the operation of the bridgeless power supply circuit of the embodiment.

  First, the components are listed. In FIG. 1, A is a bridgeless power supply circuit, T1p and T1n are first and second input terminals of an AC power supply, T2p and T2n are first and second output terminals of a DC voltage, L51 is an inductor, and Q51 is a first output terminal. 1 switching element (lower side), Q52 is a second switching element (upper side), C51 is a smoothing capacitor made of an electrolytic capacitor, D51 is a first diode (lower side) as a first unidirectional element, D52 Is a second diode (upper side) as a second unidirectional element, CT is a current transformer for detecting overcurrent, 50 is an AC power supply connected to the first and second input terminals T1p and T1n, 51 Is a control circuit, and R51 is a load resistor connected between the first and second output terminals T2p, T2n.

  One terminal of the inductor L51 is connected to the first input terminal T1p, a series circuit of the second switching element Q52 and the first switching element Q51 is connected to the other terminal of the inductor L51, and the other terminal of the series circuit is the first terminal. 2 input terminals T1n. A connection point between one terminal of the current path of the first switching element Q51 and one terminal of the current path of the second switching element Q52 is connected to one terminal (positive electrode terminal) of the smoothing capacitor C51 and the first output terminal T2p. ing. The other terminal (negative electrode terminal) of the smoothing capacitor C51 is connected to the second output terminal T2n, and the connection point P1 is connected to the second input terminal T1n via the first diode D51 which is the first unidirectional element. The inductor L51 and the second switching element Q52 are connected to a connection point P2 with the other terminal of the current path of the first switching element Q51 and through the second diode D52 which is the second unidirectional element. It is connected to a connection point P3 with the other terminal of the current path. rt1 is a first return path where the charging current for the smoothing capacitor C51 returns from the connection point P1 to the connection point P2, and rt2 is a second return path where the charging current returns from the connection point P1 to the connection point P3.

  As the first and second switching elements Q51 and Q52, here, N-channel MOSFETs (field effect transistors made of a metal oxide semiconductor) are used. The source S of the second switching element Q52 which is an NMOS is connected to a connection point P3 between the inductor L51 and the cathode K of the second diode D52, and the first switching where the drain D of the second switching element Q52 is an NMOS. The drain S of the element Q51 is connected, and the source S of the first switching element Q51 is connected to a connection point P2 between the other second input terminal T1n and the cathode K of the first diode D51. The anodes A of the first and second diodes D51 and D52 are both connected to the other terminal (negative electrode terminal) P1 of the smoothing capacitor C51. The first diode D51 is inserted in the first return path rt1, and the second diode D52 is inserted in the second return path rt2.

  The control circuit 51 is driven by receiving input power from the first and second input terminals T1p, T1n, and controls on / off control and high-speed switching control of the first and second switching elements Q51, Q52. It is configured as. The control circuit 51 receives the detection current signal from the current transformer CT and the detection signal of the output voltage Vout from the positive terminal of the smoothing capacitor C51. The two drive signal output terminals of the control circuit 51 are connected to the gates G of the first switching element Q51 and the second switching element Q52, respectively. The control circuit 51 always turns on the second switching element Q52 during the positive half-cycle period of the power supply voltage by the AC power supply 50 and switches the first switching element Q51 at high speed so that the negative power supply voltage is negative. During the half cycle period, the first switching element Q51 is always turned on, and the second switching element Q52 is switched at high speed. Further, the output voltage Vout is feedback controlled by adjusting the duty ratios of the first and second switching elements Q51 and Q52 according to the detected level of the output voltage Vout. Further, when the current detected by the current transformer CT becomes an overcurrent, the drive control for the first and second switching elements Q51 and Q52 is stopped. The control of the first and second switching elements Q51 and Q52 by the control circuit 51 is executed by a gate drive unit (not shown) built in the control circuit 51.

  Next, the operation of the bridgeless power supply circuit A having the above configuration will be described below by dividing into (A) an operation in a positive half cycle period and (B) an operation in a negative half cycle period. 2A and 2B are current state explanatory diagrams in the positive half cycle period, and FIGS. 3A and 3B are current state explanatory diagrams in the negative half cycle period. The current flow is indicated by a dashed arrow. FIG. 4 is a timing chart showing the operation of the bridgeless power supply circuit.

(A) Operation in positive half cycle period (see FIG. 2)
In the positive half cycle period of the AC power supply voltage applied from the first and second input terminals T1p and T1n, the control circuit 51 always outputs a drive signal for the gate G of the second switching element Q52 of the NMOS-FET. On the other hand, the drive signal for the gate G of the first switching element Q51 of the NMOS-FET is switched to the “H” level / “L” level at high speed.

  First, as shown in FIG. 2A, during the period in which the first switching element Q51 is conducting, the current flowing from the first input terminal T1p is the inductor L51 → second switching element Q52 → first It flows along the path of the switching element Q51 and returns to the second input terminal T1n. During this period, electromagnetic energy is accumulated in the inductor L51.

  Next, as shown in FIG. 2B, when the first switching element Q51 becomes non-conductive, a kickback counter electromotive voltage based on the stored energy of the inductor L51 is generated at the connection point between the switching elements Q51 and Q52. (Step-up), and the resulting current charges the smoothing capacitor C51. The current charged in the smoothing capacitor C51 returns to the second input terminal T1n through the first diode D51.

  Thus, in the positive half cycle period, the first switching element Q51 is turned on in FIG. 2 (a) and the off state in FIG. 2 (b) while the second switching element Q52 is always on. The alternating switching with is repeated at high speed.

  In the positive half cycle period, as shown in the time range of T (+) in FIG. 4, the input voltage Vin has a mountain-shaped positive sine wave shape, and the second switching element Q52 is always kept on and fixed. The first switching element Q51 performs high-speed switching, and a sinusoidal current pulsating in a sawtooth shape on the positive side flows in the inductor L51 in proportion to the input voltage Vin. In the second switching element Q52, a current that changes in a sine wave shape on the negative side inversely proportional to the input voltage Vin flows. In the first switching element Q51, conduction / cutoff is switched at high speed, and the envelope is positive. A comb-like current having a sine wave shape flows.

  As a result, the smoothing capacitor C51 is switched by conduction / cutoff at high speed, and is charged by a comb-like current in which the envelope has a positive sine wave shape. However, the waveform of the charging current has a form in which the “H” period and the “L” period are inverted with respect to the current flowing through the first switching element Q51.

(B) Operation in negative half cycle period (see FIG. 3)
In the negative half cycle period of the AC power supply voltage applied from the first and second input terminals T1p and T1n, the control circuit 51 constantly outputs a drive signal for the gate G of the first switching element Q51 to “H”. On the other hand, the drive signal for the gate G of the second switching element Q52 is switched to the “H” level / “L” level at high speed.

  First, as shown in FIG. 3A, during the period in which the second switching element Q52 is conducting, the current flowing from the second input terminal T1n is changed from the first switching element Q51 to the second switching element Q52. → The current flows along the path of the inductor L51 and returns to the first input terminal T1p. During this period, electromagnetic energy is accumulated in the inductor L51.

  Next, as shown in FIG. 3B, when the second switching element Q52 becomes non-conductive, a kickback counter electromotive voltage based on the energy stored in the inductor L51 is generated at the connection point between the switching elements Q51 and Q52. (Step-up), and the resulting current charges the smoothing capacitor C51. The current charged in the smoothing capacitor C51 returns to the first input terminal T1p through the second diode D52 and the inductor L51.

  In this way, in the negative half cycle period, the second switching element Q52 is turned on in FIG. 3 (a) and the off state in FIG. 3 (b) while the first switching element Q51 is always on. The alternating switching with is repeated at high speed.

  In the negative half cycle period, as shown in the time range of T (−) in FIG. 4, the input voltage Vin has a valley-shaped negative sine wave shape, and the first switching element Q51 is always ON and fixed. Thus, the second switching element Q52 performs high-speed switching, and a sinusoidal current that pulsates in a sawtooth shape on the negative side in proportion to the input voltage Vin flows through the inductor L51. In the first switching element Q51, a current that changes in a sine wave shape on the negative side in proportion to the input voltage Vin flows. In the second switching element Q52, conduction / cutoff is switched at high speed, and the envelope is on the positive side. A comb-like current having a sine wave shape flows.

  As a result, the smoothing capacitor C51 is switched by conduction / cutoff at high speed, and is charged by a comb-like current in which the envelope has a positive sine wave shape. However, the waveform of the charging current has a form in which the “H” period and the “L” period are inverted with respect to the current flowing through the second switching element Q52.

  In FIG. 4, the current flowing through the second switching element Q52 changes in inverse proportion to the input voltage Vin. This is because the direction from the drain D to the source S in the second switching element Q52 is defined as the positive direction. By doing. The current flowing through the inductor L51 changes with pulsation due to the switching operation of the first switching element Q51 or the second switching element Q52 while being substantially proportional to the input voltage Vin.

  Next, functions of the first and second diodes D51 and D52 will be described.

  The second diode D52 is interposed in the second return path rt2 from the connection point P1, which is the negative terminal of the smoothing capacitor C51, to the connection point P3 in the state shown in FIG. 3B. If the second return path rt2 is not provided, the charging current cannot be supplied to the smoothing capacitor C51. If the second diode D52 is not interposed in the second return path rt2, the AC power source 50 passes through the first input terminal T1p and the inductor L51 to the connection point P3 in the state of FIG. The reached current flows into the second return path rt2. That is, current flows through the path of T1p → L51 → P3 → rt2 → P1 → rt1 → P2 → T1n, and does not flow in the series circuit of the switching elements Q52 and Q51, and the boosting operation by switching of the first switching element Q51 Falls into trouble.

  The first diode D51 is interposed in the first return path rt1 from the connection point P1, which is the negative terminal of the smoothing capacitor C51, to the connection point P2 in the state shown in FIG. 2B. If the first return path rt1 is not provided, the charging current cannot be supplied to the smoothing capacitor C51. If the first diode D51 is not interposed in the first return path rt1, the current that reaches the connection point P2 from the AC power supply 50 through the second input terminal T1n in the state of FIG. 3A. Flows into the first return path rt1. That is, current flows through the path of T1n → P2 → rt1 → P1 → rt2 → P3 → L51 → T1p, and does not flow in the series circuit of both switching elements Q51 and Q52, and the boosting operation by switching of the second switching element Q52 Falls into trouble.

  For this reason, the first diode D51 is inserted into the first return path rt1, and the second diode D52 is inserted into the second return path rt2. Both the first diode D51 and the second diode D52 have a rectification (backflow prevention) function and a return function.

  Next, power loss will be examined.

  As shown in FIG. 2B, only one of the first diodes D51 has a current flowing in the positive half cycle period. The power loss is reduced as compared with the case of returning through the two elements of the body diode of the second switching element Q22 and the second inductor L22 as in the second conventional example of FIG. Further, the power loss is reduced as compared with the case where the charging current is passed through the two elements of the first diode D21 and the fourth diode D24 as in the third conventional example of FIG. Further, the power loss is reduced as compared with the case where the charging current is passed through the two elements of the second diode D2 and the fourth diode D4 as in the fourth conventional example of FIG.

  Similarly, even in the negative half cycle period, the power loss is reduced as compared with each conventional example.

  Next, loss calculation will be described.

  For example, the AC input voltage is 200 Vrms, the AC input current is 10 Arms, the DC output voltage is 380 V, and the conversion efficiency is 95%. The suffix “rms” of Vrms and Arms is a root mean square and means an effective value. In this setting, the AC input power is 2 kW, the DC output power is 1.9 kW, and the maximum DC output current is 5 A. A typical diode voltage drop is 1.0V.

  In the case of the fourth conventional example of FIG. 8, which is considered to have the least power loss among the four conventional examples described above, two diodes in each of the positive half cycle period or the negative half cycle period of the commercial system voltage. Are simultaneously turned on, the generated power loss is 1.0 V × 5 A × 2 = 10 W (the current flowing through the diode is the same as the maximum DC output current).

  However, in the embodiment of the present invention, since current flows through only one diode in each of the positive half cycle period or the negative half cycle period of the commercial connection voltage, the generated power loss is 1.0 V × 5A = 5W, halved. Table 1 shows a comparison between the embodiment of the present invention and the fourth conventional example of FIG.

なお、上記の実施形態では第１および第２のスイッチング素子Ｑ５１，Ｑ５２としてＮチャネル型のＭＯＳＦＥＴを用いたが、本発明においてはこれのみに限定されるものではなく、Ｐチャネル型のＭＯＳＦＥＴを用いてもよく、あるいはバイポーラトランジスタ、ＩＧＢＴ（絶縁ゲートバイポーラトランジスタ）を用いるのでもよい（ＮＰＮ型、ＰＮＰ型のいずれも可）。

In the above embodiment, N-channel MOSFETs are used as the first and second switching elements Q51 and Q52. However, the present invention is not limited to this, and P-channel MOSFETs are used. Alternatively, a bipolar transistor or an IGBT (insulated gate bipolar transistor) may be used (both NPN type and PNP type).

  INDUSTRIAL APPLICABILITY The present invention is useful as a technique for reducing power loss due to a unidirectional element such as a diode and simplifying the configuration by reducing the number of components in a bridgeless power supply circuit.

C51 Smoothing capacitor D51 First unidirectional element (diode)
D52 Second unidirectional element (diode)
Q51 First switching element (transistor)
Q52 Second switching element (transistor)
T1p, T1n input terminal T2p, T2n output terminal 50 AC power supply 51 Control circuit

Claims (1)

AC power input terminal,
DC voltage output terminal,
A smoothing capacitor connected between the output terminals;
A first switching element and a second switching element in which one terminal of each current path is connected to each other, and the connection point is connected to one terminal of the smoothing capacitor;
A first unidirectional element connected between the other terminal of the smoothing capacitor and the other terminal of the current path of the first switching element, the current direction from the former to the latter;
A second unidirectional element connected between the other terminal of the smoothing capacitor and the other terminal of the current path of the second switching element, the current direction from the former to the latter;
An inductor inserted between one of the other terminal of the first switching element or the other terminal of the second switching element and the input terminal;
During the first half cycle period of the AC power supply voltage, the second switching element is always turned on and the first switching element is switched, and during the second half cycle period of the AC power supply voltage A bridgeless power supply circuit comprising: a control circuit that always turns on the first switching element and switches the second switching element.

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