JP6030922B2 - Light source control device - Google Patents

Description

  The present invention relates to a light source control device that controls a light source.

  2. Description of the Related Art In recent years, semiconductor light sources such as LEDs (Light Emitting Diodes) that have a longer life and lower power consumption have been used in vehicular lamps such as headlamps in place of conventional halogen lamps having filaments. Since the degree of light emission, that is, the brightness of the LED depends on the magnitude of the current flowing through the LED, a lighting circuit for adjusting the current flowing through the LED is required when the LED is used as a light source.

  The present applicant proposes a technique in Patent Document 1 that employs an array of LEDs as a light source and individually turns on and off each LED in order to make the light distribution of the headlamp variable and perform fine light distribution control. doing. In the lighting circuit described in Patent Document 1, a bypass switch is provided in parallel with each LED, and the individual lighting / extinguishing of the LED is realized by turning on and off the bypass switch.

JP2011-192865A

  When the bypass method as described in Patent Document 1 is adopted, the wiring around the LED becomes relatively complicated. When wiring becomes complicated, there is a possibility that the possibility of occurrence of poor conduction such as poor contact or disconnection may increase.

  The present invention has been made in view of such circumstances, and an object of the present invention is to provide a light source control device that can appropriately cope with the occurrence of poor conduction in the wiring around the light source and the bypass switch.

  One embodiment of the present invention relates to a light source control device. The light source control device includes a drive circuit that generates a drive current flowing through a plurality of semiconductor light sources connected in series, a first bypass switch connected in parallel with a part of the plurality of semiconductor light sources, and a first bypass. A second bypass switch connected in series with the switch and in parallel with another part of the plurality of semiconductor light sources. A connection wiring for connecting a connection node between the first bypass switch and the second bypass switch and a connection node between a part of the plurality of semiconductor light sources and another part of the plurality of semiconductor light sources. The polarity of the current flowing through the connection line when the first bypass switch is off and the second bypass switch is on is opposite to the polarity of the current flowing through the connection line when the first bypass switch is on and the second bypass switch is off. is there. When a continuity failure occurs in the connection wiring, both the first bypass switch and the second bypass switch are forcibly turned on.

  According to this aspect, when a continuity failure occurs in the connection wiring, both the first bypass switch and the second bypass switch can be forcibly turned on.

  Another embodiment of the present invention is also a light source control device. This device includes a drive circuit that generates a drive current flowing through a plurality of semiconductor light sources connected in series, a bypass switch connected in parallel with at least a part of the plurality of semiconductor light sources, and a bypass switch that is normally turned on. The main control circuit that periodically turns on and off, and the voltage across the bypass switch when the bypass switch is off are held in the capacitor when the voltage across the bypass switch is lower than the first voltage or higher than the second voltage that is higher than the first voltage When the amount of charge is changed in the first direction with the first time constant and the amount of charge held in the capacitor is not changed in the first direction with the first time constant, the amount of charge held in the capacitor is changed to the first time. An abnormality detection auxiliary circuit that changes in a second direction opposite to the first direction with a second time constant longer than the constant. The main control circuit determines whether or not an abnormality has occurred based on the voltage across the capacitor. If it is determined that an abnormality has occurred, the main control circuit forcibly turns on the bypass switch.

  According to this aspect, it is possible to determine whether or not an abnormality has occurred using the difference between the first time constant and the second time constant.

  It should be noted that any combination of the above-described constituent elements and those obtained by replacing the constituent elements and expressions of the present invention with each other among apparatuses, methods, systems, etc. are also effective as an aspect of the present invention.

  ADVANTAGE OF THE INVENTION According to this invention, the light source control apparatus which can respond appropriately even if a conduction defect generate | occur | produces in the wiring around a light source or a bypass switch can be provided.

It is a circuit diagram which shows the structure of the semiconductor light source control apparatus which concerns on embodiment, and the member connected to it. FIG. 2 is a circuit diagram showing a configuration of a hysteresis width setting circuit of FIG. 1. It is a graph which shows the relationship between the absolute value of a drive voltage, and an offset voltage. FIG. 2 is a circuit diagram showing a configuration of a down converter drive circuit in FIG. 1. FIG. 3 is a circuit diagram illustrating configurations of a second bypass circuit and a third bypass circuit in FIG. 1. FIGS. 6A to 6C are graphs showing the change over time of the drive current. It is a timing chart which shows the change of the 2nd bypass switch drive signal at the time of PWM dimming, and the 2nd abnormality detection signal. It is a circuit diagram which shows the structure of the semiconductor light source lighting circuit which concerns on a comparative example. FIGS. 9A to 9C are circuit diagrams showing configurations of semiconductor light source control devices according to the first, second, and third modifications. It is a circuit diagram which shows the structure of the semiconductor light source control apparatus which concerns on a 4th modification, and the member connected to it. It is a circuit diagram which shows the structure of the 2nd bypass circuit of the semiconductor light source control apparatus which concerns on a 5th modification. It is a timing chart which shows the change of the 2nd lighting extinction control signal when disconnection generate | occur | produces in 2nd LED.

  Hereinafter, the same or equivalent components, members, and signals shown in the respective drawings are denoted by the same reference numerals, and repeated description thereof will be omitted as appropriate. In addition, in the drawings, some of the members that are not important for explanation are omitted. Moreover, the code | symbol attached | subjected to the voltage, electric current, or resistance may be used as what represents each voltage value, electric current value, or resistance value as needed.

  In this specification, “the state in which the member A is connected to the member B” means that the member A and the member B are electrically connected in addition to the case where the member A and the member B are physically directly connected. It includes the case of being indirectly connected through another member that does not affect the connection state. Similarly, “the state in which the member C is provided between the member A and the member B” refers to the case where the member A and the member C or the member B and the member C are directly connected, as well as an electrical It includes the case of being indirectly connected through another member that does not affect the connection state.

  The semiconductor light source control device according to the embodiment generates a drive current that flows through a plurality of semiconductor light sources connected in series, that is, LEDs. Each LED is provided with a bypass switch in parallel. When the bypass switch is turned on (off), the corresponding LED is turned off (lit). A bypass connection wiring is provided between a connection node between two adjacent LEDs and a connection node between two corresponding bypass switches. The semiconductor light source control device is configured to forcibly turn on two bypass switches connected to the bypass connection wiring when a disconnection or poor contact occurs in the bypass connection wiring. Thereby, when such a conduction failure occurs, the related LED can be turned off.

  FIG. 1 is a circuit diagram showing a configuration of a semiconductor light source control device 100 according to an embodiment and members connected thereto. The semiconductor light source control device 100 supplies a drive current Iout to a plurality (N) of in-vehicle LEDs 2-1 to 2-N connected in series and lights them. N is a natural number of 2 or more. The semiconductor light source control device 100 and the N LEDs 2-1 to 2-N are mounted on a vehicle lamp such as a headlight. The semiconductor light source control device 100 is connected to the in-vehicle battery 6 and the power switch 8.

  The in-vehicle battery 6 generates a DC battery voltage (power supply voltage) Vbat of 12V (or 24V). The power switch 8 is a relay switch provided for controlling on / off of the entire N LEDs 2-1 to 2-N, and is provided in series with the in-vehicle battery 6. When the power switch 8 is turned on, the battery voltage Vbat is supplied from the positive terminal of the in-vehicle battery 6 to the semiconductor light source control device 100 as an input voltage. The negative terminal of the in-vehicle battery 6 is connected to the fixed voltage terminal, that is, grounded.

  Electrostatic protection Zener diodes 252-1 to 252-N are connected to the LEDs 2-1 to 2-N in parallel and in opposite directions. That is, the cathode of the first electrostatic protection Zener diode 252-1 is connected to the anode of the first LED 2-1, and the anode of the first electrostatic protection Zener diode 252-1 is connected to the cathode of the first LED 2-1. The same applies to the second electrostatic protection Zener diode 252-2 to the Nth electrostatic protection Zener diode 252-N. The electrostatic protection zener diode protects the corresponding LED from static breakdown.

  The semiconductor light source control device 100 includes a switching regulator or flyback regulator 102, a down converter 104, a control circuit 106, a current detection resistor 108, N bypass circuits 270-1 to 270 -N, and a bypass drive circuit 112. And comprising. Control circuit 106 controls flyback regulator 102 and downconverter 104, and includes flyback drive circuit 134, downconverter drive circuit 136, and hysteresis width setting circuit 138. The bypass drive circuit 112 is realized by a microcomputer.

  The flyback regulator 102 is a voltage regulator, and converts the input battery voltage Vbat into a target voltage Vt and outputs it. Since the output terminal on the high potential side of the flyback regulator 102 is grounded, the target voltage Vt is a voltage applied to the output terminal on the low potential side of the flyback regulator 102 and has a negative polarity. The flyback regulator 102 includes an input capacitor 114, a first switching element 116, an input transformer 124, an output diode 126, an output capacitor 128, a voltage detection diode 130, and a voltage detection capacitor 132.

  The input capacitor 114 is provided in parallel with the in-vehicle battery 6 and smoothes the battery voltage Vbat. More specifically, the input capacitor 114 is provided in the vicinity of the input transformer 124 and fulfills a voltage smoothing function for the switching operation of the flyback regulator 102.

  The primary winding 118 of the input transformer 124 and the first switching element 116 are connected in series, and the series circuit is connected in parallel with the input capacitor 114 to the in-vehicle battery 6. For example, the first switching element 116 is composed of an N-channel MOSFET (Metal Oxide Semiconductor Field Effect Transistor). One end of the secondary winding 120 of the input transformer 124 is connected to one end of the output capacitor 128, and the other end of the secondary winding 120 is connected to the anode of the output diode 126. The other end of the output capacitor 128 is connected to the cathode of the output diode 126. One end of the output capacitor 128 is connected to the output terminal on the low potential side of the flyback regulator 102, and the target voltage Vt is applied. The other end of the output capacitor 128 is connected to the output terminal on the high potential side of the flyback regulator 102.

  To the control terminal (gate) of the first switching element 116, a rectangular wave-form front-stage control signal S1 generated by the flyback drive circuit 134 is applied. The first switching element 116 is turned on when the previous-stage control signal S1 is asserted, that is, when it is at a high level, and turned off when it is negated, that is, when it is at a low level.

  The voltage detection winding 122, the voltage detection diode 130, and the voltage detection capacitor 132 of the input transformer 124 constitute a positive voltage detection circuit for detecting the magnitude of the target voltage Vt as a positive voltage. One end of the voltage detection winding 122 is grounded, and the other end is connected to the anode of the voltage detection diode 130. The cathode of voltage detection diode 130 is connected to one end of voltage detection capacitor 132. The other end of the voltage detection capacitor 132 is grounded. A positive voltage corresponding to the absolute value of the target voltage Vt is applied to one end of the voltage detection capacitor 132. This voltage is supplied to the flyback drive circuit 134 as the detection voltage Vd.

  The flyback drive circuit 134 performs voltage feedback control for keeping the target voltage Vt substantially constant based on the detection voltage Vd. The flyback drive circuit 134 adjusts the frequency and duty ratio of the pre-stage control signal S1 so that the target voltage Vt approaches a set voltage of, for example, about −100V.

  The down converter 104 is provided in the subsequent stage of the flyback regulator 102 and includes a second switching element 140, a flywheel diode 142, and an inductor 144, but does not include an output voltage smoothing capacitor.

  The second switching element 140 is composed of, for example, an N channel MOSFET. The control terminal of the second switching element 140 is applied with a rectangular-wave subsequent control signal S2 generated by the down converter drive circuit 136. The second switching element 140 is turned on when the post-stage control signal S2 is at a high level, and turned off when it is at a low level. The drain of the second switching element 140 is connected to the output terminal on the high potential side of the output capacitor 128, that is, the high potential side of the flyback regulator 102. The source of the second switching element 140 is connected to the cathode of the flywheel diode 142.

  The anode of the flywheel diode 142 is connected to one end of the inductor 144. A connection node between the anode of the flywheel diode 142 and one end of the inductor 144 is connected to the low potential side of the output capacitor 128, that is, the low potential side output terminal of the flyback regulator 102. The other end of the inductor 144 is connected to the cathode side of the N LEDs 2-1 to 2-N.

  The current detection resistor 108 is provided on the path of the drive current Iout. One end of the current detection resistor 108 is connected to a connection node between the source of the second switching element 140 and the cathode of the flywheel diode 142. The other end of the current detection resistor 108 is grounded and connected to the anode side of the N LEDs 2-1 to 2-N. A voltage drop Vm proportional to the drive current Iout is generated in the current detection resistor 108.

  Since the anode sides of the N LEDs 2-1 to 2-N are grounded, the negative drive voltage Vout is applied to the cathode side of the N LEDs 2-1 to 2-N, that is, the other end of the inductor 144. . During normal lighting, the drive voltage Vout is a negative voltage having a magnitude corresponding to the number of LEDs in the light emitting state (= the corresponding bypass switch is turned off) × the forward drop voltage Vf of one LED. .

  The down converter drive circuit 136 performs current feedback control for keeping the drive current Iout within a predetermined current range based on the voltage drop Vm. The down-converter drive circuit 136 turns off the second switching element 140 when the magnitude of the drive current Iout exceeds a predetermined current upper limit value Ith1, and the current lower limit value Ith2 whose magnitude of the drive current Iout is smaller than the current upper limit value Ith1. If it falls below, the second switching element 140 is turned on. The down converter drive circuit 136 sets the post-stage control signal S2 to a low level when the magnitude of the drive current Iout exceeds the current upper limit value Ith1, and sets the post-stage control signal S2 to high when the magnitude of the drive current Iout falls below the current lower limit value Ith2. Level.

  The hysteresis width setting circuit 138 sets a hysteresis width ΔI that is the difference between the current upper limit value Ith1 and the current lower limit value Ith2 based on the drive voltage Vout. The hysteresis width setting circuit 138 increases the hysteresis width ΔI as the absolute value of the drive voltage Vout increases when the absolute value of the drive voltage Vout falls below the voltage threshold Vth that is smaller than the absolute value of the target voltage Vt. When the absolute value of the voltage Vout exceeds the voltage threshold value Vth, the hysteresis width ΔI is decreased as the absolute value of the drive voltage Vout increases.

  FIG. 2 is a circuit diagram showing a configuration of the hysteresis width setting circuit 138. The hysteresis width setting circuit 138 includes a first operational amplifier 146, a first diode 148, a first resistor 150, a second resistor 152, a third resistor 154, a fourth resistor 156, a fifth resistor 158, A reference voltage source 160. A positive control power supply voltage Vcc is applied to one end of the third resistor 154. The other end of the third resistor 154 is connected to one end of the second resistor 152, one end of the fifth resistor 158, and one end of the fourth resistor 156. The other end of the fourth resistor 156 is grounded. The drive voltage Vout is applied to the other end of the fifth resistor 158. The other end of the second resistor 152 is connected to the inverting input terminal of the first operational amplifier 146. The inverting input terminal of the first operational amplifier 146 is connected to the anode of the first diode 148 through the first resistor 150. The cathode of the first diode 148 is connected to the output terminal of the first operational amplifier 146. A reference voltage Vref generated by the reference voltage source 160 is applied to the non-inverting input terminal of the first operational amplifier 146. A voltage applied to the anode of the first diode 148 is referred to as an offset voltage Voffset. As will be described later, the offset voltage Voffset corresponds to the hysteresis width ΔI, and the hysteresis width ΔI increases as the offset voltage Voffset increases.

  The resistance value around the first operational amplifier 146 is compared with the values of the third resistor 154, the fourth resistor 156, and the fifth resistor 158 that are differential with the reference voltage Vref, and the first resistor 150 that determines the amplification factor. The value of the second resistor 152 is sufficiently increased so that the feedback current does not affect the differential with the reference voltage Vref.

  FIG. 3 is a graph showing the relationship between the absolute value of the drive voltage Vout and the offset voltage Voffset. When the absolute value of the negative drive voltage Vout is small, the voltage at the common connection node of the third resistor 154, the fourth resistor 156, and the fifth resistor 158 is larger than the reference voltage Vref, and thus the first operational amplifier 146 is The offset voltage Voffset is reduced by sinking the current. The offset voltage Voffset becomes maximum when the voltage of the common connection node becomes equal to the reference voltage Vref.

  In order to realize the control that the hysteresis width ΔI, that is, the offset voltage Voffset is maximized when the absolute value of the drive voltage Vout becomes the voltage threshold Vth, the absolute value of the drive voltage Vout is set to the reference voltage Vref. It is set to the voltage of the common connection node when it is equal to the threshold value Vth. In particular, when the set voltage of the flyback regulator 102 is −100V, the reference voltage Vref is set to the voltage of the common connection node when the drive voltage Vout = −Vth = −50V.

  When the absolute value of the drive voltage Vout increases beyond the voltage threshold value Vth, the operation of the first operational amplifier 146 is lost, and the voltage of the common connection node becomes the offset voltage Voffset as it is. The hysteresis width setting circuit 138 controls the hysteresis width ΔI by sending the offset voltage Voffset that changes in a mountain shape with respect to the absolute value of the drive voltage Vout to the down converter drive circuit 136 in this way, and the switching frequency of the down converter 104 Is within a predetermined range.

  FIG. 4 is a circuit diagram showing a configuration of down converter drive circuit 136. The down converter drive circuit 136 includes a second operational amplifier 162, a comparator 164, a gate driver 166, a first current mirror circuit 170, a seventh resistor 172, an eighth resistor 174, a tenth resistor 178, A twelfth resistor 182, a thirteenth resistor 184, a first npn bipolar transistor 190, a third switching element 202, a fourth switching element 204, and a second current mirror circuit 206 are included.

  The offset voltage Voffset is applied to the non-inverting input terminal of the second operational amplifier 162. The output terminal of the second operational amplifier 162 is connected to the base of the first npn type bipolar transistor 190, and the inverting input terminal is connected to the emitter of the first npn type bipolar transistor 190. One end of the eighth resistor 174 is connected to the emitter of the first npn-type bipolar transistor 190, and the other end is grounded. The collector of the first npn-type bipolar transistor 190 is connected to the first current mirror circuit 170 via the seventh resistor 172.

  The first current mirror circuit 170 includes a sixth resistor 168, a ninth resistor 176, an eleventh resistor 180, a first pnp bipolar transistor 192, a second pnp bipolar transistor 194, a third pnp bipolar transistor 196, Have These circuit elements are connected to each other so as to form a known current mirror circuit. The first current mirror circuit 170 receives the current flowing through the seventh resistor 172, receives the current flowing through the tenth resistor 178, and the current flowing through the third switching element 202 as an output, and the magnitude of the input current and the magnitude of the output current are substantially reduced. Make equal.

  The second current mirror circuit 206 includes a fourteenth resistor 186, a fifteenth resistor 188, a second npn-type bipolar transistor 198, and a third npn-type bipolar transistor 200. These circuit elements are connected to each other so as to form a known current mirror circuit. The second current mirror circuit 206 receives the current flowing through the tenth resistor 178 as an input and the current flowing through the fourth switching element 204 as an output, and makes the magnitude of the input current and the magnitude of the output current substantially equal.

  The third switching element 202 is composed of, for example, a P-channel MOSFET. The fourth switching element 204 is composed of, for example, an N-channel MOSFET. The source of the third switching element 202 is connected to the first current mirror circuit 170. The gate of the third switching element 202 is connected to the inverting output terminal of the comparator 164. The drain of the third switching element 202 is connected to the drain of the fourth switching element 204. The gate of the fourth switching element 204 is connected to the inverting output terminal of the comparator 164. The source of the fourth switching element 204 is connected to the second current mirror circuit 206.

  The twelfth resistor 182 and the thirteenth resistor 184 are connected in series between the control power supply voltage Vcc and the ground potential in this order. A connection node between the twelfth resistor 182 and the thirteenth resistor 184 is connected to a connection node between the drain of the third switching element 202 and the drain of the fourth switching element 204. A connection node between the drain of the third switching element 202 and the drain of the fourth switching element 204 is connected to the non-inverting input terminal of the comparator 164. A voltage drop Vm is applied to the inverting input terminal of the comparator 164.

  The non-inverting output terminal of the comparator 164 is connected to the gate driver 166. The gate driver 166 aligns the phase of the post-stage control signal S2 with the phase of the signal appearing at the non-inverting output terminal of the comparator 164. That is, when the signal appearing at the non-inverting output terminal of the comparator 164 becomes a high level (low level), the gate driver 166 sets the subsequent stage control signal S2 to a high level (low level).

  The second operational amplifier 162 and the first npn-type bipolar transistor 190 that receive the offset voltage Voffset output a current of Voffset / (resistance value of the eighth resistor 174). This current is sinked or sourced to the voltage dividing node of the twelfth resistor 182 and the thirteenth resistor 184 depending on the phase of the output of the comparator 164 that receives the voltage drop Vm. Regarding the third switching element 202 and the fourth switching element 204 in the bridge form, the third switching element 202 is turned on at the timing when the gate of the second switching element 140 is at the high level (the second switching element 140 is turned on), and the twelfth resistance The voltage at the voltage dividing node between 182 and the thirteenth resistor 184 increases, and the current upper limit value Ith1 is set. When the drive current Iout increases and reaches the current upper limit value Ith1, the fourth switching element 204 is turned on substantially simultaneously with the gate of the second switching element 140 becoming low level (the second switching element 140 is turned off). . As a result, the voltage at the voltage dividing node between the twelfth resistor 182 and the thirteenth resistor 184 decreases, and the current lower limit value Ith2 is set.

  The average value of the drive current Iout is set by the divided voltage of the twelfth resistor 182 and the thirteenth resistor 184. Further, due to the action of the hysteresis width setting circuit 138, the sink / source current increases when the absolute value of the drive voltage Vout approaches the voltage threshold value Vth. Therefore, the current upper limit value Ith1−current lower limit value Ith2 = the hysteresis width ΔI is large. Become. The hysteresis width ΔI decreases as the absolute value of the drive voltage Vout increases from the voltage threshold Vth. As will be described later, this acts to keep the switching frequency of the down converter 104 within a predetermined range.

  Returning to FIG. 1, the semiconductor light source control device 100 is configured to be able to individually control the turning on / off of the N LEDs 2-1 to 2-N. The bypass drive circuit 112 generates N lighting on / off control signals Sc1 to ScN for controlling lighting / lighting off of the LEDs 2-1 to 2-N. The bypass drive circuit 112 individually controls the levels of the lighting control signals Sc1 to ScN so that a desired luminance and light distribution pattern can be obtained. Specifically, the bypass drive circuit 112 sets the first turn-on / off control signal Sc1 to a low level when turning on the first LED 2-1, and sets the first turn-on / off control signal Sc1 to high when turning off the first LED 2-1. Level. The same applies to the second lighting on / off control signal Sc2 to the Nth lighting on / off control signal ScN. The bypass drive circuit 112 outputs the lighting control signals Sc1 to ScN to the corresponding bypass circuits 270-1 to 270-N.

  The first bypass circuit 270-1 to the Nth bypass circuit 270-N are connected in parallel with the first LED2-1 to the Nth LED2-N, respectively. The first bypass circuit 270-1 to the Nth bypass circuit 270 -N are connected in series between the high potential side output terminal and the low potential side output terminal of the down converter 104 in this order.

  When the first lighting on / off control signal Sc1 is at a high level, the first bypass circuit 270-1 conducts between both ends of the first LED 2-1, that is, connects with a lower impedance than the first LED 2-1. Thereby, the first LED 2-1 is turned off. Hereinafter, the state of the bypass circuit that turns off the LED in this way is referred to as a bypass-on state. The first bypass circuit 270-1 connects both ends of the first LED 2-1 with a higher impedance than the first LED 2-1 when the first lighting control signal Sc <b> 1 is at a low level. As a result, the first LED 2-1 is turned on. Hereinafter, the state of the bypass circuit that turns on the LED in this way is referred to as a bypass-off state.

  The first bypass circuit 270-1 generates a first abnormality detection signal Sdet 1 for detecting an abnormality in the first LED 2-1 and its surrounding wiring, and supplies the first abnormality detection signal Sdet 1 to the bypass drive circuit 112. When the first abnormality detection signal Sdet1 is at a high level, the voltage applied to the first bypass circuit 270-1 when the first bypass circuit 270-1 is in the bypass-off state is lower than the short detection voltage or shorted The disconnection detection voltage is higher than the detection voltage. The short detection voltage is set to be lower than the forward drop voltage Vf of one LED. The disconnection detection voltage is set to be higher than the forward drop voltage Vf of the LED and lower than the sum 2Vf of the forward drop voltages of the two LEDs.

  Similarly, the second bypass circuit 270-2 to the Nth bypass circuit 270-N also turn on / off the second LED2-2 to the NLED2-N based on the second lighting control signal Sc2 to the Nth lighting control signal ScN, respectively. To control. The second bypass circuit 270-2 to the Nth bypass circuit 270-N generate the second abnormality detection signal Sdet2 to the Nth abnormality detection signal SdetN, respectively, and supply them to the bypass drive circuit 112.

  When the brightness of the first LED 2-1 is reduced during normal lighting, the bypass drive circuit 112 periodically changes the first lighting control signal Sc1 at a dimming frequency f1 of several hundred Hz to several kHz, that is, in a rectangular wave shape. By such pulse modulation of the first lighting on / off control signal Sc1, the first LED 2-1 blinks at the dimming frequency f1, and the brightness perceived by human eyes is reduced. The duty ratio of the first lighting on / off control signal Sc1 is set so as to obtain a desired degree of light emission. In this case, since the fluctuation of the magnitude of the drive current flowing through the first LED 2-1 when the first LED 2-1 is turned on can be suppressed, the color shift can be suppressed. The bypass drive circuit 112 similarly has a PWM (Pulse Width Modulation) dimming function for each of the second LED 2-2 to the Nth LED 2-N.

  The bypass drive circuit 112 determines whether or not an abnormality has occurred in the first LED 2-1 or the surrounding wiring based on the first abnormality detection signal Sdet1. The bypass drive circuit 112 determines that an abnormality has occurred when the first abnormality detection signal Sdet1 becomes high level, and forcibly sets the first lighting on / off control signal Sc1 to high level. For example, according to the PWM dimming function, the bypass drive circuit 112 keeps the first lighting control signal Sc1 at the high level even during the period when the first lighting control signal Sc1 should be at the low level. Alternatively, if it is not determined that an abnormality has occurred, the bypass drive circuit 112 keeps the first turn-on / off control signal Sc1 at a high level even during the period when the first turn-on / off control signal Sc1 should be at a low level. The bypass drive circuit 112 has a similar abnormality detection function for each of the second LED 2-2 to the N-th LED 2-N.

The first bypass connection wiring 280-1 includes the first bypass side connection node NB1 of the first bypass circuit 270-1 and the second bypass circuit 270-2, and the first of the first LED 2-1 and the second LED 2-2. The load side connection node NL1 is connected. When the first bypass circuit 270-1 is in the bypass-off state and the second bypass circuit 270-2 is in the bypass-on state, the polarity of the drive current Iout flowing through the first bypass connection wiring 280-1 is from the load side to the bypass side. The direction is heading. When the first bypass circuit 270-1 is in the bypass-on state and the second bypass circuit 270-2 is in the bypass-off state, the polarity of the drive current Iout flowing through the first bypass connection wiring 280-1 is from the bypass side to the load side. The direction is heading. Therefore, the polarity of the drive current Iout flowing through the first bypass connection wiring 280-1 in the former case is opposite to the polarity of the drive current Iout flowing through the first bypass connection wiring 280-1 in the latter case.
The same applies to the second bypass connection wiring 280-2 to the (N-1) th bypass connection wiring 280- (N-1).

  The bypass drive circuit 112 and the first bypass circuit 270-1 to the Nth bypass circuit 270 -N are configured such that, when a conduction failure occurs in the bypass connection wiring, both of the two bypass circuits connected to the bypass connection wiring are It is configured to be forced into a bypass-on state.

  FIG. 5 is a circuit diagram showing configurations of the second bypass circuit 270-2 and the third bypass circuit 270-3. The second bypass circuit 270-2 includes a second switch level shift circuit 254-2, a second bypass / limiter circuit 250-2, a second disconnection detection circuit 272-2, and a second short detection circuit 274-2. And a second detection signal level shift circuit 276-2 and a second integration circuit 278-2.

  The second switch level shift circuit 254-2 receives the second lighting on / off control signal Sc2 from the bypass drive circuit 112, and uses the second LED2-2 cathode voltage as a reference, that is, a low level, as a second bypass switch drive signal Sd2. Convert to The phase of the second bypass switch drive signal Sd2 is aligned with the phase of the second lighting on / off control signal Sc2, but the low level of the second bypass switch drive signal Sd2 is the cathode voltage of the second LED 2-2. In this way, the second switch level shift circuit 254-2 shifts the level of the second lighting on / off control signal Sc2 and supplies it to the corresponding second bypass / limiter circuit 250-2.

  The second bypass / limiter circuit 250-2 includes a second bypass switch 110-2 connected in parallel with the second LED 2-2. When the second bypass switch drive signal Sd2 is at a high level (low level), the second bypass / limiter circuit 250-2 turns off the second LED 2-2 by turning on (off) the second bypass switch 110-2 ( Light up). Furthermore, the second bypass / limiter circuit 250-2 uses the second bypass switch 110-2 and the voltage across the second bypass switch 110-2 when the second bypass switch drive signal Sd2 is at a low level. Configured to limit the upper limit of.

  The second bypass / limiter circuit 250-2 includes a limiter Zener diode 256, a backflow prevention diode 258, a sixteenth resistor 260, and a second bypass switch 110-2. The second bypass switch 110-2 is composed of, for example, an N-channel MOSFET.

  The cathode of the limiter Zener diode 256 is connected to the drain of the second bypass switch 110-2. These connection nodes are connected to the first bypass connection wiring 280-1. The anode of limiter zener diode 256 is connected to the anode of backflow prevention diode 258. The second bypass switch drive signal Sd2 is input to the gate of the second bypass switch 110-2 via the sixteenth resistor 260. The source of the second bypass switch 110-2 is connected to the second bypass connection wiring 280-2.

  A series circuit of a limiter Zener diode 256 and a backflow prevention diode 258 is connected to the gate side of the second bypass switch 110-2 with respect to the second bypass switch drive signal Sd2 for turning on / off the second bypass switch 110-2. . That is, the cathode of the backflow prevention diode 258 is connected between the sixteenth resistor 260 and the gate of the second bypass switch 110-2.

  The voltage across the second bypass switch 110-2 when the Zener voltage of the limiter Zener diode 256 is 6V, Vf of the backflow prevention diode 258 is 0.5V, and the gate threshold voltage of the second bypass switch 110-2 is 2.5V. That is, when the drain-source voltage reaches 9V, the second bypass switch 110-2 starts to be turned on, so the upper limit value of the voltage across the second bypass switch 110-2, that is, the disconnection detection voltage becomes 9V.

  The Zener voltage of the limiter Zener diode 256 is lower than the Zener voltage defined by the second electrostatic protection Zener diode 252-2 so that the disconnection detection voltage is higher than the maximum value of Vf of the second LED 2-2. And is set to be lower than the sum of Vf of the second LED 2-2 and Vf of the third LED 2-3. For example, when each LED has substantially the same characteristics, when the maximum value of Vf of the LED = 6V and the Zener voltage of the second electrostatic protection Zener diode 252-2 = 20V, the Zener voltage of the limiter Zener diode 256 is It is set in the range of 3V to 9V.

  The backflow prevention diode 258 is for preventing the second bypass switch 110-2 from being turned on / off by the second bypass switch drive signal Sd2. For example, when the second LED 2-2 connected in parallel is turned off, or the second bypass switch 110-2 is forcibly turned on as a measure for disconnection or contact failure described later, The gate voltage of the bypass switch 110-2 decreases from the forward direction of the limiter Zener diode 256 via the second bypass switch 110-2 that is on. The backflow prevention diode 258 prevents this situation from occurring.

  The second disconnection detection circuit 272-2 determines whether or not a conduction failure has occurred on the path of the drive current Iout that passes through the second LED 2-2 when the second bypass switch 110-2 is off (that is, in a non-conduction state). Determine. The second short detection circuit 274-2 determines whether or not a short circuit abnormality has occurred between the second LED 2-2 and the wiring when the second bypass switch 110-2 is OFF. When an abnormality is detected in the second disconnection detection circuit 272-2 or the second short detection circuit 274-2, the integration capacitor 282 of the second integration circuit 278-2 via the second detection signal level shift circuit 276-2. Is charged with a first time constant. This charging continues while the second bypass switch drive signal Sd2 is at a low level and an abnormality is detected in the second disconnection detection circuit 272-2 or the second short detection circuit 274-2. In other periods, the integrating capacitor 282 is discharged with a second time constant longer than the first time constant.

  The second detection signal level shift circuit 276-2 includes a seventeenth resistor 284, an eighteenth resistor 286, and a fourth pnp bipolar transistor 288. One end of the seventeenth resistor 284 is connected to the emitter of the fourth pnp bipolar transistor 288. A control power supply voltage Vcc is applied to these connection nodes. The other end of the seventeenth resistor 284 is connected to one end of the eighteenth resistor 286. These connection nodes are connected to the base of the fourth pnp bipolar transistor 288.

  The second disconnection detection circuit 272-2 includes a 23rd resistor 210 and a sixth npn-type bipolar transistor 212. The collector of the sixth npn-type bipolar transistor 212 is connected to the other end of the eighteenth resistor 286. The base of the sixth npn-type bipolar transistor 212 is connected to the connection node between the anode of the limiter Zener diode 256 and the anode of the backflow prevention diode 258 through the 23rd resistor 210. The emitter of the sixth npn-type bipolar transistor 212 is connected to the second bypass connection wiring 280-2.

  The second short detection circuit 274-2 includes a nineteenth resistor 290, a twentieth resistor 292, a twenty-first resistor 294, a twenty-second resistor 296, a fourth npn-type bipolar transistor 298, a fifth npn-type bipolar transistor 208, Have The twentieth resistor 292 and the twenty-second resistor 296 are connected in series between the first bypass connection wire 280-1 and the second bypass connection wire 280-2 in this order. A connection node between the twentieth resistor 292 and the twenty-second resistor 296 is connected to one end of the twenty-first resistor 294 and the base of the fourth npn-type bipolar transistor 298. The other end of the 21st resistor 294 is connected to the cathode of the backflow prevention diode 258. The emitter of the fourth npn-type bipolar transistor 298 is connected to the second bypass connection wiring 280-2. The collector of the fourth npn-type bipolar transistor 298 is connected to one end of the nineteenth resistor 290. A control power supply voltage Vcc is applied to the other end of the nineteenth resistor 290. A connection node between the collector of the fourth npn-type bipolar transistor 298 and the nineteenth resistor 290 is connected to the base of the fifth npn-type bipolar transistor 208. The emitter of the fifth npn bipolar transistor 208 is connected to the second bypass connection wiring 280-2, and the collector is connected to the other end of the eighteenth resistor 286.

  The second integration circuit 278-2 includes an integration capacitor 282, a 24th resistor 214, and a 25th resistor 216. One end of the 24th resistor 214 is grounded to a ground potential substantially equal to the ground potential of the microcomputer, and the other end is connected to one end of the 25th resistor 216. A connection node between the 24th resistor 214 and the 25th resistor 216 is connected to the collector of the fourth pnp bipolar transistor 288. The other end of the 25th resistor 216 is connected to one end of the integrating capacitor 282. The other end of the integrating capacitor 282 is grounded to a ground potential substantially equal to the ground potential of the microcomputer. The voltage at one end of the integration capacitor 282 is supplied to the bypass drive circuit 112 as the second abnormality detection signal Sdet2.

  When conduction failure occurs on the path of the drive current Iout, the voltage across the second bypass switch 110-2 rises above the forward drop voltage Vf during the OFF period of the second bypass switch 110-2. When the voltage between both ends exceeds the disconnection detection voltage, a current flows through the limiter Zener diode 256. Then, the sixth npn-type bipolar transistor 212 of the second disconnection detection circuit 272-2 is turned on.

Further, when a short circuit occurs in the second LED 2-2, the voltage across the second bypass switch 110-2 is lower than the forward drop voltage Vf during the OFF period of the second bypass switch 110-2. When the both-end voltage falls below the short-circuit detection voltage, the fourth npn-type bipolar transistor 298 of the second short-circuit detection circuit 274-2 is turned off, and the fifth npn-type bipolar transistor 208 is turned on.
The resistance values of the twentieth resistor 292, the twenty-first resistor 294, and the twenty-second resistor 296 are set so as to satisfy the following three conditions.
(1) When a high level is applied to the other end of the twenty-first resistor 294, the fourth npn-type bipolar transistor 298 is turned on.
(2) When the low level is applied to the other end of the 21st resistor 294 and when the voltage across the second bypass switch 110-2 is equal to or higher than the short detection voltage, the fourth npn type bipolar transistor 298 Turn on.
(3) When the low level is applied to the other end of the 21st resistor 294, and when the voltage across the second bypass switch 110-2 is lower than the short detection voltage, the fourth npn bipolar transistor 298 is turned off.

  When either the sixth npn-type bipolar transistor 212 or the fifth npn-type bipolar transistor 208 is turned on, the fourth pnp-type bipolar transistor 288 of the second detection signal level shift circuit 276-2 is turned on. The control power supply voltage Vcc is a positive voltage based on the ground potential of the microcomputer. When an abnormality is detected in the second disconnection detection circuit 272-2 or the second short detection circuit 274-2, the control power supply voltage Vcc is supplied to the second integration circuit 278-2. The integration capacitor 282 is charged via the 25th resistor 216 and discharged via the 24th resistor 214 and the 25th resistor 216. By setting the resistance value of the twenty-fourth resistor 214 to be larger than the resistance value of the twenty-fifth resistor 216, an average current of 10% lighting (second bypass switch 110-2 on = 90%, off = 10%), It is possible to detect an abnormality even during dimming when the period for detecting an abnormality (that is, the off period of the second bypass switch 110-2) is short.

The third bypass circuit 270-3 is configured in the same manner as the second bypass circuit 270-2, and includes a third switch level shift circuit 254-3 corresponding to the second switch level shift circuit 254-2, and a second bypass / Corresponds to the third bypass / limiter circuit 250-3 corresponding to the limiter circuit 250-2, the third disconnection detection circuit 272-3 corresponding to the second disconnection detection circuit 272-2, and the second short detection circuit 274-2. The third short detection circuit 274-3, the third detection signal level shift circuit 276-3 corresponding to the second detection signal level shift circuit 276-2, and the third short circuit corresponding to the second integration circuit 278-2. Integrating circuit 278-3.
Each of the first bypass circuit 270-1, the fourth bypass circuit 270-4 to the Nth bypass circuit 270-N is configured similarly to the second bypass circuit 270-2.

The operation of the semiconductor light source control device 100 having the above configuration will be described.
FIGS. 6A to 6C are graphs showing the time change of the drive current Iout. Consider a situation where only one LED is lit, then about half are lit, and then all LEDs are lit. Here, PWM dimming is not considered. FIG. 6A shows the time change of the drive current Iout when only one LED is turned on, and the remaining N−1 LEDs are turned off by turning on the corresponding bypass switch. FIG. 6B shows a change with time of the drive current Iout when about half, that is, N / 2 LEDs are turned on and the remaining LEDs are turned off. FIG. 6C shows a time change of the drive current Iout when all the LEDs are turned on.

  FIGS. 6A to 6C show a case where the hysteresis width ΔI is adjusted so that the switching frequency of the second switching element 140, that is, the switching period Ts becomes substantially constant regardless of the number of turned on / off of LEDs. It is. However, in the present embodiment, the hysteresis width ΔI only needs to be controlled so that the change in the switching period Ts due to the change in the number of LEDs that are turned on and off can be controlled by those skilled in the art who have touched this specification. Understood.

  Referring to FIG. 6A, when the number of LEDs to be lit is small, the driving current Iout rises relatively quickly during the on time Ton of the second switching element 140, and the driving current during the off time Toff of the second switching element 140. Iout falls relatively slowly. The hysteresis width at this time is expressed as ΔI1. The absolute value of the drive voltage Vout is relatively low, and the offset voltage Voffset generated by the hysteresis width setting circuit 138 is also relatively low.

  Referring to FIG. 6B, when the number of LEDs to be turned on and the number of LEDs to be turned off are about the same number, the drive voltage Vout is about half of the set voltage of the flyback regulator 102, and the second switching element 140 The on time Ton and the off time Toff antagonize. The overall speed of change of the drive current Iout is greater than when the number of LEDs to be lit is small.

  The hysteresis width setting circuit 138 generates a higher offset voltage Voffset as shown in FIG. The down converter drive circuit 136 receives the high offset voltage Voffset, and makes the hysteresis width ΔI2 larger than the hysteresis width ΔI1 when the number of lighting LEDs is 1. As a result, the increase in the overall change speed of the drive current Iout is canceled out, and the switching period Ts is kept substantially constant.

  Referring to FIG. 6C, when the number of LEDs to be turned off is small or not, the drive current Iout rises relatively slowly during the on-time Ton of the second switching element 140, and the off-time Toff of the second switching element 140. The drive current Iout falls relatively quickly. The overall speed of change of the drive current Iout is smaller than when the number of LED lighting and the number of light extinguishing antagonize. The absolute value of the drive voltage Vout is relatively high, and the offset voltage Voffset generated by the hysteresis width setting circuit 138 is relatively low.

  The down-converter driving circuit 136 receives a low offset voltage Voffset, and makes the hysteresis width ΔI3 smaller than the hysteresis width ΔI2 when the number of LED lighting and the number of light-offs compete. As a result, the decrease in the overall change rate of the drive current Iout is canceled out, and the switching period Ts is kept substantially constant.

  FIG. 7 is a timing chart showing changes in the second bypass switch drive signal Sd2 and the second abnormality detection signal Sdet2 during PWM dimming. Here, the second bypass switch drive signal Sd2 corresponds to lighting with an average current of 20%. That is, the length Toff of the period during which the second bypass switch drive signal Sd2 is at the low level is about one quarter of the length Ton of the period during which the second bypass switch drive signal Sd2 is at the high level.

  When no abnormality has occurred, the second abnormality detection signal Sdet2 is kept constant at the ground potential of the microcomputer, that is, near 0V, regardless of the change of the second bypass switch drive signal Sd2. Here, it is assumed that conduction failure or disconnection occurs at time t1 during the period in which the second bypass switch drive signal Sd2 is at the high level. At time t1, the second bypass switch drive signal Sd2 is at a high level, that is, the second bypass switch 110-2 is on (that is, conductive), so both the sixth npn bipolar transistor 212 and the fifth npn bipolar transistor 208 Keeps off. Accordingly, the potential of the second abnormality detection signal Sdet2 does not substantially change.

  At time t2 when the second bypass switch drive signal Sd2 transitions from the high level to the low level, either the sixth npn-type bipolar transistor 212 or the fifth npn-type bipolar transistor 208 is turned on due to the abnormality that has occurred, and the integrating capacitor 282 Charging is started.

  The integration capacitor 282 is fully charged at time t3 during the period when the second bypass switch drive signal Sd2 is at the low level. The voltage of the second abnormality detection signal Sdet2 when the integration capacitor 282 is fully charged is higher than the level threshold value Vg for determining the level of the second abnormality detection signal Sdet2 in the bypass drive circuit 112.

At time t4 when the second bypass switch drive signal Sd2 transitions from the low level to the high level, both the sixth npn-type bipolar transistor 212 and the fifth npn-type bipolar transistor 208 are turned off, and discharging of the integrating capacitor 282 is started.
After time t3, the second abnormality detection signal Sdet2 becomes higher than the level threshold value Vg, but the operation speed of the bypass drive circuit 112, which is a microcomputer, is relatively slow, so the level of the second bypass switch drive signal Sd2 for a while. The transition continues.
When the abnormality that has occurred continues at time t5 when the second bypass switch drive signal Sd2 next transitions from the high level to the low level, recharging of the integration capacitor 282 is started. Thereafter, the integrating capacitor 282 is fully charged again.

Since the second time constant related to the discharge of the integrating capacitor 282 is set longer than the first time constant related to the charging, the time between time t4 and time t5 when the second bypass switch drive signal Sd2 is at the high level. Even during this period, the voltage of the second abnormality detection signal Sdet2 remains higher than the level threshold value Vg. Thus, after the second bypass switch drive signal Sd2 repeats level transition several times, the bypass drive circuit 112 determines that the second abnormality detection signal Sdet2 has transitioned from the low level to the high level.
FIG. 7 illustrates the case where the integration capacitor 282 reaches full charge in one off period of the second bypass switch 110-2. However, the present invention is not limited to this, and the integration capacitor 282 is full whenever the off period is repeated. It may be approaching charging.

  The operation of the semiconductor light source control device 100 when a conduction failure occurs in any of the first bypass connection wiring 280-1 to the (N-1) th bypass connection wiring 280- (N-1) will be described. As an example, let us consider a case where a continuity failure occurs in the second bypass connection wiring 280-2, that is, a case where a continuity failure occurs at a location indicated by “x” indicated by reference numeral 218 of the circuit shown in FIG.

  When conduction failure occurs in the second bypass connection wiring 280-2, the potential of the connection node between the source of the second bypass switch 110-2 and the drain of the third bypass switch 110-3 is floated. When both the second bypass switch 110-2 and the third bypass switch 110-3 are turned off, the connection is caused by the difference in characteristics between the second bypass switch 110-2 and the third bypass switch 110-3. The potential of the node approaches the potential of the source of the second bypass switch 110-2 or the drain of the third bypass switch 110-3. For example, when the potential of the connection node sufficiently approaches the potential of the drain of the third bypass switch 110-3, the third short detection circuit 274-3 determines that a short circuit abnormality has occurred. The third abnormality detection signal Sdet3 becomes high level, and the bypass drive circuit 112 fixes the third lighting on / off control signal Sc3 to high level.

Then, the third bypass switch 110-3 is turned on, and the voltage across the second bypass switch 110-2 is substantially 2 Vf. Since the disconnection detection voltage is lower than 2 Vf, the second disconnection detection circuit 272-2 determines that a conduction failure has occurred. The second abnormality detection signal Sdet2 becomes high level, and the bypass drive circuit 112 fixes the second lighting on / off control signal Sc2 at high level.
As a result, both the second bypass switch 110-2 and the third bypass switch 110-3 are forcibly turned on, and both the second LED 2-2 and the third LED 2-3 are maintained in the off state.
When the potential of the connection node between the source of the second bypass switch 110-2 and the drain of the third bypass switch 110-3 approaches the potential of the source of the second bypass switch 110-2, the second short detection circuit 274 is used. -2 detects a short circuit abnormality, the third disconnection detection circuit 272-3 detects a disconnection abnormality, and both the second bypass switch 110-2 and the third bypass switch 110-3 are forcibly turned on.

Temporarily, both the two bypass switches are not forcedly turned on, the second bypass switch drive signal Sd2 is set to low level to turn on the second LED 2-2, and the third LED 2-3 is turned off. When the third bypass switch drive signal Sd3 is set to the high level, the third bypass switch 110-3 is turned on, but the drive current Iout does not flow through the third bypass switch 110-3 but flows through the third LED 2-3. That is, the third LED 2-3 cannot be turned off. When the second bypass switch drive signal Sd2 is at a high level and the third bypass switch drive signal Sd3 is at a low level, the second bypass switch 110-2 is turned on, but the drive current Iout is supplied to the second bypass switch 110-2. It flows through the second LED 2-2 without flowing. That is, the second LED 2-2 cannot be turned off.
As described above, when a conduction failure occurs in the second bypass connection wiring 280-2, it is difficult to individually control the on / off states of the second LED 2-2 and the third LED 2-3.

  In the semiconductor light source control device 100 according to the present embodiment, when a conduction failure occurs in any of the first bypass connection wiring 280-1 to the (N-1) th bypass connection wiring 280- (N-1). Both of the two bypass switches connected to the bypass connection wiring in which the conduction failure has occurred are forcibly turned on. As a result, both the two LEDs connected to the bypass connection wiring are turned off.

  When a continuity failure occurs in the bypass connection wiring, it is difficult to individually control the on / off states of the two LEDs connected to the bypass connection wiring as described above. For example, consider a case where the semiconductor light source control device 100 is used for a high beam of a vehicle headlamp, and the semiconductor light source control device 100 has a function of turning off the corresponding LED so as not to give glare to a preceding vehicle or an oncoming vehicle. . In this case, if a continuity failure occurs in the bypass connection wiring connected to the corresponding LED, the corresponding LED cannot be turned off and glare can be given. Therefore, in this embodiment, when a conduction failure occurs in the bypass connection wiring, both of the two bypass switches connected to the bypass connection wiring are forcibly turned on, so that the bypass connection wiring in which the conduction failure has occurred It becomes irrelevant to the Iout bypass path, and the corresponding LED can be maintained in the extinguished state. Thereby, it can avoid giving glare.

Moreover, the semiconductor light source control device 100 according to the present embodiment has a PWM dimming function that adjusts the brightness of the corresponding LED by turning on and off the bypass switch at a relatively high speed. When this bypass switch is on, the voltage across the bypass switch is close to 0V, but this is not abnormal, so it should not be determined as a short circuit abnormality. Therefore, the semiconductor light source control device 100 is configured not to detect short circuit abnormality or disconnection abnormality when the bypass switch is on, but to detect those when the bypass switch is off.
Therefore, when an abnormality occurs in the LED or wiring that is dimmed by PWM, the state in which the bypass switch is off and the abnormality appears and the state in which the bypass switch is on regardless of whether there is an abnormality are repeated at high speed. It is necessary to make an abnormality judgment. A microcomputer is often used as a main device for controlling the turning on / off of the LED and the PWM dimming function. Usually, the microcomputer operates at a comparatively long time interval of about several tens of milliseconds, so that it is not suitable for picking up and judging only abnormalities of many LEDs at high speed. For example, when the dimming frequency f1 is several kHz and the average current is 10%, the length of the off period of the bypass switch is on the order of several hundred microseconds. Thus, in order to determine abnormality / normality in a relatively short period, it is necessary to employ an expensive microcomputer having a relatively high operation speed.

  Therefore, the semiconductor light source control apparatus 100 according to the present embodiment generates an abnormality detection signal that is at a high level during an abnormality and at a low level during a normal operation, regardless of PWM dimming, by the action of the integration circuit. The bypass drive circuit 112, which is a microcomputer, determines normality / abnormality based on the abnormality detection signal. As a result, when the PWM dimming function using the bypass switch is employed, it is possible to detect an abnormality in the LED or wiring without employing an expensive microcomputer.

  In addition, according to the semiconductor light source control device 100 according to the present embodiment, it is possible to suppress an increase in voltage applied to the bypass switch even when a contact failure such as contact failure or disconnection occurs on the path of the drive current Iout. . For example, when the first LED 2-1 is in a lit state, that is, when the first bypass switch 110-1 is off, the upstream side of the connection node between the anode of the first LED 2-1 and the cathode of the first electrostatic protection Zener diode 252-1 Let us consider a case in which a contact failure or disconnection occurs in the wiring indicated by “x” indicated by reference numeral 262 of the circuit shown in FIG.

  When such a contact failure or disconnection occurs, the voltage across the first bypass switch increases, and the first abnormality detection signal Sdet1 transitions from a low level to a high level. When the bypass drive circuit 112 detects such a transition of the first abnormality detection signal Sdet1, it determines that an abnormality has occurred with respect to the first LED 2-1, and turns on the first bypass switch 110-1 in the circuit shown in FIG. Then, measures are taken so that other LEDs can be turned on.

  However, since the operation speed of the bypass drive circuit 112, which is a microcomputer, is relatively slow as described above, this measure usually takes several tens of milliseconds to several hundreds of milliseconds. Here, when the semiconductor light source control device does not have the limiter function according to the present embodiment, immediately after the contact failure or disconnection as described above occurs due to the absence of the output voltage smoothing capacitor, A relatively high voltage such as several kV (absolute value) determined by the energy stored in the inductor 144 and the parasitic capacitance of the first bypass switch is output. Such a high voltage is applied to the first bypass switch before the first bypass switch is turned on. Therefore, it is necessary to select an element having a withstand voltage of several kV as the first bypass switch in consideration of poor contact or disconnection, although only a voltage of several volts is applied during normal lighting.

  On the other hand, according to the semiconductor light source control device 100 having a limiter function according to the present embodiment, the drain-source voltage of the first bypass switch 110-1 rises when the above disconnection or poor contact occurs. However, the rise of the voltage is limited by the action of the limiter Zener diode 256 and the first bypass switch 110-1 itself. Therefore, an element having a lower withstand voltage can be selected as the first bypass switch 110-1 even in consideration of poor contact or disconnection.

  Here, when disconnection or poor contact occurs, about 10 V × 1A = 10 W is applied to the first bypass switch 110-1 as an example for several tens of milliseconds to several hundreds of milliseconds. Since it is necessary to use a large device, the influence on the device size and cost is small.

  For example, when the first LED 2-1 is lit, that is, when the first bypass switch 110-1 is off, the downstream side of the connection node between the anode of the first LED 2-1 and the cathode of the first electrostatic protection Zener diode 252-1 Let us consider a case in which a contact failure or disconnection occurs in the wiring shown in FIG. When the semiconductor light source control device does not have the limiter function according to the present embodiment, most of the energy stored in the inductor 144 is consumed by the first electrostatic protection Zener diode. Therefore, it is necessary to select an element that can withstand such a large power consumption as the first electrostatic protection Zener diode. Alternatively, as the first electrostatic protection zener diode, an element having a zener voltage higher than a voltage of several kV that may occur when a contact failure or disconnection occurs may be considered. If it is high, it cannot play the role of original static electricity protection.

  On the other hand, according to the semiconductor light source control device 100 having the limiter function according to the present embodiment, the upper limit value of the voltage across the first bypass switch 110-1 is defined by the first electrostatic protection Zener diode 252-1. It is set to be lower than the zener voltage. Therefore, a relatively small Zener diode can be selected as the first electrostatic protection Zener diode 252-1.

  Similarly, when a similar contact failure or disconnection occurs in any of the second LED 2-2 to the N-th LED 2-N, the upper limit of the voltage applied to the corresponding bypass switch or electrostatic protection zener diode is limited. The Therefore, an element having a lower withstand voltage can be employed as the corresponding bypass switch, and a relatively small Zener diode can be employed as the corresponding electrostatic protection Zener diode.

  In the semiconductor light source control device 100 according to the present embodiment, the bypass switch for controlling turning on / off of the LED is also used as a switch for realizing a limiter function with respect to the voltage across the LED. That is, the bypass switch is shared by the lighting on / off control function and the limiter function. As a result, an increase in the number of elements can be suppressed while realizing the lighting on / off control function and the limiter function.

  In the semiconductor light source control device 100 according to the present embodiment, since no smoothing capacitor is provided in the output stage to the N LEDs 2-1 to 2-N, the drive current Iout follows the second switching element 140. Sex is better. In particular, when the second switching element 140 is turned off, the drive current Iout decreases, and when the second switching element 140 is turned on, the drive current Iout increases. In order to stabilize the drive current Iout near the target value, hysteresis control of the drive current Iout is employed instead of smoothing. As a result, the response in the current feedback can be speeded up. For example, when the number of lighted LEDs changes due to the action of the bypass drive circuit 112 and the bypass switch, the drive current lout can be made to follow the load variation more quickly. In particular, it is possible to suppress undershoot of the drive current Iout when the number of lighting of the LED is increased and overshoot of the drive current Iout when the number of lighting is decreased.

In the semiconductor light source control device 100 according to the present embodiment, the flyback regulator 102 at the front stage has a negative output, and the down converter 104 at the rear stage also has a negative output. As a result, an N-channel MOSFET having better characteristics can be employed as the bypass switch.
In addition to the negative output, the inductor 144 is provided not between the cathode and the output of the flywheel diode 142 but between the anode and the output, so that the second switching element 140 of the down converter 104 has a better N characteristic. A channel MOSFET can be employed. Further, the drive voltage Vout can be detected stably.

  In addition, when the semiconductor light source control device has a positive output, the detection of the drive current is often performed on the high side in consideration of the case where the LED is grounded. Here, when the load changes, the potential at the detection location also changes, so that it is difficult to accurately detect the drive current. Also, the configuration of the detection circuit can be more complicated. Therefore, in the semiconductor light source control device 100 according to the present embodiment, the negative output is adopted, and the current detection resistor 108 is provided on the output on the positive side, that is, the ground side. As a result, even if the load (drive voltage Vout) changes, the change does not affect the potential of the detection location of the drive current Iout, and the drive current Iout can be detected stably. In addition, the configuration of the detection circuit can be simplified.

  When the drive current lout is subjected to hysteresis control, if the input voltage to the down converter 104 and / or the drive voltage Vout change, the slope of the rise or fall of the drive current lout changes, so the switching frequency of the second switching element 140 changes. Yes. Therefore, in the semiconductor light source control device 100 according to the present embodiment, the hysteresis width ΔI is adjusted so that the change of the switching frequency is suppressed. In particular, by setting the target switching frequency so as to avoid a known frequency band of radio noise, adverse effects on the semiconductor light source control device 100 due to radio noise can be suppressed.

  Moreover, in the semiconductor light source control device 100 according to the present embodiment, the fluctuation of the input voltage to the down converter 104 due to the fluctuation of the battery voltage Vbat is suppressed by the action of the flyback regulator 102. Therefore, a change in switching frequency due to a change in input voltage to the down converter 104 can be suppressed. In other words, it is not necessary to select the hysteresis width ΔI based on the combination of the input voltage to the down converter 104 and the drive voltage Vout, and it is only necessary to select the hysteresis width ΔI based mainly on the drive voltage Vout, so the hysteresis width ΔI is adjusted. The control for this is further simplified. This contributes to downsizing and speeding up of the control circuit.

  In the semiconductor light source control device 100 according to the present embodiment, an output capacitor 128 is provided at the output stage of the flyback regulator 102. If the second switching element 140 is turned on when the bypass switch is turned on, the charge stored in the output capacitor 128 tends to flow to the LED at once. However, since the semiconductor light source control device 100 is provided with the inductor 144 on the path of the drive current Iout, such a charge flow is smoothed, and the overshoot of the drive current Iout is suppressed. Similarly, when the bypass switch is turned off, the undershoot of the drive current Iout is suppressed.

Consider a semiconductor light source lighting circuit 300 according to the following comparative example, which was originally created to suppress overshoot and undershoot of the drive current Iout when the bypass switch is switched.
FIG. 8 is a circuit diagram showing a configuration of a semiconductor light source lighting circuit 300 according to a comparative example. The semiconductor light source lighting circuit 300 is basically a forward converter that does not use a smoothing capacitor. The semiconductor light source lighting circuit 300 includes a control circuit 302, an input capacitor 306, a reset circuit 308, a transformer 310, a fifth switching element 312, a second diode 314, a third diode 316, an inductor 318, a current And a detection resistor 320.
The control circuit 302 turns off the fifth switching element 312 when the magnitude of the drive current exceeds a predetermined current upper limit value, and turns on the fifth switching element 312 when the magnitude of the drive current falls below the current lower limit value.

For the semiconductor light source lighting circuit 300, the winding ratio of the transformer 310 is N _{s / p} , the inductance of the inductor 318 is Ls ′, the hysteresis width of the drive current is ΔI ′, the input voltage is Vin, the output voltage is Vout (<0), If the on-time of the fifth switching element 312 is Ton ′, the off-time is Toff ′, the switching frequency is F ′, and the forward drop voltage of the rectifier diode is small, so that it can be ignored, F ′ can be obtained from the following equation. .

In the semiconductor light source lighting circuit 300, when the winding ratio of the transformer 310 = 16.7 (input = 6V is converted to output = 100V), the inductance of the inductor 318 = 500 μH, and the hysteresis width = 0.1 A, Equation 1 The relationship between Vin, Vout, and F ′ obtained from the equation (1) is as shown in Table 1 below. Here, input voltage fluctuation = 6V to 20V, output (load) voltage fluctuation = −4V to −88V (22 LEDs in series with Vf = 4V) were assumed.

  In this case, the switching frequency F 'varies about 17 times at the maximum / minimum. If the inductance is increased, this fluctuation range can be suppressed, but the circuit becomes larger. In addition, when the function of suppressing the large fluctuation of the switching frequency F ′ from the input voltage and the output voltage and suppressing it to a predetermined range is realized, the control circuit scale becomes large.

The same calculation is performed for the semiconductor light source control apparatus 100 according to the present embodiment. Regarding the semiconductor light source control device 100, assuming that the inductance of the inductor 144 is Ls, the switching frequency is F, and the forward voltage drop of the flywheel diode 142 is small, F can be obtained from the following equation.

In the semiconductor light source control device 100, when the target voltage Vt = −100V, the inductance of the inductor 144 = 500 μH, and the hysteresis width = 0.1 A, the relationship between Vt, Vout, and F obtained from Equation 2 is as follows. It is as 2 tables.

  In this case, the fluctuation of the switching frequency F is suppressed to about 6.5 times. The main parameter causing the fluctuation is the drive voltage Vout and the target voltage Vt is substantially fixed. Therefore, the scale of the control circuit for adjusting the hysteresis width ΔI so as to suppress the fluctuation of the switching frequency F is relatively small. it can.

  Looking at the theoretical calculation value of the switching frequency F in Table 2, the switching frequency F increases as the drive voltage Vout decreases from −4V to −44V, and as the drive voltage Vout decreases from −44V to −88V. The switching frequency F decreases. The increase / decrease boundary of the switching frequency F is −50 V, which is about half of the output voltage of the first stage (front stage) flyback regulator 102 (the input voltage of the second stage (rear stage) down converter 104). Therefore, it is possible to easily keep the switching frequency F within a predetermined range by controlling so that the hysteresis width ΔI is increased as the drive voltage Vout is lower at Vout> −50V, and is decreased at Vout <−50V. It becomes.

  Further, in the present embodiment, it has been found that the boundary of the increase / decrease of the switching frequency F is at about half of the output voltage of the flyback regulator 102. However, in other embodiments involving other circuit arrangements, This boundary may be one third or one quarter of the output voltage. What can be said in common is that Vout that gives the maximum value of the switching frequency F when the hysteresis width is constant may exist between the maximum value and the minimum value of Vout. Therefore, if such a Vout is found through experiments or simulations, and the circuit is configured such that the hysteresis width ΔI is minimized at such Vout, fluctuations in the switching frequency F can be more suitably suppressed.

Ｖｏｆｆｓｅｔは、図２に示されるヒステリシス幅設定回路１３８の回路定数を調整し、Ｖｏｕｔ＝−５０Ｖ付近で図３に示されるグラフのように電圧値が高くなるようにしたものである。下限電圧・上限電圧は、図４に示されるダウンコンバータ駆動回路１３６の第１２抵抗１８２と第１３抵抗１８４との分圧ノードの電圧であり、電流下限値Ｉｔｈ２、電流上限値Ｉｔｈ１にそれぞれ対応する。下限電圧・上限電圧は、第８抵抗１７４、第１２抵抗１８２、第１３抵抗１８４の各抵抗値と制御電源電圧Ｖｃｃを設定し、オフセット電圧Ｖｏｆｆｓｅｔから算出したものである。平均電流は電流上限値Ｉｔｈ１と電流下限値Ｉｔｈ２との平均値である。スイッチング周波数は、式２と同じ算出式において、ΔＩ＝Ｉｔｈ１−Ｉｔｈ２、Ｖｔ＝−１００Ｖ、Ｌｓ＝２００μＨとして求めたものである。
Ｌｓを５００μＨから２００μＨへと小さくしても、スイッチング周波数を５５０ｋＨｚ強〜４００ｋＨｚ弱の範囲に収めることができることが分かる。すなわち、本実施の形態に係る半導体光源制御装置１００によると、駆動電流Ｉｏｕｔを平滑化するためのインダクタンスを小型化できる。 A parameter setting example for the semiconductor light source control apparatus 100 according to the present embodiment is shown in Table 3 below.

Voffset is obtained by adjusting the circuit constant of the hysteresis width setting circuit 138 shown in FIG. 2 so that the voltage value becomes high as shown in the graph shown in FIG. 3 near Vout = −50V. The lower limit voltage and the upper limit voltage are voltages of voltage dividing nodes of the twelfth resistor 182 and the thirteenth resistor 184 of the down converter drive circuit 136 shown in FIG. 4, and correspond to the current lower limit value Ith2 and the current upper limit value Ith1, respectively. . The lower limit voltage and the upper limit voltage are calculated from the offset voltage Voffset by setting the resistance values of the eighth resistor 174, the twelfth resistor 182, and the thirteenth resistor 184 and the control power supply voltage Vcc. The average current is an average value of the current upper limit value Ith1 and the current lower limit value Ith2. The switching frequency is obtained in the same calculation formula as Expression 2 with ΔI = Ith1−Ith2, Vt = −100V, and Ls = 200 μH.
It can be seen that even if Ls is reduced from 500 μH to 200 μH, the switching frequency can be kept in the range of slightly higher than 550 kHz to slightly lower than 400 kHz. That is, according to the semiconductor light source control device 100 according to the present embodiment, the inductance for smoothing the drive current Iout can be reduced.

  Further, when comparing the semiconductor light source lighting circuit 300 according to the comparative example and the semiconductor light source control device 100 according to the present embodiment, the semiconductor light source control device 100 uses the output capacitor 128 of the flyback regulator 102 and the second switching of the down converter 104. Although the number of elements 140 is increased, the reset circuit 308 can be reduced from the semiconductor light source lighting circuit 300, so that the circuit scale is almost the same.

  The configuration and operation of the semiconductor light source control device according to the embodiment have been described above. This embodiment is an exemplification, and it is understood by those skilled in the art that various modifications can be made to each component and combination of processes, and such modifications are within the scope of the present invention.

  In the embodiment, the case where the second switching element 140 is arranged on the cathode side of the flywheel diode 142 and the inductor 144 is arranged on the anode side of the flywheel diode 142 as the element arrangement of the down converter 104 has been described. I can't. The flywheel diode may be connected in parallel with the output capacitor 128 of the flyback regulator 102. If the second switching element is provided on the path of the drive current Iout from one end of the output capacitor 128 to the LED and back from the LED to the other end of the output capacitor 128, the second switching element is provided between the output capacitor 128 and the flywheel diode. Good. On / off of the second switching element may be controlled based on the drive current. The inductor 144 may be provided on the path of the drive current Iout and provided between the flywheel diode and the LED.

  FIGS. 9A to 9C are circuit diagrams showing configurations of semiconductor light source control devices 400, 500, and 600 according to the first, second, and third modifications. FIG. 9A shows a configuration of a semiconductor light source control device 400 according to the first modification. One end of the second switching element 440 is connected to the high potential side output of the flyback regulator 102, and the other end is connected to the cathode of the flywheel diode 442. One end of the inductor 444 is connected to a connection node between the other end of the second switching element 440 and the cathode of the flywheel diode 442. The other end of the inductor 444 is grounded and serves as a high potential side output terminal to the LED. The anode of the flywheel diode 442 is connected to the low potential side output of the flyback regulator 102 and serves as a low potential side output terminal to the LED.

  FIG. 9B shows a configuration of a semiconductor light source control device 500 according to the second modification. The cathode of the flywheel diode 542 is connected to the high potential side output of the flyback regulator 102 and grounded to form a high potential side output to the LED. One end of the second switching element 540 is connected to the low potential side output of the flyback regulator 102, and the other end is connected to the anode of the flywheel diode 542. One end of the inductor 544 is connected to a connection node between the other end of the second switching element 540 and the anode of the flywheel diode 542. The other end of the inductor 544 serves as a low potential side output terminal to the LED.

  FIG. 9C shows a configuration of a semiconductor light source control device 600 according to the third modification. One end of the second switching element 640 is connected to the low potential side output of the flyback regulator 102, and the other end is connected to the anode of the flywheel diode 642. A connection node between the other end of the second switching element 640 and the anode of the flywheel diode 642 forms a low potential side output to the LED. The cathode of flywheel diode 642 is connected to one end of inductor 644. A connection node between the cathode of the flywheel diode 642 and one end of the inductor 644 is connected to the high potential side output of the flyback regulator 102. The other end of the inductor 644 is grounded and becomes a high potential side output terminal to the LED.

  According to each of the semiconductor light source control devices 400, 500, and 600 according to the first, second, and third modified examples, as with the semiconductor light source control device 100 according to the embodiment, overshoot and undershoot of the drive current Iout are performed. Can be reduced.

  In the embodiment, the case where the negative output is realized by grounding the high potential side of the output, that is, the anode side of the plurality of LEDs, is not limited to this. For example, the anode side of the plurality of LEDs is connected to the battery voltage. It may be connected to a terminal to which a DC voltage such as Vbat is applied.

  In the embodiment, the switching frequency is not measured in real time, and instead, the relationship between the drive voltage Vout and the hysteresis width ΔI is determined based on the known relationship between the drive voltage Vout and the switching frequency, and the hysteresis width ΔI is in that relationship. Therefore, although the case where the circuit is configured to change is described, the present invention is not limited to this. For example, the semiconductor light source control device may include a circuit that measures the switching frequency of the second switching element 140, and adjust the hysteresis width so that the switching frequency thus measured falls within the target frequency range.

  In the embodiment, the case where the semiconductor light source control device 100 includes N bypass switches 110-1 to 110-N has been described. However, the present invention is not limited to this, and the bypass switch is provided separately from the semiconductor light source control device. May be.

  In the embodiment, the case where the hysteresis control of the drive current is performed has been described. However, the present invention is not limited to this. For example, the second switching element is set so that the voltage obtained by appropriately filtering the voltage drop Vm approaches the reference voltage corresponding to the target current. A duty ratio of 140 may be controlled.

  In the embodiment, a case has been described in which a drive circuit that performs control to generate a drive current Iout by combining the flyback regulator 102 and the downconverter 104 and bring the magnitude of the drive current Iout closer to a target value has been described. However, the present invention is not limited to this, and for example, the circuit shown in FIG. 8 may be adopted as such a drive circuit, or a flyback regulator controlled by current feedback may be adopted.

  FIG. 10 is a circuit diagram showing a configuration of a semiconductor light source control device 700 according to a fourth modification and members connected thereto. The semiconductor light source control device 700 includes a flyback regulator 702, a current detection resistor 708, N bypass circuits 270-1 to 270 -N, and a bypass drive circuit 112.

  In the semiconductor light source control apparatus 700 according to the present modification, when a conduction failure occurs at a location indicated by “x” indicated by reference numeral 718 of the second bypass connection wiring 280-2, the semiconductor light source control according to the present embodiment. Similar to the apparatus 100, both the second bypass switch and the third bypass switch are forcibly turned on by the action of the second bypass circuit 270-2, the third bypass circuit 270-3, and the bypass drive circuit 112.

  Further, the limit value of the maximum voltage output from the flyback regulator 702 is set to be equal to or greater than the sum of Vf in consideration of the case where all N LEDs connected in series are lit. For example, when the maximum value of Vf of one LED is 6V and 30 LEDs are connected in series, the limit value is set to 180V or more. Here, since the drive current Iout does not flow to the LED at the moment when contact failure or disconnection occurs in the wiring of “x” indicated by the reference numeral 762 in FIG. 10, the output voltage of the flyback regulator 702 is 180V. Ascend toward. When the control circuit (not shown) detects that the drive current Iout no longer flows, it checks which wiring or LED is broken, turns on the first bypass switch 110-1 in the circuit of FIG. Take measures to turn on other LEDs. This measure usually takes tens of milliseconds to hundreds of milliseconds

  Here, when the semiconductor light source control device does not have the limiter Zener diode 256 and the backflow prevention diode 258, the output voltage of the flyback regulator 702 reaches 180V before the first bypass switch is turned on. At that time, assuming that the average value (at room temperature) of Vf of the LED being used is 4V, and 3V when almost no current flows, the first bypass switch has a voltage of 180V-3V × 30 = 90V. Applied. Therefore, in any of the 30 bypass switches, although only a voltage of several volts is normally applied, an element having a withstand voltage of 100 V must be selected in consideration of disconnection and poor contact.

  Next, when disconnection or poor contact occurs in the wiring indicated by “x” indicated by reference numeral 764 in FIG. 10, when almost no current flows through the first electrostatic protection zener diode, the above 90 V is controlled. When current flows, 180V-4V × 30 = 60V is applied. Here, assuming that the Zener voltage of the first electrostatic protection Zener diode is 20V, 90V and 60V are voltages higher than 20V. Therefore, when the control current is 1A, the voltage is 20V × 1A for several tens of milliseconds to several hundreds of milliseconds. = 20 W is applied to the first electrostatic protection Zener diode, and it is necessary to select an element that can withstand this. In order to avoid this, the Zener voltage of the first electrostatic protection Zener diode may be set to 90 V or more, but it becomes difficult to play the role of the original electrostatic protection.

Therefore, the semiconductor light source control device 700 according to the present modification includes the first bypass / limiter circuit 250-1, so that the upper limit of the voltage applied to the first bypass switch 110-1 even if a contact failure or disconnection occurs. It is suppressed. Therefore, an element having a high breakdown voltage of 100 V or more may not be selected as the first bypass switch 110-1. Further, if the limiting voltage by the first bypass / limiter circuit 250-1 is set to be equal to or lower than the Zener voltage of the first electrostatic protection Zener diode 252-1, a small Zener diode can be selected.
In the semiconductor light source control device 100 according to the embodiment, the withstand voltage in the kV order is required for the bypass switch when there is no limiter function, and therefore the withstand voltage suppression effect by providing the limiter function is more remarkable in the embodiment. is there.

  In the embodiment, the case where the LEDs correspond to the bypass switches on a one-to-one basis has been described. However, the present invention is not limited to this. For example, when one bypass switch is connected to a series of two LEDs, the total maximum value of LED Vf = 12V and the Zener voltage of the electrostatic protection Zener diode = 40V. Therefore, the Zener voltage of the limiter Zener diode May be in the range of 9V to 21V. When the Zener voltage of the limiter Zener diode is set to 20V, the limit value of the voltage between both ends is 23V. Therefore, a 30V withstand voltage element may be selected as the bypass switch.

  In the embodiment, when not only the second bypass connection wiring 280-2 but also the “x” mark indicated by the reference numeral 220 (see FIG. 5) of the third bypass connection wiring 280-3 occurs. , The second bypass circuit 270-2, the third bypass circuit 270-3, the fourth bypass circuit 270-4, and the bypass drive circuit 112 cause all of the second bypass switch, the third bypass switch, and the fourth bypass switch to Forced on.

  In the embodiment, when a continuity failure occurs in the bypass connection wiring, which of the two bypass switches connected to the bypass connection wiring is forcibly turned on due to disconnection abnormality and which is forcibly turned on due to short circuit abnormality. Although the case where it depends on the characteristic of a bypass switch etc. was demonstrated, it is not restricted to this. For example, a resistor may be provided in parallel with each of the odd-numbered bypass circuits (first bypass circuit 270-1, third bypass circuit 270-3,...). Alternatively, a resistor may be provided in parallel with each of the even-numbered bypass circuits. In these cases, when a conduction failure occurs in the bypass connection wiring, the bypass switch on the side where such a resistor is provided is forcibly turned on due to a short circuit abnormality, and the side on which such a resistor is not provided The bypass switch is forcibly turned on due to disconnection abnormality.

  In the embodiment, the case where the disconnection detection function and the short-circuit detection function of the bypass circuit and the bypass drive circuit are used to cope with the conduction failure of the bypass connection wiring has been described. However, the present invention is not limited to this. For example, a means for detecting a continuity failure of the bypass connection wiring, such as a current measuring means, is provided in the bypass connection wiring, and it is determined whether or not a continuity failure has occurred in the bypass connection wiring using the means. May be. When it is determined that a conduction failure has occurred, the bypass drive circuit may forcibly turn on the corresponding two bypass switches.

  In the embodiment, the case has been described in which the integration capacitor is charged when a conduction failure or a short circuit abnormality is detected when the bypass switch is off, and in other cases, the integration capacitor is discharged. However, the present invention is not limited to this. For example, the integration capacitor may be discharged with a first time constant when a continuity failure or a short circuit abnormality is detected when the bypass switch is off, and the integration capacitor may be charged with a second time constant otherwise.

  In the semiconductor light source control device 100 according to the embodiment, a PWM dimming function is employed. During PWM dimming, the bypass drive circuit 112 periodically changes the voltage level of the lighting control signal at the dimming frequency f1. If no abnormality has occurred, the corresponding bypass switch is turned off when the lighting control signal is at a low level, and the corresponding bypass switch is turned on when the lighting control signal is at a high level. Therefore, the low level of the lighting control signal is a state corresponding to turning off the corresponding bypass switch, and the high level of the lighting control signal is a state corresponding to turning on of the corresponding bypass switch.

  The bypass drive circuit 112 is a microcomputer and operates at a relatively long time interval. In particular, the bypass drive circuit 112 determines whether or not an abnormality has occurred for each LED by monitoring an abnormality detection signal over a predetermined abnormality determination period. When it is determined that an abnormality has occurred in a certain LED, the bypass drive circuit 112 fixes the lighting control signal corresponding to that LED at a high level. If this is explained based on the concept of abnormality confirmation / unconfirmation, it can be said that when the bypass drive circuit 112 detects an abnormal state continuously during the abnormality determination period, it is determined as abnormal and the bypass switch is on-latch controlled.

  In general, the abnormality determination period is longer than the PWM dimming period (= 1 / f1). Therefore, the semiconductor light source control device 100 according to the embodiment introduces an integration circuit, so that even when PWM dimming is performed, an error occurs. Enables more accurate detection of short circuit and disconnection errors with less detection.

  Consider a case in which a disconnection abnormality occurs in LED constant lighting (bypass switch = always off) control without PWM dimming. By the action of the limiter Zener diode of the bypass / limiter circuit corresponding to the LED until the bypass switch corresponding to the LED is on-latch controlled after the disconnection of the LED occurs, the bypass switch realizes the limiter function. It works as a switch to do. Thereby, the path | route of the electric current to other LED can be maintained. However, in this case, since the bypass switch basically operates in the linear region, power of “disconnection detection voltage × drive current Iout” is continuously applied to the bypass switch, and power loss in the bypass switch increases. As a result, a large-sized switch element having a large withstand capacity is required as a bypass switch, which may increase the size and cost of the circuit. Here, if the bypass switch is on-latch controlled immediately after detection of the disconnection abnormality, the power loss is reduced. However, in this case, the number of determinations in the bypass drive circuit 112 is small (the determination time is short), particularly in the PWM dimming mode. The possibility of erroneous detection (malfunction) of disconnection abnormality increases.

Even if a disconnection occurs during PWM dimming, the bypass switch operates as a switch for realizing the limiter function when the lighting control signal is at a low level. There is concern about an increase in loss. The ratio of the on-period of the bypass switch (= LED extinguishing period) in one cycle (= 1 / f1) of PWM dimming is referred to as an on-duty ratio. The power loss due to disconnection increases as the on-duty ratio decreases.
Further, the power loss in the bypass switch when the bypass switch is turned on is small mainly due to the on-resistance of the bypass switch, and is much smaller than the power loss particularly when operating in the linear region.

  FIG. 11 is a circuit diagram showing a configuration of the second bypass circuit 870-2 of the semiconductor light source control device according to the fifth modification. Other bypass circuits are configured similarly to the second bypass circuit 870-2.

The difference between the second bypass circuit 270-2 shown in FIG. 5 and the second bypass circuit 870-2 shown in FIG. 11 is mainly the configuration of the second switch level shift circuit. The second switch level shift circuit 854-2 of the second bypass circuit 870-2 receives the second lighting on / off control signal Sc2 'from the bypass drive circuit 812 and converts it into the second bypass switch drive signal Sd2. In particular, the second switch level shift circuit 854-2 sets the second bypass switch drive signal Sd2 to the low level when the second lighting control signal Sc2 ′ is at the high level, and sets the second lighting control signal Sc2 ′ to the low level. At this time, the second bypass switch drive signal Sd2 is set to the high level.
Note that the second bypass / limiter circuit 850-2 of the second bypass circuit 870-2 does not have to have a resistance corresponding to the sixteenth resistor 260.

The second switch level shift circuit 854-2 includes a 26th resistor 822, a 27th resistor 814, a 28th resistor 816, a 29th resistor 818, and a fifth pnp type bipolar transistor 820. One end of the 26th resistor 822 is connected to a terminal of the bypass drive circuit 812. The second lighting on / off control signal Sc2 ′ is output from the terminal of the bypass drive circuit 812. The other end of the 26th resistor 822 is connected to the base of the fifth pnp bipolar transistor 820. One end of the 27th resistor 814 is connected to a connection node between the other end of the 26th resistor 822 and the base of the fifth pnp bipolar transistor 820. A control power supply voltage Vcc is applied to the other end of the 27th resistor 814 and the emitter of the fifth pnp type bipolar transistor 820. The collector of the fifth pnp bipolar transistor 820 is connected to one end of the 28th resistor 816. The other end of the 28th resistor 816 is connected to one end of the 29th resistor 818. The other end of the 29th resistor 818 is connected to the cathode of the second LED 2-2.
A connection node between the other end of the 28th resistor 816 and one end of the 29th resistor 818 is connected to the gate of the second bypass switch 110-2. A signal generated at the connection node is the second bypass switch drive signal Sd2.

  When the second lighting on / off control signal Sc2 'is at a high level, the second bypass switch drive signal Sd2 is at a low level, and the second bypass switch 110-2 is turned off (the second LED 2-2 is lit). When the second lighting on / off control signal Sc2 'is at a low level, the second bypass switch drive signal Sd2 is at a high level and the second bypass switch 110-2 is turned on (the second LED 2-2 is turned off). When PWM dimming is performed, the bypass drive circuit 812 uses the second lighting control signal Sc2 'as a high level / low level repetition signal with a period of several milliseconds.

  When the voltage across the integration capacitor 282 exceeds a predetermined threshold voltage, the bypass drive circuit 812 applies to the control input terminal or gate of the second bypass switch 110-2 in synchronization with the second turn-on / off control signal Sc2 ′. The on-duty ratio and period of the applied signal are increased from those of normal PWM dimming. In particular, when the voltage of the second abnormality detection signal Sdet2 exceeds the level threshold value Vg, the bypass drive circuit 812 starts an abnormality determination period for the second LED 2-2 and occupies one cycle of the second lighting on / off control signal Sc2 ′. Increase the percentage of low level duration. In one example, the cycle of the second lighting on / off control signal Sc2 'increases to several hundred milliseconds, and the on-duty ratio increases to 90% or more.

  FIG. 12 is a timing chart showing changes in the second lighting on / off control signals Sc2 and Sc2 'when the second LED 2-2 is disconnected. The waveform shown in the upper part of FIG. 12 represents the second lighting control signal Sc2 ′ when the semiconductor light source control device according to the fifth modified example performs PWM dimming with the first period PT1 of about several milliseconds at the time of normal lighting. Show. The waveform shown in the middle part of FIG. 12 represents the second lighting on / off control signal Sc2 when the semiconductor light source control device 100 according to the embodiment performs regular lighting during normal lighting. The waveform shown in the lower part of FIG. 12 indicates the second lighting on / off control signal Sc2 when the semiconductor light source control device 100 according to the embodiment performs PWM dimming with the first period PT1 during normal lighting.

  When the disconnection occurs in the second LED 2-2 in the fifth modification, the cycle of the second lighting on / off control signal Sc2 'becomes the second cycle PT2 of about several hundred milliseconds that is larger than the first cycle PT1. In particular, the bypass drive circuit 812 realizes such an increase in the period by extending the low-level duration while maintaining the high-level duration in one cycle of the second lighting on / off control signal Sc2 ′. To do. The second period PT2 may be smaller than the second time constant related to the discharge of the second integration circuit 278-2.

  In the abnormality determination period PT3 of about several seconds, the bypass drive circuit 812 determines that the second abnormality detection signal Sdet2 when the second lighting on / off control signal Sc2 ′ is at the high level is higher than the level threshold value Vg. The on-duty ratio and cycle of the signal applied to the gate of the second bypass switch 110-2 are kept increased in synchronization with the lighting / extinguishing control signal Sc2 ′. When the voltage of the second abnormality detection signal Sdet2 falls below the level threshold value Vg, the bypass drive circuit 812 returns the second lighting on / off control signal Sc2 'to a state having the first cycle PT1. If the state where the voltage of the second abnormality detection signal Sdet2 is higher than the level threshold value Vg continues until the abnormality determination period PT3 expires, the bypass drive circuit 812 determines that an abnormality has occurred (that is, determined abnormality), The second lighting on / off control signal Sc2 ′ is fixed at a low level.

  When the disconnection occurs in the second LED 2-2 in the embodiment, the state of the second lighting on / off control signal Sc2 remains the same as that before the occurrence of the abnormality during the abnormality determination period PT3. When the abnormality determination period PT3 expires, it is determined that an abnormality has occurred, and the second lighting on / off control signal Sc2 is fixed at a high level.

  As described above, according to the semiconductor light source control device according to the fifth modified example, the bypass switch OFF period in the abnormality determination period can be compared with the semiconductor light source control device 100 according to the embodiment in both the constant lighting and the PWM dimming. The total length can be shortened. Since this off period corresponds to a period in which the bypass switch operates in the linear region when disconnection occurs, according to the fifth modification, power loss in the bypass switch can be reduced. As a result, a low-resistance switch element can be employed as the bypass switch, and the circuit can be reduced in size and cost.

  Especially in a situation where it is not appropriate to determine that an abnormality has occurred immediately after the transition of the level of the abnormality detection signal for the convenience of performing PWM dimming, according to the fifth modification, by monitoring the abnormality detection signal over the abnormality determination period, It is possible to more accurately determine whether or not an abnormality has occurred, and to reduce power loss of the bypass switch during the abnormality determination period.

  6 on-vehicle battery, 8 power switch, 100 semiconductor light source control device, 102 flyback regulator, 104 down converter, 106 control circuit, 108 current detection resistor, 128 output capacitor, 142 flywheel diode, 144 inductor.

Claims (7)

A drive circuit for generating a drive current flowing through a plurality of semiconductor light sources connected in series;
A first bypass switch connected in parallel with a part of the plurality of semiconductor light sources;
A second bypass switch connected in series with the first bypass switch and in parallel with another part of the plurality of semiconductor light sources,
A connection for connecting a connection node between the first bypass switch and the second bypass switch, and a connection node between a part of the plurality of semiconductor light sources and another part of the plurality of semiconductor light sources. Regarding the wiring, the polarity of the current flowing through the connection wiring when the first bypass switch is off and the second bypass switch is on is the same as the polarity when the first bypass switch is on and the second bypass switch is off. It is opposite to the polarity of the current flowing in the connection wiring,
The light source control device according to claim 1, wherein both of the first bypass switch and the second bypass switch are forcibly turned on when a continuity failure occurs in the connection wiring. The voltage across the first bypass switch when the first bypass switch is off is lower than the first voltage for short circuit detection or higher than the second voltage for disconnection detection higher than the first voltage. The first bypass switch is forcibly turned on, and the voltage across the second bypass switch when the second bypass switch is off is lower than the third voltage for short detection or the third 2. The light source control device according to claim 1, further comprising a switch control circuit that forcibly turns on the second bypass switch when the voltage is higher than a fourth voltage for detecting disconnection higher than the voltage. The plurality of semiconductor light sources are a plurality of light emitting diodes;
Each of the second voltage and the fourth voltage is a forward voltage drop defined by a part of the plurality of semiconductor light sources and a forward direction defined by another part of the plurality of semiconductor light sources. The light source control device according to claim 2, wherein the light source control device is set to be lower than a sum of the drop voltage. The switch control circuit includes:
A main control circuit for periodically turning on and off the first bypass switch during normal lighting;
Has a capacitor, is held in (i) the case where the first bypass switch is higher than the voltage across the first bypass switch is lower than the first voltage or the second voltage when turned off, the capacitor (Ii) the voltage across the first bypass switch when the first bypass switch is off is higher than the first voltage and the second voltage is changed in the first direction with a first time constant . When the voltage is lower than the voltage, or when the first bypass switch is on, the amount of charge held in the capacitor is set to a second time constant longer than the first time constant and opposite to the first direction. An abnormality detection auxiliary circuit that changes in two directions,
4. The main control circuit according to claim 2, wherein the main control circuit determines whether to forcibly turn on the first bypass switch according to a comparison result between a voltage across the capacitor and a threshold value. Light source control device. A drive circuit for generating a drive current flowing through a plurality of semiconductor light sources connected in series;
A bypass switch connected in parallel with at least some of the plurality of semiconductor light sources;
A main control circuit that periodically turns on and off the bypass switch during normal lighting;
And (i) a second voltage for detecting disconnection when the voltage across the bypass switch when the bypass switch is off is lower than the first voltage for short detection or higher than the first voltage. is higher than the amount of charge held in the capacitor is changed to the first direction at a first time constant, than the voltage across the bypass switch is the first voltage when the (ii) the bypass switch is turned off Higher and lower than the second voltage, or when the bypass switch is on, the amount of charge held in the capacitor is set to the first direction with a second time constant longer than the first time constant. Comprises an anomaly detection auxiliary circuit that changes in the opposite second direction,
The main control circuit determines whether a short circuit or disconnection abnormality has occurred based on the voltage across the capacitor, and forcibly turns on the bypass switch when it is determined that the abnormality has occurred. A light source control device characterized by the above. A drive circuit for generating a drive current flowing through a plurality of semiconductor light sources connected in series;
A bypass switch connected in parallel with at least a part of the plurality of semiconductor light sources and controlled to be turned on and off by a control signal;
The control signal periodically repeats a state corresponding to turning on the bypass switch and a state corresponding to turning off the bypass switch during normal lighting,
The light source control device further includes
A limiter configured to limit the upper limit of the voltage across at least some of the plurality of semiconductor light sources using the bypass switch when a control signal is in a state corresponding to turning off the bypass switch. Circuit,
Includes a capacitor, when the limiter circuit limits the upper limit of the voltage across, the amount of charge held in the capacitor is changed to the first direction at a first time constant, the limiter circuit to limit the voltage across An abnormality detection auxiliary circuit that, when not limited , changes the amount of electric charge held in the capacitor in a second direction opposite to the first direction with a second time constant longer than the first time constant;
When the ratio of the duration of the state corresponding to the on-state of the bypass switch in one cycle of the control signal is referred to as an on-duty ratio, the on-duty ratio of the control signal when the voltage across the capacitor crosses a predetermined threshold voltage A main control circuit for increasing
The main control circuit determines whether an abnormality has occurred by monitoring the voltage across the capacitor over an abnormality determination period longer than the cycle of the control signal, and if it is determined that an abnormality has occurred, A light source control device, wherein the state of the control signal is fixed to a state corresponding to the on state of the bypass switch.   The light source control device according to claim 6, wherein the main control circuit increases a cycle of a control signal when a voltage across the capacitor crosses the threshold voltage.

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