JP6134312B2 - Downconverter control circuit and vehicle lamp - Google Patents

Description

  The present invention relates to a light source control device that controls a light source.

  2. Description of the Related Art In recent years, semiconductor light sources such as LEDs (Light Emitting Diodes) that have a longer life and lower power consumption have been used in vehicular lamps such as headlamps in place of conventional halogen lamps having filaments. Since the degree of light emission, that is, the brightness of the LED depends on the magnitude of the current flowing through the LED, a lighting circuit for adjusting the current flowing through the LED is required when the LED is used as a light source.

  The present applicant proposes a technique in Patent Document 1 that employs an array of LEDs as a light source and individually turns on and off each LED in order to make the light distribution of the headlamp variable and perform fine light distribution control. doing. In the lighting circuit described in Patent Document 1, a bypass switch is provided in parallel with each LED, and the individual lighting / extinguishing of the LED is realized by turning on and off the bypass switch.

JP2011-192865A

  However, in the bypass method described in Patent Document 1, the load varies when each bypass switch is switched on and off. Depending on the response speed of the drive circuit that generates the current to be supplied to the LED, the current may deviate from the target value due to load fluctuations.

  Such a problem is not limited to the bypass method, and may occur for any light source control device whose load may vary.

  The present invention has been made in view of such circumstances, and an object of the present invention is to provide a light source control device with better response to load fluctuations.

  One embodiment of the present invention relates to a light source control device. This light source control device includes a switching regulator that converts an input voltage into a target voltage, a flywheel diode that is connected in parallel with an output capacitor of the switching regulator, a light source to be driven from one end of the output capacitor, and a light source to an output capacitor. A switching element provided on the path of the current returning to the other end of the switch, provided between the output capacitor and the flywheel diode, and provided between the flywheel diode and the light source. And an inductor provided.

  According to this aspect, the current flowing through the light source to be driven passes through the inductor.

  It should be noted that any combination of the above-described constituent elements and those obtained by replacing the constituent elements and expressions of the present invention with each other among apparatuses, methods, systems, etc. are also effective as an aspect of the present invention.

  ADVANTAGE OF THE INVENTION According to this invention, the responsiveness of the light source control apparatus with respect to the fluctuation | variation of load can be improved.

It is a circuit diagram which shows the structure of the semiconductor light source control apparatus which concerns on embodiment, and the member connected to it. FIG. 2 is a circuit diagram showing a configuration of a hysteresis width setting circuit of FIG. 1. It is a graph which shows the relationship between the absolute value of a drive voltage, and an offset voltage. FIG. 2 is a circuit diagram showing a configuration of a down converter drive circuit in FIG. 1. FIG. 5A to FIG. 5C are graphs showing the time change of the drive current. It is a circuit diagram which shows the structure of the semiconductor light source lighting circuit which concerns on a comparative example. FIGS. 7A to 7C are circuit diagrams showing configurations of semiconductor light source control devices according to the first, second, and third modifications.

  Hereinafter, the same or equivalent components, members, and signals shown in the respective drawings are denoted by the same reference numerals, and repeated description thereof will be omitted as appropriate. In addition, in the drawings, some of the members that are not important for explanation are omitted. Moreover, the code | symbol attached | subjected to the voltage, electric current, or resistance may be used as what represents each voltage value, electric current value, or resistance value as needed.

  In this specification, “the state in which the member A is connected to the member B” means that the member A and the member B are electrically connected in addition to the case where the member A and the member B are physically directly connected. It includes the case of being indirectly connected through another member that does not affect the connection state. Similarly, “the state in which the member C is provided between the member A and the member B” refers to the case where the member A and the member C or the member B and the member C are directly connected, as well as an electrical condition. It includes the case of being indirectly connected through another member that does not affect the connection state.

  The semiconductor light source control device according to the embodiment receives a DC input voltage from a DC power source such as an in-vehicle battery, and DC / DC converts it to a DC target voltage using a switching regulator. This target voltage becomes substantially constant in the vicinity of a preset set voltage and is relatively independent of the input voltage. The semiconductor light source control device generates a drive voltage to be applied to a plurality of LEDs to be driven and a drive current to be passed to the plurality of LEDs by down-converting a substantially constant target voltage in this way. The down converter does not have an output capacitor, and instead, hysteresis control of the drive current is performed. Thereby, even when the on / off of the LEDs is individually switched, it is possible to suppress an undesired change in the drive current due to charging / discharging of the output capacitor. In particular, when the LEDs are individually switched on and off, changes in the drive current caused by the output capacitor of the switching regulator are smoothed by the inductor of the downstream down converter.

  FIG. 1 is a circuit diagram showing a configuration of a semiconductor light source control device 100 according to an embodiment and members connected thereto. The semiconductor light source control device 100 supplies a drive current Iout to a plurality (N) of in-vehicle LEDs 2-1 to 2-N connected in series and lights them. N is a natural number of 2 or more. The semiconductor light source control device 100 and the plurality of LEDs 2-1 to 2-N are mounted on a vehicle lamp such as a headlight. The semiconductor light source control device 100 is connected to the in-vehicle battery 6 and the power switch 8.

  The in-vehicle battery 6 generates a DC battery voltage (power supply voltage) Vbat of 12V (or 24V). The power switch 8 is a relay switch provided to control the entire on / off of the plurality of LEDs 2-1 to 2 -N, and is provided in series with the in-vehicle battery 6. When the power switch 8 is turned on, the battery voltage Vbat is supplied from the positive terminal of the in-vehicle battery 6 to the semiconductor light source control device 100 as an input voltage. The negative terminal of the in-vehicle battery 6 is connected to the fixed voltage terminal, that is, grounded.

  The semiconductor light source control device 100 includes a switching regulator or flyback regulator 102, a down converter 104, a control circuit 106, a current detection resistor 108, a plurality of bypass switches 110-1 to 110-N, and a bypass drive circuit 112. . Control circuit 106 controls flyback regulator 102 and downconverter 104, and includes flyback drive circuit 134, downconverter drive circuit 136, and hysteresis width setting circuit 138.

  The flyback regulator 102 is a voltage regulator, and converts the input battery voltage Vbat into a target voltage Vt and outputs it. Since the output terminal on the high potential side of the flyback regulator 102 is grounded, the target voltage Vt is a voltage applied to the output terminal on the low potential side of the flyback regulator 102 and has a negative polarity. The flyback regulator 102 includes an input capacitor 114, a first switching element 116, an input transformer 124, an output diode 126, an output capacitor 128, a voltage detection diode 130, and a voltage detection capacitor 132.

  The input capacitor 114 is provided in parallel with the in-vehicle battery 6 and smoothes the battery voltage Vbat. More specifically, the input capacitor 114 is provided in the vicinity of the input transformer 124 and fulfills a voltage smoothing function for the switching operation of the flyback regulator 102.

  The primary winding 118 of the input transformer 124 and the first switching element 116 are connected in series, and the series circuit is connected in parallel with the input capacitor 114 to the in-vehicle battery 6. For example, the first switching element 116 is composed of an N-channel MOSFET (Metal Oxide Semiconductor Field Effect Transistor). One end of the secondary winding 120 of the input transformer 124 is connected to one end of the output capacitor 128, and the other end of the secondary winding 120 is connected to the anode of the output diode 126. The other end of the output capacitor 128 is connected to the cathode of the output diode 126. One end of the output capacitor 128 is connected to the output terminal on the low potential side of the flyback regulator 102, and the target voltage Vt is applied. The other end of the output capacitor 128 is connected to the output terminal on the high potential side of the flyback regulator 102.

  To the control terminal (gate) of the first switching element 116, a rectangular wave-form front-stage control signal S1 generated by the flyback drive circuit 134 is applied. The first switching element 116 is turned on when the pre-stage control signal S1 is asserted, that is, when it is at a high level, and turned off when it is negated, that is, when it is at a low level.

  The voltage detection winding 122, the voltage detection diode 130, and the voltage detection capacitor 132 of the input transformer 124 constitute a positive voltage detection circuit for detecting the magnitude of the target voltage Vt as a positive voltage. One end of the voltage detection winding 122 is grounded, and the other end is connected to the anode of the voltage detection diode 130. The cathode of voltage detection diode 130 is connected to one end of voltage detection capacitor 132. The other end of the voltage detection capacitor 132 is grounded. A positive voltage corresponding to the absolute value of the target voltage Vt is applied to one end of the voltage detection capacitor 132. This voltage is supplied to the flyback drive circuit 134 as the detection voltage Vd.

  The flyback drive circuit 134 performs voltage feedback control for keeping the target voltage Vt substantially constant based on the detection voltage Vd. The flyback drive circuit 134 adjusts the frequency and duty ratio of the pre-stage control signal S1 so that the target voltage Vt approaches a set voltage of, for example, about −100V.

  The down converter 104 is provided in the subsequent stage of the flyback regulator 102 and includes a second switching element 140, a flywheel diode 142, and an inductor 144, but does not include an output voltage smoothing capacitor.

  The second switching element 140 is composed of, for example, an N channel MOSFET. The control terminal of the second switching element 140 is applied with a rectangular-wave subsequent control signal S2 generated by the down converter drive circuit 136. The second switching element 140 is turned on when the post-stage control signal S2 is at a high level, and turned off when it is at a low level. The drain of the second switching element 140 is connected to the output terminal on the high potential side of the output capacitor 128, that is, the high potential side of the flyback regulator 102. The source of the second switching element 140 is connected to the cathode of the flywheel diode 142.

  The anode of the flywheel diode 142 is connected to one end of the inductor 144. A connection node between the anode of the flywheel diode 142 and one end of the inductor 144 is connected to the low potential side of the output capacitor 128, that is, the low potential side output terminal of the flyback regulator 102. The other end of the inductor 144 is connected to the cathode side of the plurality of LEDs 2-1 to 2-N.

  The current detection resistor 108 is provided on the path of the drive current Iout. One end of the current detection resistor 108 is connected to a connection node between the source of the second switching element 140 and the cathode of the flywheel diode 142. The other end of the current detection resistor 108 is grounded and connected to the anode side of the plurality of LEDs 2-1 to 2-N. A voltage drop Vm proportional to the drive current Iout is generated in the current detection resistor 108.

  Since the anode sides of the plurality of LEDs 2-1 to 2-N are grounded, the negative drive voltage Vout is applied to the cathode side of the plurality of LEDs 2-1 to 2-N, that is, the other end of the inductor 144. During normal lighting, the drive voltage Vout is a negative voltage having a magnitude corresponding to the number of LEDs in a light emitting state (= the corresponding bypass switch is off) × the forward drop voltage of one LED.

  The down converter drive circuit 136 performs current feedback control for keeping the drive current Iout within a predetermined current range based on the voltage drop Vm. The down-converter drive circuit 136 turns off the second switching element 140 when the magnitude of the drive current Iout exceeds a predetermined current upper limit value Ith1, and the current lower limit value Ith2 whose magnitude of the drive current Iout is smaller than the current upper limit value Ith1. If it falls below, the second switching element 140 is turned on. The down converter drive circuit 136 sets the post-stage control signal S2 to a low level when the magnitude of the drive current Iout exceeds the current upper limit value Ith1, and sets the post-stage control signal S2 to high when the magnitude of the drive current Iout falls below the current lower limit value Ith2. Level.

  The semiconductor light source control device 100 is configured to be able to individually control turning on / off of the plurality of LEDs 2-1 to 2-N. Each of the plurality of bypass switches 110-1 to 110-N is connected in parallel with the corresponding LED, and is configured by, for example, an N-channel MOSFET. The bypass drive circuit 112 supplies a control signal to the control terminals of the bypass switches 110-1 to 110-N. The bypass drive circuit 112 individually controls the level of each control signal so as to obtain a desired luminance and light distribution pattern.

  The hysteresis width setting circuit 138 sets a hysteresis width ΔI that is the difference between the current upper limit value Ith1 and the current lower limit value Ith2 based on the drive voltage Vout. The hysteresis width setting circuit 138 increases the hysteresis width ΔI as the absolute value of the drive voltage Vout increases when the absolute value of the drive voltage Vout falls below the voltage threshold Vth that is smaller than the absolute value of the target voltage Vt. When the absolute value of the voltage Vout exceeds the voltage threshold value Vth, the hysteresis width ΔI is decreased as the absolute value of the drive voltage Vout increases.

  FIG. 2 is a circuit diagram showing a configuration of the hysteresis width setting circuit 138. The hysteresis width setting circuit 138 includes a first operational amplifier 146, a first diode 148, a first resistor 150, a second resistor 152, a third resistor 154, a fourth resistor 156, a fifth resistor 158, A reference voltage source 160. A control power supply voltage Vcc is applied to one end of the third resistor 154. The other end of the third resistor 154 is connected to one end of the second resistor 152, one end of the fifth resistor 158, and one end of the fourth resistor 156. The other end of the fourth resistor 156 is grounded. The drive voltage Vout is applied to the other end of the fifth resistor 158. The other end of the second resistor 152 is connected to the inverting input terminal of the first operational amplifier 146. The inverting input terminal of the first operational amplifier 146 is connected to the anode of the first diode 148 through the first resistor 150. The cathode of the first diode 148 is connected to the output terminal of the first operational amplifier 146. A reference voltage Vref generated by the reference voltage source 160 is applied to the non-inverting input terminal of the first operational amplifier 146. A voltage applied to the anode of the first diode 148 is referred to as an offset voltage Voffset. As will be described later, the offset voltage Voffset corresponds to the hysteresis width ΔI, and the hysteresis width ΔI increases as the offset voltage Voffset increases.

  The resistance value around the first operational amplifier 146 is compared with the values of the third resistor 154, the fourth resistor 156, and the fifth resistor 158 that are differential with the reference voltage Vref, and the first resistor 150 that determines the amplification factor. The value of the second resistor 152 is sufficiently increased so that the feedback current does not affect the differential with the reference voltage Vref.

  FIG. 3 is a graph showing the relationship between the absolute value of the drive voltage Vout and the offset voltage Voffset. When the absolute value of the negative drive voltage Vout is small, the voltage at the common connection node of the third resistor 154, the fourth resistor 156, and the fifth resistor 158 is larger than the reference voltage Vref, and thus the first operational amplifier 146 is The offset voltage Voffset is reduced by sinking the current. The offset voltage Voffset becomes maximum when the voltage of the common connection node becomes equal to the reference voltage Vref.

  In order to realize the control that the hysteresis width ΔI, that is, the offset voltage Voffset is maximized when the absolute value of the drive voltage Vout becomes the voltage threshold Vth, the absolute value of the drive voltage Vout is set to the reference voltage Vref. It is set to the voltage of the common connection node when it is equal to the threshold value Vth. In particular, when the set voltage of the flyback regulator 102 is −100V, the reference voltage Vref is set to the voltage of the common connection node when the drive voltage Vout = −Vth = −50V.

  When the absolute value of the drive voltage Vout increases beyond the voltage threshold value Vth, the operation of the first operational amplifier 146 is lost, and the voltage of the common connection node becomes the offset voltage Voffset as it is. The hysteresis width setting circuit 138 controls the hysteresis width ΔI by sending the offset voltage Voffset that changes in a mountain shape with respect to the absolute value of the drive voltage Vout to the down converter drive circuit 136 in this way, and the switching frequency of the down converter 104 Is within a predetermined range.

  FIG. 4 is a circuit diagram showing a configuration of down converter drive circuit 136. The down converter drive circuit 136 includes a second operational amplifier 162, a comparator 164, a gate driver 166, a first current mirror circuit 170, a seventh resistor 172, an eighth resistor 174, a tenth resistor 178, A twelfth resistor 182, a thirteenth resistor 184, a first npn bipolar transistor 190, a third switching element 202, a fourth switching element 204, and a second current mirror circuit 206 are included.

  The offset voltage Voffset is applied to the non-inverting input terminal of the second operational amplifier 162. The output terminal of the second operational amplifier 162 is connected to the base of the first npn type bipolar transistor 190, and the inverting input terminal is connected to the emitter of the first npn type bipolar transistor 190. One end of the eighth resistor 174 is connected to the emitter of the first npn-type bipolar transistor 190, and the other end is grounded. The collector of the first npn-type bipolar transistor 190 is connected to the first current mirror circuit 170 via the seventh resistor 172.

  The first current mirror circuit 170 includes a sixth resistor 168, a ninth resistor 176, an eleventh resistor 180, a first pnp bipolar transistor 192, a second pnp bipolar transistor 194, a third pnp bipolar transistor 196, Have These circuit elements are connected to each other so as to form a known current mirror circuit. The first current mirror circuit 170 receives the current flowing through the seventh resistor 172, receives the current flowing through the tenth resistor 178, and the current flowing through the third switching element 202 as an output, and the magnitude of the input current and the magnitude of the output current are substantially reduced. Make equal.

  The second current mirror circuit 206 includes a fourteenth resistor 186, a fifteenth resistor 188, a second npn-type bipolar transistor 198, and a third npn-type bipolar transistor 200. These circuit elements are connected to each other so as to form a known current mirror circuit. The second current mirror circuit 206 receives the current flowing through the tenth resistor 178 as an input and the current flowing through the fourth switching element 204 as an output, and makes the magnitude of the input current and the magnitude of the output current substantially equal.

  The third switching element 202 is composed of, for example, a P-channel MOSFET. The fourth switching element 204 is composed of, for example, an N-channel MOSFET. The source of the third switching element 202 is connected to the first current mirror circuit 170. The gate of the third switching element 202 is connected to the inverting output terminal of the comparator 164. The drain of the third switching element 202 is connected to the drain of the fourth switching element 204. The gate of the fourth switching element 204 is connected to the inverting output terminal of the comparator 164. The source of the fourth switching element 204 is connected to the second current mirror circuit 206.

  The twelfth resistor 182 and the thirteenth resistor 184 are connected in series between the control power supply voltage Vcc and the ground potential in this order. A connection node between the twelfth resistor 182 and the thirteenth resistor 184 is connected to a connection node between the drain of the third switching element 202 and the drain of the fourth switching element 204. A connection node between the drain of the third switching element 202 and the drain of the fourth switching element 204 is connected to the non-inverting input terminal of the comparator 164. A voltage drop Vm is applied to the inverting input terminal of the comparator 164.

  The non-inverting output terminal of the comparator 164 is connected to the gate driver 166. The gate driver 166 aligns the phase of the post-stage control signal S2 with the phase of the signal appearing at the non-inverting output terminal of the comparator 164. That is, when the signal appearing at the non-inverting output terminal of the comparator 164 becomes a high level (low level), the gate driver 166 sets the subsequent stage control signal S2 to a high level (low level).

  The second operational amplifier 162 and the first npn-type bipolar transistor 190 that receive the offset voltage Voffset output a current of Voffset / (resistance value of the eighth resistor 174). This current is sinked or sourced to the voltage dividing node of the twelfth resistor 182 and the thirteenth resistor 184 depending on the phase of the output of the comparator 164 that receives the voltage drop Vm. Regarding the third switching element 202 and the fourth switching element 204 in the bridge form, the third switching element 202 is turned on at the timing when the gate of the second switching element 140 is at the high level (the second switching element 140 is turned on), and the twelfth resistance The voltage at the voltage dividing node between 182 and the thirteenth resistor 184 increases, and the current upper limit value Ith1 is set. When the drive current Iout increases and reaches the current upper limit value Ith1, the fourth switching element 204 is turned on substantially simultaneously with the gate of the second switching element 140 becoming low level (the second switching element 140 is turned off). . As a result, the voltage at the voltage dividing node between the twelfth resistor 182 and the thirteenth resistor 184 decreases, and the current lower limit value Ith2 is set.

  The average value of the drive current Iout is set by the divided voltage of the twelfth resistor 182 and the thirteenth resistor 184. Further, due to the action of the hysteresis width setting circuit 138, the sink / source current increases when the absolute value of the drive voltage Vout approaches the voltage threshold value Vth. Therefore, the current upper limit value Ith1−current lower limit value Ith2 = the hysteresis width ΔI is large. Become. The hysteresis width ΔI decreases as the absolute value of the drive voltage Vout increases from the voltage threshold Vth. As will be described later, this acts to keep the switching frequency of the down converter 104 within a predetermined range.

The operation of the semiconductor light source control device 100 having the above configuration will be described.
FIGS. 5A to 5C are graphs showing a change with time of the drive current Iout. Consider a situation where only one LED is lit, then about half are lit, and then all LEDs are lit. FIG. 5A shows the time change of the drive current Iout when only one LED is turned on and the remaining N−1 LEDs are turned off by turning on the corresponding bypass switch. FIG. 5B shows the change over time in the drive current Iout when about half, that is, N / 2 LEDs are turned on and the remaining LEDs are turned off. FIG.5 (c) shows the time change of the drive current Iout when all LED are lighted.

  FIGS. 5A to 5C show a case where the hysteresis width ΔI is adjusted so that the switching frequency of the second switching element 140, that is, the switching cycle Ts becomes substantially constant regardless of the number of turned on / off of LEDs. It is. However, in the present embodiment, the hysteresis width ΔI only needs to be controlled so that the change in the switching period Ts due to the change in the number of LEDs that are turned on and off can be controlled by those skilled in the art who have touched this specification. Understood.

  Referring to FIG. 5A, when the number of LEDs to be lit is small, the drive current Iout rises relatively quickly during the on time Ton of the second switching element 140, and the drive current during the off time Toff of the second switching element 140. Iout falls relatively slowly. The hysteresis width at this time is expressed as ΔI1. The absolute value of the drive voltage Vout is relatively low, and the offset voltage Voffset generated by the hysteresis width setting circuit 138 is also relatively low.

  Referring to FIG. 5B, when the number of LEDs to be turned on and the number of LEDs to be turned off are about the same number, the drive voltage Vout is about half of the set voltage of the flyback regulator 102, and the second switching element 140 The on time Ton and the off time Toff antagonize. The overall speed of change of the drive current Iout is greater than when the number of LEDs to be lit is small.

  The hysteresis width setting circuit 138 generates a higher offset voltage Voffset as shown in FIG. The down converter drive circuit 136 receives the high offset voltage Voffset, and makes the hysteresis width ΔI2 larger than the hysteresis width ΔI1 when the number of lighting LEDs is 1. As a result, the increase in the overall change speed of the drive current Iout is canceled out, and the switching period Ts is kept substantially constant.

  Referring to FIG. 5C, when the number of LEDs to be turned off is small or not, the drive current Iout rises relatively slowly during the on-time Ton of the second switching element 140, and the off-time Toff of the second switching element 140. The drive current Iout falls relatively quickly. The overall speed of change of the drive current Iout is smaller than when the number of LED lighting and the number of light extinguishing antagonize. The absolute value of the drive voltage Vout is relatively high, and the offset voltage Voffset generated by the hysteresis width setting circuit 138 is relatively low.

  The down-converter driving circuit 136 receives a low offset voltage Voffset, and makes the hysteresis width ΔI3 smaller than the hysteresis width ΔI2 when the number of LED lighting and the number of light-offs compete. As a result, the decrease in the overall change rate of the drive current Iout is canceled out, and the switching period Ts is kept substantially constant.

  In the semiconductor light source control device 100 according to the present embodiment, since the smoothing capacitor is not provided in the output stage to the plurality of LEDs 2-1 to 2-N, the followability of the drive current Iout with respect to the second switching element 140. Will be better. In particular, when the second switching element 140 is turned off, the drive current Iout decreases, and when the second switching element 140 is turned on, the drive current Iout increases. In order to stabilize the drive current Iout near the target value, hysteresis control of the drive current Iout is employed instead of smoothing. As a result, the response in the current feedback can be speeded up. For example, when the number of lighted LEDs changes due to the action of the bypass drive circuit 112 and the bypass switch, the drive current lout can be made to follow the load variation more quickly. In particular, it is possible to suppress undershoot of the drive current Iout when the number of lighting of the LED is increased and overshoot of the drive current Iout when the number of lighting is decreased.

In the semiconductor light source control device 100 according to the present embodiment, the flyback regulator 102 at the front stage has a negative output, and the down converter 104 at the rear stage also has a negative output. As a result, an N-channel MOSFET having better characteristics can be employed as the bypass switch.
In addition to the negative output, the inductor 144 is provided not between the cathode and the output of the flywheel diode 142 but between the anode and the output, so that the second switching element 140 of the down converter 104 has a better N characteristic. A channel MOSFET can be employed. Further, the drive voltage Vout can be detected stably.

  In addition, when the semiconductor light source control device has a positive output, the detection of the drive current is often performed on the high side in consideration of the case where the LED is grounded. Here, when the load changes, the potential at the detection location also changes, so that it is difficult to accurately detect the drive current. Also, the configuration of the detection circuit can be more complicated. Therefore, in the semiconductor light source control device 100 according to the present embodiment, the negative output is adopted, and the current detection resistor 108 is provided on the output on the positive side, that is, the ground side. As a result, even if the load (drive voltage Vout) changes, the change does not affect the potential of the detection location of the drive current Iout, and the drive current Iout can be detected stably. In addition, the configuration of the detection circuit can be simplified.

  When the drive current lout is subjected to hysteresis control, if the input voltage to the down converter 104 and / or the drive voltage Vout change, the slope of the rise or fall of the drive current lout changes, so the switching frequency of the second switching element 140 changes. Yes. Therefore, in the semiconductor light source control device 100 according to the present embodiment, the hysteresis width ΔI is adjusted so that the change of the switching frequency is suppressed. In particular, by setting the target switching frequency so as to avoid a known frequency band of radio noise, adverse effects on the semiconductor light source control device 100 due to radio noise can be suppressed.

  Moreover, in the semiconductor light source control device 100 according to the present embodiment, the fluctuation of the input voltage to the down converter 104 due to the fluctuation of the battery voltage Vbat is suppressed by the action of the flyback regulator 102. Therefore, a change in switching frequency due to a change in input voltage to the down converter 104 can be suppressed. In other words, it is not necessary to select the hysteresis width ΔI based on the combination of the input voltage to the down converter 104 and the drive voltage Vout, and it is only necessary to select the hysteresis width ΔI based mainly on the drive voltage Vout, so the hysteresis width ΔI is adjusted. The control for this is further simplified. This contributes to downsizing and speeding up of the control circuit.

  In the semiconductor light source control device 100 according to the present embodiment, an output capacitor 128 is provided at the output stage of the flyback regulator 102. If the second switching element 140 is turned on when the bypass switch is turned on, the charge stored in the output capacitor 128 tends to flow to the LED at once. However, since the semiconductor light source control device 100 is provided with the inductor 144 on the path of the drive current Iout, such a charge flow is smoothed, and the overshoot of the drive current Iout is suppressed. Similarly, when the bypass switch is turned off, the undershoot of the drive current Iout is suppressed.

Consider a semiconductor light source lighting circuit 300 according to the following comparative example, which was originally created to suppress overshoot and undershoot of the drive current Iout when the bypass switch is switched.
FIG. 6 is a circuit diagram showing a configuration of a semiconductor light source lighting circuit 300 according to a comparative example. The semiconductor light source lighting circuit 300 is basically a forward converter that does not use a smoothing capacitor. The semiconductor light source lighting circuit 300 includes a control circuit 302, an input capacitor 306, a reset circuit 308, a transformer 310, a fifth switching element 312, a second diode 314, a third diode 316, an inductor 318, a current And a detection resistor 320.
The control circuit 302 turns off the fifth switching element 312 when the magnitude of the drive current exceeds a predetermined current upper limit value, and turns on the fifth switching element 312 when the magnitude of the drive current falls below the current lower limit value.

For the semiconductor light source lighting circuit 300, the winding ratio of the transformer 310 is N _{s / p} , the inductance of the inductor 318 is Ls ′, the hysteresis width of the drive current is ΔI ′, the input voltage is Vin, the output voltage is Vout (<0), If the on-time of the fifth switching element 312 is Ton ′, the off-time is Toff ′, the switching frequency is F ′, and the forward drop voltage of the rectifier diode is small, so that it can be ignored, F ′ can be obtained from the following equation. .

In the semiconductor light source lighting circuit 300, when the winding ratio of the transformer 310 = 16.7 (input = 6V is converted to output = 100V), the inductance of the inductor 318 = 500 μH, and the hysteresis width = 0.1 A, Equation 1 The relationship between Vin, Vout, and F ′ obtained from the equation (1) is as shown in Table 1 below. Here, input voltage fluctuation = 6V to 20V, output (load) voltage fluctuation = −4V to −88V (22 LEDs in series with Vf = 4V) were assumed.

  In this case, the switching frequency F 'varies about 17 times at the maximum / minimum. If the inductance is increased, this fluctuation range can be suppressed, but the circuit becomes larger. In addition, when the function of suppressing the large fluctuation of the switching frequency F ′ from the input voltage and the output voltage and suppressing it to a predetermined range is realized, the control circuit scale becomes large.

The same calculation is performed for the semiconductor light source control apparatus 100 according to the present embodiment. Regarding the semiconductor light source control device 100, assuming that the inductance of the inductor 144 is Ls, the switching frequency is F, and the forward voltage drop of the flywheel diode 142 is small, F can be obtained from the following equation.

In the semiconductor light source control device 100, when the target voltage Vt = −100V, the inductance of the inductor 144 = 500 μH, and the hysteresis width = 0.1 A, the relationship between Vt, Vout, and F obtained from Equation 2 is as follows. It is as 2 tables.

  In this case, the fluctuation of the switching frequency F is suppressed to about 6.5 times. The main parameter causing the fluctuation is the drive voltage Vout and the target voltage Vt is substantially fixed. Therefore, the scale of the control circuit for adjusting the hysteresis width ΔI so as to suppress the fluctuation of the switching frequency F is relatively small. it can.

  Looking at the theoretical calculation value of the switching frequency F in Table 2, the switching frequency F increases as the drive voltage Vout decreases from −4V to −44V, and as the drive voltage Vout decreases from −44V to −88V. The switching frequency F decreases. The increase / decrease boundary of the switching frequency F is −50 V, which is about half of the output voltage of the first stage (front stage) flyback regulator 102 (the input voltage of the second stage (rear stage) down converter 104). Therefore, it is possible to easily keep the switching frequency F within a predetermined range by controlling so that the hysteresis width ΔI is increased as the drive voltage Vout is lower at Vout> −50V, and is decreased at Vout <−50V. It becomes.

  Further, in the present embodiment, it has been found that the boundary of the increase / decrease of the switching frequency F is at about half of the output voltage of the flyback regulator 102. However, in other embodiments involving other circuit arrangements, This boundary may be one third or one quarter of the output voltage. What can be said in common is that Vout that gives the maximum value of the switching frequency F when the hysteresis width is constant may exist between the maximum value and the minimum value of Vout. Therefore, if such a Vout is found through experiments or simulations, and the circuit is configured such that the hysteresis width ΔI is minimized at such Vout, fluctuations in the switching frequency F can be more suitably suppressed.

Ｖｏｆｆｓｅｔは、図２に示されるヒステリシス幅設定回路１３８の回路定数を調整し、Ｖｏｕｔ＝−５０Ｖ付近で図３に示されるグラフのように電圧値が高くなるようにしたものである。下限電圧・上限電圧は、図４に示されるダウンコンバータ駆動回路１３６の第１２抵抗１８２と第１３抵抗１８４との分圧ノードの電圧であり、電流下限値Ｉｔｈ２、電流上限値Ｉｔｈ１にそれぞれ対応する。下限電圧・上限電圧は、第８抵抗１７４、第１２抵抗１８２、第１３抵抗１８４の各抵抗値と制御電源電圧Ｖｃｃを設定し、オフセット電圧Ｖｏｆｆｓｅｔから算出したものである。平均電流は電流上限値Ｉｔｈ１と電流下限値Ｉｔｈ２との平均値である。スイッチング周波数は、式２と同じ算出式において、ΔＩ＝Ｉｔｈ１−Ｉｔｈ２、Ｖｔ＝−１００Ｖ、Ｌｓ＝２００μＨとして求めたものである。
Ｌｓを５００μＨから２００μＨへと小さくしても、スイッチング周波数を５５０ｋＨｚ強〜４００ｋＨｚ弱の範囲に収めることができることが分かる。すなわち、本実施の形態に係る半導体光源制御装置１００によると、駆動電流Ｉｏｕｔを平滑化するためのインダクタンスを小型化できる。A parameter setting example for the semiconductor light source control apparatus 100 according to the present embodiment is shown in Table 3 below.

Voffset is obtained by adjusting the circuit constant of the hysteresis width setting circuit 138 shown in FIG. 2 so that the voltage value becomes high as shown in the graph shown in FIG. 3 near Vout = −50V. The lower limit voltage and the upper limit voltage are voltages of voltage dividing nodes of the twelfth resistor 182 and the thirteenth resistor 184 of the down converter drive circuit 136 shown in FIG. 4, and correspond to the current lower limit value Ith2 and the current upper limit value Ith1, respectively. . The lower limit voltage and the upper limit voltage are calculated from the offset voltage Voffset by setting the resistance values of the eighth resistor 174, the twelfth resistor 182, and the thirteenth resistor 184 and the control power supply voltage Vcc. The average current is an average value of the current upper limit value Ith1 and the current lower limit value Ith2. The switching frequency is obtained in the same calculation formula as Expression 2 with ΔI = Ith1−Ith2, Vt = −100V, and Ls = 200 μH.
It can be seen that even if Ls is reduced from 500 μH to 200 μH, the switching frequency can be kept in the range of slightly higher than 550 kHz to slightly lower than 400 kHz. That is, according to the semiconductor light source control device 100 according to the present embodiment, the inductance for smoothing the drive current Iout can be reduced.

  Further, when comparing the semiconductor light source lighting circuit 300 according to the comparative example and the semiconductor light source control device 100 according to the present embodiment, the semiconductor light source control device 100 uses the output capacitor 128 of the flyback regulator 102 and the second switching of the down converter 104. Although the number of elements 140 is increased, the reset circuit 308 can be reduced from the semiconductor light source lighting circuit 300, so that the circuit scale is almost the same.

  The configuration and operation of the semiconductor light source control device according to the embodiment have been described above. This embodiment is an exemplification, and it is understood by those skilled in the art that various modifications can be made to each component and combination of processes, and such modifications are within the scope of the present invention.

  In the embodiment, the case where the second switching element 140 is arranged on the cathode side of the flywheel diode 142 and the inductor 144 is arranged on the anode side of the flywheel diode 142 as the element arrangement of the down converter 104 has been described. I can't. The flywheel diode may be connected in parallel with the output capacitor 128 of the flyback regulator 102. If the second switching element is provided on the path of the drive current Iout from one end of the output capacitor 128 to the LED and back from the LED to the other end of the output capacitor 128, the second switching element is provided between the output capacitor 128 and the flywheel diode. Good. On / off of the second switching element may be controlled based on the drive current. The inductor 144 may be provided on the path of the drive current Iout and provided between the flywheel diode and the LED.

  FIGS. 7A to 7C are circuit diagrams showing configurations of the semiconductor light source control devices 400, 500, and 600 according to the first, second, and third modifications. FIG. 7A shows a configuration of a semiconductor light source control device 400 according to the first modification. One end of the second switching element 440 is connected to the high potential side output of the flyback regulator 102, and the other end is connected to the cathode of the flywheel diode 442. One end of the inductor 444 is connected to a connection node between the other end of the second switching element 440 and the cathode of the flywheel diode 442. The other end of the inductor 444 is grounded and serves as a high potential side output terminal to the LED. The anode of the flywheel diode 442 is connected to the low potential side output of the flyback regulator 102 and serves as a low potential side output terminal to the LED.

  FIG. 7B shows a configuration of a semiconductor light source control device 500 according to the second modification. The cathode of the flywheel diode 542 is connected to the high potential side output of the flyback regulator 102 and grounded to form a high potential side output to the LED. One end of the second switching element 540 is connected to the low potential side output of the flyback regulator 102, and the other end is connected to the anode of the flywheel diode 542. One end of the inductor 544 is connected to a connection node between the other end of the second switching element 540 and the anode of the flywheel diode 542. The other end of the inductor 544 serves as a low potential side output terminal to the LED.

  FIG. 7C shows a configuration of a semiconductor light source control device 600 according to the third modification. One end of the second switching element 640 is connected to the low potential side output of the flyback regulator 102, and the other end is connected to the anode of the flywheel diode 642. A connection node between the other end of the second switching element 640 and the anode of the flywheel diode 642 forms a low potential side output to the LED. The cathode of flywheel diode 642 is connected to one end of inductor 644. A connection node between the cathode of the flywheel diode 642 and one end of the inductor 644 is connected to the high potential side output of the flyback regulator 102. The other end of the inductor 644 is grounded and becomes a high potential side output terminal to the LED.

  According to each of the semiconductor light source control devices 400, 500, and 600 according to the first, second, and third modified examples, as with the semiconductor light source control device 100 according to the embodiment, overshoot and undershoot of the drive current Iout are performed. Can be reduced.

  In the embodiment, the case where the negative output is realized by grounding the high potential side of the output, that is, the anode side of the plurality of LEDs, is not limited to this. For example, the anode side of the plurality of LEDs is connected to the battery voltage. It may be connected to a terminal to which a DC voltage such as Vbat is applied.

  In the embodiment, the switching frequency is not measured in real time, and instead, the relationship between the drive voltage Vout and the hysteresis width ΔI is determined based on the known relationship between the drive voltage Vout and the switching frequency, and the hysteresis width ΔI is in that relationship. Therefore, although the case where the circuit is configured to change is described, the present invention is not limited to this. For example, the semiconductor light source control device may include a circuit that measures the switching frequency of the second switching element 140, and adjust the hysteresis width so that the switching frequency thus measured falls within the target frequency range.

  In the embodiment, the semiconductor light source control device 100 has been described as including a plurality of bypass switches 110-1 to 110-N. However, the present invention is not limited to this, and the bypass switch is provided separately from the semiconductor light source control device. Also good.

  In the embodiment, the case where the hysteresis control of the drive current is performed has been described. However, the present invention is not limited to this. For example, the second switching element is set so that the voltage obtained by appropriately filtering the voltage drop Vm approaches the reference voltage corresponding to the target current. A duty ratio of 140 may be controlled.

  6 on-vehicle battery, 8 power switch, 100 semiconductor light source control device, 102 flyback regulator, 104 down converter, 106 control circuit, 108 current detection resistor, 128 output capacitor, 142 flywheel diode, 144 inductor.

  The present invention can be used in a light source control device that controls a light source.

Claims (6)

A control circuit for a down converter that includes a switching element and supplies a driving current to a light source,
  When the magnitude of the current flowing through the light source exceeds a predetermined current upper limit value, the switching element is turned off, and when the current magnitude falls below a current lower limit value smaller than the current upper limit value, the switching element is turned on,
  A control circuit, wherein a hysteresis width which is a difference between the current upper limit value and the current lower limit value is changed based on a voltage between both ends of the light source. The light source includes a plurality of semiconductor light sources connected in series ,
The control circuit according to claim 1 , wherein a bypass switch is provided in parallel with at least a part of the plurality of semiconductor light sources.   3. The control circuit according to claim 1, wherein the hysteresis width is reduced as the voltage between both ends of the light source decreases in a region where the voltage between the both ends of the light source is smaller than a predetermined threshold value.   4. The control circuit according to claim 1, wherein the hysteresis width is reduced as the voltage across the light source increases in a region where the voltage across the light source is greater than a predetermined threshold value. .   The control circuit includes:
  A hysteresis width setting circuit that generates an offset voltage that takes a peak when the voltage across the light source has a predetermined voltage value;
  A down converter drive circuit for generating a control signal based on the offset voltage and a voltage proportional to the current flowing through the light source, and driving the switching element based on the control signal;
  With
  The down converter drive circuit is
  A first resistor and a second resistor for dividing a predetermined voltage;
  (I) When the control signal is at the first level, a current corresponding to the offset voltage is sourced to the voltage dividing node of the first resistor and the second resistor, and (ii) when the control signal is at the second level. A circuit for sinking a current corresponding to the offset voltage from a voltage dividing node of the first resistor and the second resistor;
  A comparator for comparing the voltage of the voltage dividing node with a voltage proportional to a current flowing through the light source and generating the control signal indicating a comparison result;
  A gate driver for driving the switching element based on the control signal;
  The control circuit according to claim 1, further comprising:   A light source;
  A down converter for supplying a driving current to the light source;
  The control circuit according to any one of claims 1 to 5, which controls the down converter;
  A vehicular lamp characterized by comprising:

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