JP6020223B2

JP6020223B2 - Overcurrent detection circuit

Info

Publication number: JP6020223B2
Application number: JP2013022225A
Authority: JP
Inventors: 吉田　順一; 順一吉田; 佐竹　弘之; 弘之佐竹; 博樹角井
Original assignee: Denso Corp
Current assignee: Denso Corp
Priority date: 2013-02-07
Filing date: 2013-02-07
Publication date: 2016-11-02
Anticipated expiration: 2033-02-07
Also published as: JP2014154669A

Description

本発明は、出力端子の一端が負荷に接続される駆動用ＭＯＳＦＥＴに流れる過電流を検出する回路に関する。 The present invention relates to a circuit for detecting an overcurrent flowing in a driving MOSFET having one end of an output terminal connected to a load.

駆動用ＭＯＳＦＥＴを介して流れる過電流を検出するものとして、例えば検出用のＭＯＳＦＥＴ及び抵抗素子の直列回路を駆動用ＭＯＳＦＥＴに並列に接続し、検出用抵抗素子の電位を閾値電圧と比較する構成の回路がある。しかし一般的に、閾値電圧を設定する部分の回路が温度特性を有しているので、閾値のばらつきが大きくなる傾向にある。そのためのマージンを考慮すると、駆動用ＭＯＳＦＥＴのサイズを大きくする必要があり、チップ面積が大きくなることが問題となる。そこで、閾値電圧のばらつきを低減するための構成が、例えば特許文献１に開示されている。 For detecting overcurrent flowing through the drive MOSFET, for example, a series circuit of a detection MOSFET and a resistance element is connected in parallel to the drive MOSFET, and the potential of the detection resistance element is compared with a threshold voltage. There is a circuit. However, in general, since the circuit of the part where the threshold voltage is set has temperature characteristics, the variation of the threshold tends to increase. Considering the margin for that purpose, it is necessary to increase the size of the driving MOSFET, and there is a problem that the chip area increases. Therefore, for example, Patent Document 1 discloses a configuration for reducing variation in threshold voltage.

特開２００２−１７０３６号公報JP 2002-17036 A

しかしながら、特許文献１の構成では、閾値電圧を設定するために供給する定電流値をダイナミックに調整するため、比較的複雑な回路構成を採用している。
本発明は上記事情に鑑みてなされたものであり、その目的は、より簡単な構成で、駆動用ＭＯＳＦＥＴを介して流れる過電流の検出を精度良く行うことができる過電流検出回路を提供することにある。 However, in the configuration of Patent Document 1, a relatively complicated circuit configuration is adopted in order to dynamically adjust the constant current value supplied to set the threshold voltage.
The present invention has been made in view of the above circumstances, and an object thereof is to provide an overcurrent detection circuit capable of accurately detecting an overcurrent flowing through a driving MOSFET with a simpler configuration. It is in.

請求項１記載の過電流検出回路によれば、駆動用ＭＯＳＦＥＴに対し、電流検出用ＭＯＳＦＥＴ及び電流検出用抵抗素子の直列回路を並列に接続し、比較器は、電流検出用抵抗素子の端子電圧を閾値電圧と比較して過電流検出を行う。そして、前記閾値電圧を発生させる閾値設定回路を、電流源と、前記駆動用ＭＯＳＦＥＴと同一の温度特性を有する温度特性補正用ＭＯＳＦＥＴと、閾値設定用抵抗素子との直列回路で構成する。 According to the overcurrent detection circuit of claim 1, a series circuit of a current detection MOSFET and a current detection resistance element is connected in parallel to the drive MOSFET, and the comparator has a terminal voltage of the current detection resistance element. Is compared with the threshold voltage to detect overcurrent. The threshold setting circuit for generating the threshold voltage is configured by a series circuit of a current source, a temperature characteristic correcting MOSFET having the same temperature characteristics as the driving MOSFET, and a threshold setting resistance element.

このように構成すれば、電流検出用抵抗素子に流れる電流により発生する検出電圧と、閾値設定用抵抗素子に流れる電流により発生する閾値電圧の温度特性が同一になる。これにより双方の電圧の温度特性がキャンセルされるので、過電流の検出を高い精度で行うことができる。したがって、極めて簡単な構成で検出精度を向上させることができ、従来のように駆動用ＭＯＳＦＥＴの電流駆動能力に余裕を持たせる必要が無くなり、素子サイズを小型にすることができる。 With this configuration, the temperature characteristics of the detection voltage generated by the current flowing through the current detection resistance element and the threshold voltage generated by the current flowing through the threshold setting resistance element are the same. As a result, the temperature characteristics of both voltages are canceled, so that overcurrent can be detected with high accuracy. Therefore, the detection accuracy can be improved with an extremely simple configuration, and it is not necessary to provide a margin for the current driving capability of the driving MOSFET as in the prior art, and the element size can be reduced.

請求項２記載の過電流検出回路によれば、温度特性補正用ＭＯＳＦＥＴのオン抵抗値をＲ１，閾値設定用抵抗素子の抵抗値をＲ２，電流検出用抵抗素子の端子電圧をＶｘ，温度特性補正用ＭＯＳＦＥＴ及び閾値設定用抵抗素子の共通接続点の端子電圧をＶｙとする。そして、抵抗値Ｒ１，Ｒ２を、低温時，高温時でそれぞれ電流検出用抵抗素子に流れる電流の変化を考慮し、低温時，高温時の電圧Ｖｘと電圧Ｖｙとが何れも等しくなることを条件に決定する。 According to the overcurrent detection circuit of claim 2, the on-resistance value of the temperature characteristic correction MOSFET is R1, the resistance value of the threshold setting resistance element is R2, the terminal voltage of the current detection resistance element is Vx, and the temperature characteristic correction is performed. The terminal voltage at the common connection point of the MOSFET for use and the threshold setting resistor is Vy. The resistance values R1 and R2 are set under the condition that the voltage Vx and the voltage Vy at the low temperature and the high temperature are equal to each other in consideration of a change in the current flowing through the current detection resistor element at the low temperature and the high temperature. To decide.

例えば、駆動用，検出用，温度特性補正用ＭＯＳＦＥＴにそれぞれ流れる電流をＩ１〜Ｉ３とし、各ＭＯＳＦＥＴのオン抵抗は、低温時を基準として高温時にα倍になるものとする。また、駆動用，検出用ＭＯＳＦＥＴのオン抵抗をそれぞれＡ，Ｂとし、検出用抵抗素子の抵抗値をＣとする。すると、低温時において電流Ｉ１が過電流となった場合に、検出用ＭＯＳＦＥＴに流れる電流Ｉ２は、
Ｉ２＝Ａ／（Ｂ＋Ｃ）・Ｉ１ …（１）
となり、高温時において電流Ｉ１が過電流となった場合に流れる電流Ｉ２は、
Ｉ２＝αＡ／（αＢ＋Ｃ）・Ｉ１ …（２）
となる。 For example, the currents flowing through the driving, detection, and temperature characteristic correction MOSFETs are I1 to I3, respectively, and the on-resistance of each MOSFET is α times higher when the temperature is higher than the lower temperature. The on-resistances of the driving and detection MOSFETs are A and B, respectively, and the resistance value of the detection resistance element is C. Then, when the current I1 becomes an overcurrent at a low temperature, the current I2 flowing through the detection MOSFET is
I2 = A / (B + C) · I1 (1)
The current I2 that flows when the current I1 becomes an overcurrent at a high temperature is
I2 = αA / (αB + C) · I1 (2)
It becomes.

そして、低温時の電圧Ｖｘ，電圧Ｖｙは、
Ｖｘ＝Ａ／（Ｂ＋Ｃ）・Ｉ１・Ｃ …（３）
Ｖｙ＝（Ｒ１＋Ｒ２）・Ｉ３ …（４）
高温時の電圧Ｖｘ，電圧Ｖｙは、
Ｖｘ＝αＡ／（αＢ＋Ｃ）・Ｉ１・Ｃ …（５）
Ｖｙ＝（αＲ１＋Ｒ２）・Ｉ３ …（６）
となる。 And the voltage Vx and the voltage Vy at the time of low temperature are:
Vx = A / (B + C) · I1 · C (3)
Vy = (R1 + R2) · I3 (4)
The voltage Vx and voltage Vy at high temperature are
Vx = αA / (αB + C) · I1 · C (5)
Vy = (αR1 + R2) · I3 (6)
It becomes.

これらより、（３）式と（４）式，（５）式と（６）式がそれぞれ等しくなることが条件となるから、これらの式を解くと、
Ｒ１＝Ｉ１／Ｉ３｛αＡ・Ｃ／（αＢ＋Ｃ）−Ａ・Ｃ／（Ｂ＋Ｃ）｝
／（α−１） …（７）
Ｒ２＝Ｉ１／Ｉ３［Ａ・Ｃ／（Ｂ＋Ｃ）−｛αＡ・Ｃ／（αＢ＋Ｃ）
−Ａ・Ｃ／（Ｂ＋Ｃ）｝／（α−１）］ …（８）
が得られる。したがって、温度特性補正用ＭＯＳＦＥＴのオン抵抗値と閾値設定用抵抗素子の抵抗値とを、具体的に決定することができる。 From these, since it is a condition that the equations (3) and (4), (5) and (6) are equal, solving these equations,
R1 = I1 / I3 {αA · C / (αB + C) −A · C / (B + C)}
/ (Α-1) (7)
R2 = I1 / I3 [A · C / (B + C) − {αA · C / (αB + C)
−A · C / (B + C)} / (α−1)] (8)
Is obtained. Therefore, the ON resistance value of the temperature characteristic correcting MOSFET and the resistance value of the threshold value setting resistance element can be specifically determined.

第１実施形態であり、過電流検出回路の構成を示す図The figure which is 1st Embodiment and shows the structure of an overcurrent detection circuit 第２実施形態を示す図１相当図FIG. 1 equivalent view showing the second embodiment 第３実施形態を示す図１相当図FIG. 1 equivalent diagram showing the third embodiment （ａ）は第４実施形態を示す図１相当図、（ｂ）は電流源の構成を示す図(A) is a diagram corresponding to FIG. 1 showing the fourth embodiment, (b) is a diagram showing the configuration of the current source. 第５実施形態であり、ハーフブリッジ回路を構成した場合の図１相当図FIG. 1 is a diagram corresponding to FIG. 1 when a half bridge circuit is configured according to the fifth embodiment. 第６実施形態であり、Ｈブリッジ回路を構成した場合の図１相当図FIG. 1 is a diagram corresponding to FIG. 1 when an H-bridge circuit is configured according to the sixth embodiment.

（第１実施形態）
以下、第１実施形態について説明する。図１（ａ）において、図示しない負荷が接続される出力端子ＯＵＴとグランドとの間には、駆動用ＭＯＳＦＥＴ１と、検出用ＭＯＳＦＥＴ２及び検出用抵抗素子３の直列回路とが接続されている。これらのＭＯＳＦＥＴ１及び２は何れもＮチャネル型であり、これらのゲートには、ゲート制御部４により共通のゲート信号が与えられる。ゲート制御部４には、例えば５Ｖの電源ＶＣＣが動作用電源として供給されている。尚、検出用ＭＯＳＦＥＴ２のサイズは、駆動用ＭＯＳＦＥＴ１のサイズ以下に設定されている。 (First embodiment)
The first embodiment will be described below. In FIG. 1A, a drive MOSFET 1, a series circuit of a detection MOSFET 2 and a detection resistance element 3 are connected between an output terminal OUT to which a load (not shown) is connected and the ground. These MOSFETs 1 and 2 are both N-channel type, and a common gate signal is given to these gates by the gate controller 4. For example, a power supply VCC of 5V is supplied to the gate controller 4 as an operation power supply. The size of the detection MOSFET 2 is set to be equal to or smaller than the size of the drive MOSFET 1.

検出用ＭＯＳＦＥＴ２のソースは、ヒステリシス付きコンパレータである比較器５の非反転入力端子に接続されている。電源ＶＣＣとグランドとの間には、電流源６，温度特性補正用ＭＯＳＦＥＴ７及び閾値設定用抵抗素子８の直列回路が接続されている。この直列回路は閾値設定回路９を構成している。温度特性補正用ＭＯＳＦＥＴ７もＮチャネル型であり、そのドレインは、比較器５の反転入力端子に接続されており、ゲートは電源ＶＣＣにプルアップされている。 The source of the detection MOSFET 2 is connected to the non-inverting input terminal of the comparator 5 which is a comparator with hysteresis. A series circuit of a current source 6, a temperature characteristic correcting MOSFET 7 and a threshold setting resistance element 8 is connected between the power supply VCC and the ground. This series circuit constitutes a threshold setting circuit 9. The temperature characteristic correcting MOSFET 7 is also an N-channel type, its drain is connected to the inverting input terminal of the comparator 5, and its gate is pulled up to the power supply VCC.

ここで、温度特性補正用ＭＯＳＦＥＴ７には、駆動用ＭＯＳＦＥＴ１と同一の温度特性を有する素子が選択されている。尚、検出用ＭＯＳＦＥＴ２の温度特性が駆動用ＭＯＳＦＥＴ１と同一であることは言うまでもない。また、閾値設定用抵抗素子８についても、検出用抵抗素子３と同一の温度特性を有する素子が選択されている。 Here, an element having the same temperature characteristic as that of the driving MOSFET 1 is selected as the temperature characteristic correcting MOSFET 7. Needless to say, the temperature characteristics of the detection MOSFET 2 are the same as those of the drive MOSFET 1. As the threshold value setting resistance element 8, an element having the same temperature characteristics as the detection resistance element 3 is selected.

比較器５は、検出用抵抗素子３の端子電圧（検出電圧）を、温度特性補正用ＭＯＳＦＥＴ７のドレイン電圧である閾値電圧と比較し、前者が後者を超えると出力端子をハイレベルにして、過電流検出信号をゲート制御部４に出力する。ゲート制御部４は、上記過電流検出信号が入力されると、駆動用ＭＯＳＦＥＴ１の駆動を停止する。以上において、駆動用ＭＯＳＦＥＴ１及びゲート制御部４を除いたものが過電流検出回路１０を構成している。 The comparator 5 compares the terminal voltage (detection voltage) of the detection resistance element 3 with a threshold voltage that is the drain voltage of the temperature characteristic correction MOSFET 7, and if the former exceeds the latter, the output terminal is set to a high level. The current detection signal is output to the gate control unit 4. When the overcurrent detection signal is input, the gate controller 4 stops driving the driving MOSFET 1. In the above description, the overcurrent detection circuit 10 is configured except for the driving MOSFET 1 and the gate control unit 4.

次に、本実施形態の作用について説明する。ここで、駆動用，検出用，温度特性補正用ＭＯＳＦＥＴ１，２，７にそれぞれ流れる電流をＩ１〜Ｉ３とする。ゲート制御部４がゲート駆動信号をハイレベルにすると、検出用ＭＯＳＦＥＴ２は駆動用ＭＯＳＦＥＴ１と同時にオンして、両者のサイズ比に応じた電流Ｉ２を検出用抵抗素子３に流す。したがって、比較器５の非反転入力端子に与えられる検出電圧は、検出用抵抗素子３の抵抗値をＣΩとすると、Ｃ・Ｉ２となる。一方、閾値設定回路９においては、電流源６より供給される電流Ｉ３が、温度特性補正用ＭＯＳＦＥＴ７を介して閾値設定用抵抗素子８に流れる。したがって、比較器５の反転入力端子に与えられる閾値電圧は、閾値設定用抵抗素子８の抵抗値をＥΩとすると、Ｅ・Ｉ３となる。 Next, the operation of this embodiment will be described. Here, currents flowing through the driving, detecting, and temperature characteristic correcting MOSFETs 1, 2, and 7 are I1 to I3, respectively. When the gate control unit 4 sets the gate drive signal to the high level, the detection MOSFET 2 is turned on simultaneously with the drive MOSFET 1, and a current I2 corresponding to the size ratio of both is supplied to the detection resistance element 3. Therefore, the detection voltage applied to the non-inverting input terminal of the comparator 5 is C · I2 when the resistance value of the detection resistance element 3 is CΩ. On the other hand, in the threshold setting circuit 9, the current I 3 supplied from the current source 6 flows to the threshold setting resistor 8 through the temperature characteristic correcting MOSFET 7. Therefore, the threshold voltage applied to the inverting input terminal of the comparator 5 is E · I3 when the resistance value of the threshold setting resistance element 8 is EΩ.

駆動用，検出用，温度特性補正用ＭＯＳＦＥＴ１，２，７が何れも同じ温度特性を有しているので、検出用抵抗素子３，閾値設定用抵抗素子８に流れる電流Ｉ２，Ｉ３により発生する検出電圧と閾値電圧とは、同じ温度特性を有することになる。したがって、上述のように、比較器５が検出電圧を閾値電圧と比較して駆動用ＭＯＳＦＥＴ１に流れる電流Ｉ１が過電流となったことを、高い精度で検出できる。 Since the drive, detection, and temperature characteristic correction MOSFETs 1, 2, and 7 all have the same temperature characteristics, detection is generated by the currents I2 and I3 flowing through the detection resistance element 3 and the threshold setting resistance element 8. The voltage and the threshold voltage have the same temperature characteristics. Therefore, as described above, the comparator 5 compares the detection voltage with the threshold voltage and can detect with high accuracy that the current I1 flowing through the driving MOSFET 1 has become an overcurrent.

ここで、駆動用，検出用ＭＯＳＦＥＴ１，２のオン抵抗をそれぞれＡΩ，ＢΩとし、検出用抵抗素子の抵抗値をＣΩとする。また、温度特性補正用ＭＯＳＦＥＴ７のオン抵抗をＤΩ，閾値設定用抵抗素子８の抵抗値ＥΩとして、オン抵抗Ｄ（＝Ｒ１）Ω，抵抗値Ｅ（＝Ｒ２）Ωを決定する方法の一例を示す。各ＭＯＳＦＥＴ１，２，７のオン抵抗は、低温時を基準として高温時にα倍になるものとする。 Here, the ON resistances of the driving and detection MOSFETs 1 and 2 are AΩ and BΩ, respectively, and the resistance value of the detection resistance element is CΩ. Further, an example of a method for determining the on-resistance D (= R1) Ω and the resistance value E (= R2) Ω, where the on-resistance of the temperature characteristic correction MOSFET 7 is DΩ and the resistance value EΩ of the threshold setting resistor 8 is shown. . The on-resistances of the MOSFETs 1, 2, and 7 are assumed to be α times at high temperatures with reference to low temperatures.

すると、低温時，高温時に検出用ＭＯＳＦＥＴ２に流れる電流Ｉ２は、それぞれ前述の（１）式，（２）式で表される。また、比較器５の非反転入力端子，反転入力端子の電位をそれぞれＶｘ，Ｖｙとすると、低温時の電圧Ｖｘ，電圧Ｖｙはそれぞれ（３）式，（４）式で表され、高温時の電圧Ｖｘ，電圧Ｖｙはそれぞれ（５）式，（６）式で表される。 Then, the current I2 flowing through the detection MOSFET 2 at the low temperature and at the high temperature is expressed by the above-described equations (1) and (2), respectively. Further, assuming that the potentials of the non-inverting input terminal and the inverting input terminal of the comparator 5 are Vx and Vy, respectively, the low-temperature voltage Vx and the voltage Vy are expressed by the equations (3) and (4), respectively. The voltage Vx and the voltage Vy are expressed by equations (5) and (6), respectively.

そして、オン抵抗ＤΩ，抵抗値ＥΩは、低温時の電圧Ｖｘ，電圧Ｖｙと、高温時の電圧Ｖｘ，電圧Ｖｙとがそれぞれ等しくなるように決定すれば良い。すなわち、
Ａ／（Ｂ＋Ｃ）・Ｉ１・Ｃ＝Ｉ３・（Ｄ＋Ｅ） …（１０）
αＡ・Ｃ／（αＢ＋Ｃ）・Ｉ１・Ｃ＝Ｉ３・（αＤ＋Ｅ） …（１１）
を解いて、オン抵抗ＤΩ，抵抗値ＥΩを求めると（７）式，（８）式が得られる。 The on-resistance DΩ and the resistance value EΩ may be determined so that the low-temperature voltage Vx and the voltage Vy are equal to the high-temperature voltage Vx and the voltage Vy, respectively. That is,
A / (B + C) · I1 · C = I3 · (D + E) (10)
αA · C / (αB + C) · I1 · C = I3 · (αD + E) (11)
To obtain the on-resistance DΩ and the resistance value EΩ, the equations (7) and (8) are obtained.

また、図１（ｂ）に示すように、各電流値及び抵抗値に具体的な数値を付与した場合について、オン抵抗ＤΩ，抵抗値ＥΩを求めると以下のようになる。
Ａ＝１００ｍ，Ｂ＝１００，Ｃ＝１０，電流Ｉ１の過電流値を１［Ａ］，Ｉ３＝２０μ［Ａ］とし、α＝２とする。低温時に検出用ＭＯＳＦＥＴに流れる電流Ｉ２は、（１）式より、
Ｉ２＝１００・１０^−３／（１００＋１０）・１＝０．９０９［ｍＡ］
となり、高温時に流れる電流Ｉ２は、（２）式より、
Ｉ２＝２・１００・１０^−３／（２・１００＋１０）・１＝０．９５２［ｍＡ］
低温時の電圧Ｖｘ，電圧Ｖｙは、（３）式，（４）式より、
Ｖｘ＝０．９０９・１０^−３・１０＝９．０９［ｍＶ］ …（１２）
Ｖｙ＝２０・１０^−６・（Ｄ＋Ｅ） …（１３）
一方、高温時の電圧Ｖｘ，電圧Ｖｙは、（５）式，（６）式より、
Ｖｘ＝０．９５２・１０^−３・１０＝９．５２［ｍＶ］ …（１４）
Ｖｙ＝２０・１０^−６・（２Ｄ＋Ｅ） …（１５）
となる。 Further, as shown in FIG. 1B, when a specific numerical value is given to each current value and resistance value, the ON resistance DΩ and the resistance value EΩ are obtained as follows.
A = 100 m, B = 100, C = 10, the overcurrent value of the current I1 is 1 [A], I3 = 20 μ [A], and α = 2. The current I2 flowing through the detection MOSFET at a low temperature is expressed by the following equation (1):
I2 = 100 · 10 ⁻³ /(100+10)·1=0.909 [mA]
The current I2 flowing at a high temperature is
I2 = 2 · 100 · 10 ⁻³ /(2·100+10)·1=0.925 [mA]
The voltage Vx and the voltage Vy at the time of low temperature are obtained from the equations (3) and (4).
Vx = 0.909 · 10 ⁻³ · 10 = 9.09 [mV] (12)
Vy = 20 · 10 ⁻⁶ · (D + E) (13)
On the other hand, the voltage Vx and the voltage Vy at the time of high temperature are obtained from the equations (5) and (6),
Vx = 0.952 · 10 ⁻³ · 10 = 9.52 [mV] (14)
Vy = 20 · 10 ⁻⁶ · (2D + E) (15)
It becomes.

そして、（１２）式と（１３）式，（１４）式と（１５）式がそれぞれ等しくなることが条件となるから、
２０・１０^−６・（Ｄ＋Ｅ）＝９．０９・１０^−３ …（１６）
２０・１０^−６・（２Ｄ＋Ｅ）＝９．５２・１０^−３ …（１７）
となり、これらの式を解くと、
Ｄ＝２１．５［Ω］
Ｅ＝４３３［Ω］
が得られる。 And, since the condition is that the expressions (12) and (13), the expressions (14) and (15) are equal,
20 · 10 ⁻⁶ · (D + E) = 9.09 · 10 ⁻³ (16)
20 · 10 ⁻⁶ · (2D + E) = 9.52 · 10 ⁻³ (17)
And solving these equations,
D = 21.5 [Ω]
E = 433 [Ω]
Is obtained.

以上のように本実施形態によれば、過電流検出回路１０は、駆動用ＭＯＳＦＥＴ１に対し、電流検出用ＭＯＳＦＥＴ２及び電流検出用抵抗素子３の直列回路を並列に接続し、比較器５は、電流検出用抵抗素子３の端子電圧を閾値電圧と比較して過電流検出を行う。そして、閾値電圧を発生させる閾値設定回路９を、電流源６と、駆動用ＭＯＳＦＥＴ１と同一の温度特性を有する温度特性補正用ＭＯＳＦＥＴ７と、閾値設定用抵抗素子８との直列回路で構成した。 As described above, according to the present embodiment, the overcurrent detection circuit 10 connects the series circuit of the current detection MOSFET 2 and the current detection resistance element 3 in parallel to the drive MOSFET 1, and the comparator 5 Overcurrent detection is performed by comparing the terminal voltage of the detection resistance element 3 with a threshold voltage. The threshold setting circuit 9 for generating the threshold voltage is constituted by a series circuit of a current source 6, a temperature characteristic correcting MOSFET 7 having the same temperature characteristics as the driving MOSFET 1, and a threshold setting resistance element 8.

このように構成すれば、電流検出用抵抗素子３に流れる電流Ｉ２により発生する検出電圧と、閾値設定用抵抗素子８に流れる電流Ｉ３により発生する閾値電圧の温度特性が同一になり，双方の電圧の温度特性がキャンセルされるので、過電流の検出を高い精度で行うことができる。したがって、極めて簡単な構成で検出精度の向上を図ることができ、従来のように駆動用ＭＯＳＦＥＴＱの電流駆動能力に余裕を持たせる必要が無くなり、素子サイズを小型にできる。 With this configuration, the temperature characteristics of the detection voltage generated by the current I2 flowing through the current detection resistor element 3 and the threshold voltage generated by the current I3 flowing through the threshold setting resistor element 8 are the same, and both voltages Therefore, the overcurrent can be detected with high accuracy. Therefore, the detection accuracy can be improved with an extremely simple configuration, and it is not necessary to provide a margin for the current driving capability of the driving MOSFET Q as in the prior art, and the element size can be reduced.

また、温度特性補正用ＭＯＳＦＥＴ７のオン抵抗値をＤ，閾値設定用抵抗素子８の抵抗値をＥ，比較器５の非反転入力端子，反転入力端子の電位をそれぞれＶｘ，Ｖｙとして、抵抗値Ｄ，Ｅを、低温時，高温時でそれぞれ電流検出用抵抗素子３に流れる電流の変化を考慮し、低温時，高温時の電圧Ｖｘと電圧Ｖｙとが何れも等しくなることを条件に決定する。したがって、温度特性補正用ＭＯＳＦＥＴ７のオン抵抗値と閾値設定用抵抗素子８の抵抗値とを、具体的に決定することができる。そして、電流検出用抵抗素子３と閾値設定用抵抗素子８とを同一の温度特性を有する抵抗素子としたので、過電流の検出精度を一層高めることができる。 Further, the resistance value D is set such that the on-resistance value of the temperature characteristic correcting MOSFET 7 is D, the resistance value of the threshold setting resistor 8 is E, and the potentials of the non-inverting input terminal and the inverting input terminal of the comparator 5 are Vx and Vy, respectively. , E are determined on the condition that the voltage Vx and the voltage Vy at the low temperature and the high temperature are equal in consideration of changes in the current flowing through the current detection resistor element 3 at the low temperature and the high temperature, respectively. Therefore, the on-resistance value of the temperature characteristic correcting MOSFET 7 and the resistance value of the threshold-setting resistance element 8 can be specifically determined. Since the current detection resistance element 3 and the threshold setting resistance element 8 are resistance elements having the same temperature characteristics, the overcurrent detection accuracy can be further enhanced.

（第２実施形態）
以下、第１実施形態と同一部分には同一符号を付して説明を省略し、異なる部分について説明する。図２に示す電圧検出回路１０’は、第１実施形態の閾値設定回路９を、ＮチャネルＭＯＳＦＥＴ７と閾値設定用抵抗素子８との直列接続順を入れ替えた、閾値設定回路９’に置き換えたものである。このように構成される第２実施形態による場合も、第１実施形態と同様の効果が得られる。 (Second Embodiment)
Hereinafter, the same parts as those in the first embodiment are denoted by the same reference numerals, description thereof will be omitted, and different parts will be described. The voltage detection circuit 10 ′ shown in FIG. 2 is obtained by replacing the threshold setting circuit 9 of the first embodiment with a threshold setting circuit 9 ′ in which the series connection order of the N-channel MOSFET 7 and the threshold setting resistance element 8 is switched. It is. In the case of the second embodiment configured as described above, the same effect as that of the first embodiment can be obtained.

（第３実施形態）
第１実施形態は、ローサイド駆動方式であったが、図３に示す過電流検出回路１１はハイサイド駆動方式に適用したものである。駆動用ＭＯＳＦＥＴ１２と、検出用抵抗素子３及び検出用ＭＯＳＦＥＴ１３の直列回路とは、１４Ｖの電源ＶＢと出力端子ＯＵＴとの間に接続されている。この場合、ＭＯＳＦＥＴ１２及び１３はＰチャネル型であり、これらのゲートには、ゲート制御部４に替わるゲート制御部１４より共通のゲート駆動信号が与えられる。ゲート制御部１４には、電源ＶＣＣに加えて、ゲート駆動用電源として電源ＶＢも供給されている。 (Third embodiment)
The first embodiment is a low-side drive method, but the overcurrent detection circuit 11 shown in FIG. 3 is applied to the high-side drive method. The driving MOSFET 12 and the series circuit of the detection resistive element 3 and the detection MOSFET 13 are connected between a 14 V power supply VB and the output terminal OUT. In this case, the MOSFETs 12 and 13 are P-channel type, and a common gate drive signal is given to these gates from a gate control unit 14 in place of the gate control unit 4. In addition to the power supply VCC, the gate control unit 14 is also supplied with a power supply VB as a gate drive power supply.

電源ＶＢとグランドとの間には、抵抗素子８，温度特性補正用ＭＯＳＦＥＴ１５（Ｐチャネル型）及び電流源６の直列回路が接続されており、これらが閾値設定回路１６を構成している。温度特性補正用ＭＯＳＦＥＴ１５のゲートはグランドにプルダウンされている。そして、検出用ＭＯＳＦＥＴ１３のソースは、比較器５の非反転入力端子に接続されている。また、温度特性補正用ＭＯＳＦＥＴ１５は、駆動用ＭＯＳＦＥＴ１２と同一の温度特性を有する素子が選択されている。 Between the power supply VB and the ground, a series circuit of a resistance element 8, a temperature characteristic correction MOSFET 15 (P channel type) and a current source 6 is connected, and these constitute a threshold setting circuit 16. The gate of the temperature characteristic correcting MOSFET 15 is pulled down to the ground. The source of the detection MOSFET 13 is connected to the non-inverting input terminal of the comparator 5. For the temperature characteristic correcting MOSFET 15, an element having the same temperature characteristic as that of the driving MOSFET 12 is selected.

次に、第３実施形態の作用について説明する。ゲート制御部１４がゲート駆動信号をローレベルにすると、検出用ＭＯＳＦＥＴ１３は駆動用ＭＯＳＦＥＴ１２と同時にオンして、両者のサイズ比に応じた電流Ｉ２を検出用抵抗素子３に流す。したがって、比較器５の非反転入力端子に与えられる検出電圧は、検出用抵抗素子３の抵抗値をＣΩとすると、（ＶＢ−Ｃ・Ｉ２）となる。一方、閾値設定回路１６においては、電流源６がシンクする電流Ｉ３が、閾値設定用抵抗素子８を介して温度特性補正用ＭＯＳＦＥＴ１５に流れる。したがって、比較器５の反転入力端子に与えられる閾値電圧は、温度特性補正用ＭＯＳＦＥＴ１５のオン抵抗をＤΩ，閾値設定用抵抗素子８の抵抗値をＥΩとすると、｛ＶＢ−（Ｄ＋Ｅ）・Ｉ３｝となる。 Next, the operation of the third embodiment will be described. When the gate control unit 14 sets the gate drive signal to a low level, the detection MOSFET 13 is turned on simultaneously with the drive MOSFET 12, and a current I2 corresponding to the size ratio of both is supplied to the detection resistance element 3. Therefore, the detection voltage applied to the non-inverting input terminal of the comparator 5 is (VB−C · I2), where the resistance value of the detection resistance element 3 is CΩ. On the other hand, in the threshold setting circuit 16, the current I 3 sunk by the current source 6 flows to the temperature characteristic correction MOSFET 15 via the threshold setting resistor 8. Therefore, the threshold voltage applied to the inverting input terminal of the comparator 5 is {VB− (D + E) · I3}, where DΩ is the on-resistance of the temperature characteristic correction MOSFET 15 and EΩ is the resistance value of the threshold setting resistor element 8. It becomes.

以上のように構成される第３実施形態によれば、過電流検出回路１１を、Ｐチャネル型の駆動用ＭＯＳＦＥＴ１２を用いたハイサイド駆動方式に適用した場合も、第１実施形態と同様の効果が得られる。 According to the third embodiment configured as described above, even when the overcurrent detection circuit 11 is applied to the high-side drive method using the P-channel type drive MOSFET 12, the same effect as the first embodiment is obtained. Is obtained.

（第４実施形態）
第４実施形態の過電流検出回路２１は、第１実施形態の過電流検出回路１を構成していた閾値設定回路９を、閾値設定回路２２に置き換えたもので、閾値設定回路２２は、電流源２３と閾値設定用抵抗素子８との直列回路で構成されている。図４（ｂ）に示す電流源２３において、電源ＶＣＣにソースが接続される２つのＰチャネルＭＯＳＦＥＴ２４ａ，２４ｂは、カレントミラー回路２４を構成している。ＰチャネルＭＯＳＦＥＴ２４ａ，２４ｂのドレイン（それぞれ主電流経路，ミラー電流経路）は、それぞれ制御用抵抗素子２５，閾値設定用抵抗素子８を介してグランドに接続されている。 (Fourth embodiment)
The overcurrent detection circuit 21 of the fourth embodiment is obtained by replacing the threshold setting circuit 9 constituting the overcurrent detection circuit 1 of the first embodiment with a threshold setting circuit 22, and the threshold setting circuit 22 It is composed of a series circuit of a source 23 and a threshold setting resistance element 8. In the current source 23 shown in FIG. 4B, two P-channel MOSFETs 24 a and 24 b whose sources are connected to the power supply VCC constitute a current mirror circuit 24. The drains (main current path and mirror current path) of the P-channel MOSFETs 24a and 24b are connected to the ground via the control resistance element 25 and the threshold setting resistance element 8, respectively.

また、電源ＶＣＣとグランドとの間には、抵抗素子２６及び温度特性補正用ＭＯＳＦＥＴ２７の直列回路が接続されている。温度特性補正用ＭＯＳＦＥＴ２７はＮチャネル型であり、そのゲートは電源ＶＣＣにプルアップされている。また、温度特性補正用ＭＯＳＦＥＴ２７は、駆動用ＭＯＳＦＥＴ１と同一の温度特性を有する素子が選択されている。これらは基準電圧生成回路２８を構成している。 Further, a series circuit of a resistance element 26 and a temperature characteristic correcting MOSFET 27 is connected between the power supply VCC and the ground. The temperature characteristic correcting MOSFET 27 is an N-channel type, and its gate is pulled up to the power supply VCC. For the temperature characteristic correcting MOSFET 27, an element having the same temperature characteristic as that of the driving MOSFET 1 is selected. These constitute the reference voltage generation circuit 28.

制御用抵抗素子２６及び温度特性補正用ＭＯＳＦＥＴ２７の共通接続点とＰチャネルＭＯＳＦＥＴ２４ａのドレインとは、オペアンプ２９（電流制御回路）の非反転入力端子と反転入力端子とにそれぞれ接続されている。そして、オペアンプ２９の出力端子は、ＰチャネルＭＯＳＦＥＴ２４ａ，２４ｂのゲートに接続されている。 A common connection point of the control resistance element 26 and the temperature characteristic correction MOSFET 27 and the drain of the P-channel MOSFET 24a are connected to the non-inverting input terminal and the inverting input terminal of the operational amplifier 29 (current control circuit), respectively. The output terminal of the operational amplifier 29 is connected to the gates of the P-channel MOSFETs 24a and 24b.

次に、第４実施形態の作用について説明する。オペアンプ２９の非反転入力端子の電位は、電源ＶＣＣの電圧を、制御用抵抗素子２６の抵抗値と、温度特性補正用ＭＯＳＦＥＴ２７のオン抵抗とで分圧したもの（基準電圧）となる。オペアンプ２９は、ＰチャネルＭＯＳＦＥＴ２４ａ，２４ｂのゲート電位を、抵抗素子２５の端子電圧と前記制御電圧との差に応じた電圧とするように制御する。したがって、ＰチャネルＭＯＳＦＥＴ２４ｂを介して閾値設定用抵抗素子８に流れる電流Ｉ４は、基準電圧によって制御される。 Next, the operation of the fourth embodiment will be described. The potential of the non-inverting input terminal of the operational amplifier 29 is a voltage (reference voltage) obtained by dividing the voltage of the power supply VCC by the resistance value of the control resistance element 26 and the ON resistance of the temperature characteristic correction MOSFET 27. The operational amplifier 29 controls the gate potentials of the P-channel MOSFETs 24a and 24b so as to be a voltage corresponding to the difference between the terminal voltage of the resistance element 25 and the control voltage. Therefore, the current I4 flowing through the threshold setting resistance element 8 via the P-channel MOSFET 24b is controlled by the reference voltage.

そして、温度特性補正用ＭＯＳＦＥＴ２７は、駆動用ＭＯＳＦＥＴ１と同一の温度特性を有しているので、閾値設定用抵抗素子８の端子電圧；閾値電圧は、検出電圧と同様に変化することになり、温度特性がキャンセルされる。 Since the temperature characteristic correcting MOSFET 27 has the same temperature characteristic as that of the driving MOSFET 1, the terminal voltage of the threshold setting resistance element 8; the threshold voltage changes in the same manner as the detection voltage. The property is cancelled.

以上のように第４実施形態によれば、過電流検出回路２１を構成する電流源２３を、主電流経路に制御用抵抗素子２５が接続され、ミラー電流経路に閾値設定用抵抗素子８が接続されるカレントミラー回路２４と、基準電圧生成回路２８と、制御用抵抗素子２８の端子電圧と基準電圧とがそれぞれ入力端子に与えられ、カレントミラー回路２４に流れるミラー電流を制御するオペアンプ２９とで構成する。そして、基準電圧生成回路２８は、駆動用ＭＯＳＦＥＴ１と同一の温度特性を有する温度特性補正用ＭＯＳＦＥＴ２７のオン抵抗を用いて基準電圧を発生させる。このような構成を用いた場合も、第１実施形態と同様の効果が得られる。 As described above, according to the fourth embodiment, the current source 23 constituting the overcurrent detection circuit 21 is connected to the control resistor element 25 in the main current path, and the threshold setting resistor element 8 is connected to the mirror current path. A current mirror circuit 24, a reference voltage generation circuit 28, and a terminal voltage and a reference voltage of the control resistance element 28 are applied to input terminals, respectively, and an operational amplifier 29 that controls a mirror current flowing in the current mirror circuit 24. Configure. The reference voltage generation circuit 28 generates a reference voltage using the on-resistance of the temperature characteristic correction MOSFET 27 having the same temperature characteristic as that of the driving MOSFET 1. Even when such a configuration is used, the same effect as in the first embodiment can be obtained.

（第５実施形態）
図５は、電源ＶＢとグランド（ＶＳＳ）との間に、駆動用ＭＯＳＦＥＴ１２及び１の直列回路を接続してハーフブリッジ回路３１を構成した場合に、駆動用ＭＯＳＦＥＴ１２及び１のそれぞれに過電流検出回路１１及び１０を配置したものである。出力端子ＯＵＴに接続される負荷としては、例えばコイル３２及びコンデンサ３３から成るＬＣ共振回路等である。 (Fifth embodiment)
FIG. 5 shows an overcurrent detection circuit for each of the driving MOSFETs 12 and 1 when a half bridge circuit 31 is configured by connecting a series circuit of the driving MOSFETs 12 and 1 between the power supply VB and the ground (VSS). 11 and 10 are arranged. The load connected to the output terminal OUT is, for example, an LC resonance circuit including a coil 32 and a capacitor 33.

（第６実施形態）
図６は、電源ＶＢとグランドとの間に、駆動用ＭＯＳＦＥＴ１２Ａ及び１Ａの直列回路と、駆動用ＭＯＳＦＥＴ１２Ｂ及び１Ｂの直列回路とを接続してＨブリッジ回路４１を構成した場合に、駆動用ＭＯＳＦＥＴ１２及び１のそれぞれに過電流検出回路１１及び１０を配置したものである。２つの出力端子ＯＵＴ＿Ａ，ＯＵＴ＿Ｂ間に接続される負荷としては、例えばＤＣモータ４２等である。 (Sixth embodiment)
FIG. 6 shows a case where an H bridge circuit 41 is configured by connecting a series circuit of driving MOSFETs 12A and 1A and a series circuit of driving MOSFETs 12B and 1B between a power source VB and a ground. 1 are provided with overcurrent detection circuits 11 and 10 respectively. An example of the load connected between the two output terminals OUT_A and OUT_B is the DC motor 42.

本発明は上記した、又は図面に記載した実施例にのみ限定されるものではなく、以下の用な変形又は拡張が可能である。
閾値設定用抵抗素子８と検出用抵抗素子３との温度特性が異なる場合でも、第１実施形態で示した計算と同様の考え方により、それぞれの抵抗値を最適化するように決定できる。
電流制御回路はオペアンプ２９に限らず、例えば複数のトランジスタを組み合わせて構成しても良い。 The present invention is not limited to the embodiments described above or shown in the drawings, and the following modifications or expansions are possible.
Even when the temperature characteristics of the threshold setting resistance element 8 and the detection resistance element 3 are different, the respective resistance values can be determined to be optimized based on the same idea as the calculation shown in the first embodiment.
The current control circuit is not limited to the operational amplifier 29, and may be configured by combining a plurality of transistors, for example.

図面中、１は駆動用ＭＯＳＦＥＴ、２は検出用ＭＯＳＦＥＴ、３は検出用抵抗素子、５は比較器、６は電流源、７は温度特性補正用ＭＯＳＦＥＴ、８は閾値設定用抵抗素子、９は閾値設定回路、１０は過電流検出回路を示す。 In the drawing, 1 is a drive MOSFET, 2 is a detection MOSFET, 3 is a detection resistance element, 5 is a comparator, 6 is a current source, 7 is a temperature characteristic correction MOSFET, 8 is a threshold setting resistance element, and 9 is A threshold setting circuit 10 indicates an overcurrent detection circuit.

Claims

A series circuit of a current detection MOSFET (2, 13) and a current detection resistor element (3) connected in parallel to a drive MOSFET (1, 12) having one end of an output terminal connected to a load;
A comparator (5) for comparing a terminal voltage of the current detection resistance element with a threshold voltage;
It comprises a series circuit of a current source (6), temperature characteristic correcting MOSFETs (7, 15) having the same temperature characteristics as the driving MOSFET, and a threshold setting resistance element (8), and the threshold voltage is An overcurrent detection circuit (1, 10, 10 ′, 11), characterized by comprising a threshold setting circuit (9) to be generated.

The on-resistance value of the temperature characteristic correction MOSFET is R1, the resistance value of the threshold setting resistance element is R2, the terminal voltage of the current detection resistance element is Vx, the temperature characteristic correction MOSFET and the threshold setting resistance element If the terminal voltage at the common connection point is Vy,
The resistance values R1 and R2 take into account changes in the current flowing through the current detection resistor element at low temperature and high temperature, and the voltage Vx and the voltage Vy at low temperature and high temperature are both 2. The overcurrent detection circuit (1) according to claim 1, wherein the overcurrent detection circuit (1) is determined on condition that they are equal.

A series circuit of a current detection MOSFET (2) and a current detection resistor element (3) connected in parallel to the drive MOSFET (1) connected to the load at one end of the output terminal;
A comparator (5) for comparing a terminal voltage of the current detection resistance element with a threshold voltage;
A threshold setting circuit (22) configured by a series circuit of a current source (23) and a threshold setting resistance element (8);
The current source includes a current mirror circuit (24) in which a control resistor element (25) is connected to a main current path, and the threshold setting resistor element is connected to a mirror current path;
A reference voltage generation circuit (28) for generating a reference voltage;
A terminal voltage of the control resistance element and the reference voltage are applied to the input terminals, respectively, and a current control circuit (29) for controlling a mirror current flowing in the current mirror circuit;
An overcurrent detection circuit (21), wherein the reference voltage generation circuit generates the reference voltage using an on-resistance of a temperature characteristic correction MOSFET (27) having the same temperature characteristic as that of the driving MOSFET.

4. The current detection resistance element (3) and the threshold value setting resistance element (8) are resistance elements having the same temperature characteristic, according to claim 1. Overcurrent detection circuit.