JP5763970B2 - Voltage detection circuit - Google Patents

Description

  Various technical features disclosed in the present specification relate to a semiconductor device having no reference voltage source.

  Conventionally, many semiconductor devices have a reference voltage source (such as a bandgap power supply circuit) that generates a predetermined reference voltage from an external power supply voltage, and an operation setting (regulator circuit) inside the semiconductor device using the output thereof. Output target value setting, abnormality protection circuit detection threshold setting, etc.). However, in order to reduce the circuit scale as much as possible depending on the use of the semiconductor device, a model that does not have the reference voltage source has been proposed.

  Note that Patent Documents 1 to 6 can be cited as examples of conventional techniques related to various technical features disclosed in the present specification.

Japanese Patent Laid-Open No. 10-201080 Japanese Patent No. 2006-351944 JP-A-5-087840 Japanese Patent No. 2004-309164 JP-A-5-243858 Japanese Patent No. 9-130157

<Series regulator circuit>
FIG. 10 is a circuit diagram showing a conventional example of a series regulator circuit. The series regulator circuit X of the conventional example does not require a separate reference voltage source for setting the target value of the output voltage Vout. However, since the npn type bipolar transistor is used as the output transistor XC in the conventional series regulator circuit X, only the output voltage Vout lower than the base-emitter voltage Vf of the output transistor XC from the input voltage Vin is generated at most. There was a problem that it was not possible. In addition, the series regulator circuit X of the conventional example has a problem that the chip area is increased because the booster XB is required to increase the gain of the amplifier XA.

  Note that the pnp bipolar transistor or P-channel MOS [Metal Oxide Semiconductor] field effect transistor is used as the output transistor XC, the booster XB is removed, and the amplifier XA and the output transistor XC are directly connected. It seems that the problem can be solved easily. However, in such a configuration, the amplifier XA does not operate when the output voltage Vout is not output, and the series regulator circuit X cannot be started, and thus cannot be a practical solution.

  In view of the above problems, the first technical feature disclosed in the present specification does not require a separate reference voltage source for setting the output target value, and expands the output voltage range and increases the circuit scale. It is a first object of the present invention to provide a regulator circuit capable of realizing reduction.

<Temperature protection circuit (temperature detection circuit)>
11A and 11B are circuit diagrams illustrating a first conventional example and a second conventional example of a temperature protection circuit, respectively.

  The temperature protection circuit Y1 of the first conventional example easily generates the temperature protection signal TSD by utilizing the negative temperature characteristic (−2 mV / ° C.) of the base-emitter voltage Vf of the npn bipolar transistor YA. can do. However, the temperature protection circuit Y1 of the first conventional example has a problem that a separate reference voltage source REF is required to set the detection threshold (threshold voltage Vth).

  The temperature protection circuit Y2 of the second conventional example does not require a separate reference voltage source for setting the detection threshold. However, in the temperature protection circuit Y2 of the second conventional example, the number of elements constituting the circuit is larger than that of the first conventional example, and the current consumption (particularly the current consumption in the comparators YH and YL and the logic circuit LOGIC) is large. There was a problem of becoming larger.

  The second technical feature disclosed in the present specification, in view of the above-described problems, does not require a separate reference voltage source for setting the detection threshold, and reduces the circuit scale and current consumption. It is a second object to provide a temperature detection circuit that can be realized.

<Overvoltage protection circuit (voltage detection circuit)>
FIG. 12 is a circuit diagram showing a conventional example of an overvoltage protection circuit. The overvoltage protection circuit Z of the conventional example can easily generate the overvoltage protection signal OVP by comparing the monitor voltage Vm corresponding to the output voltage Vout with a predetermined reference voltage Vth by the comparator ZA. However, the conventional overvoltage protection circuit Z has a problem that a separate reference voltage source is required to set the detection threshold (reference voltage Vth).

  A third technical feature disclosed in the present specification is to provide a voltage detection circuit that does not require a separate reference voltage source for setting a detection threshold in view of the above problems. To do.

<First technical features>
Among various technical features disclosed in the present specification, a regulator circuit according to a first technical feature is a pnp type connected between an input voltage application terminal and an output voltage application terminal, or A P-channel output transistor, a feedback voltage generation circuit that generates a feedback voltage corresponding to the output voltage, and a collector current that flows through each transistor upon receiving the feedback voltage at a common base of transistor pairs having different emitter areas. A configuration having an amplifier circuit that outputs a voltage signal corresponding to a magnitude relationship, and a drive signal generation circuit that amplifies the voltage signal by a grounded emitter type or a source grounded type amplification stage to generate a drive signal of the output transistor (1-1 configuration).

  In the regulator circuit having the above-described configuration 1-1, the drive signal generation circuit is connected between an input voltage application terminal and a control electrode of the output transistor, and the voltage signal is applied to the control electrode. And a pnp-type or P-channel type amplification transistor; a current source connected between the control electrode of the output transistor and a ground terminal (1-2 configuration).

  Further, in the regulator circuit having the above configuration 1-2, the drive signal generation circuit further includes a pull-up resistor connected between the input voltage application terminal and the control electrode of the output transistor. (Configuration 1-3) may be used.

  Further, in the regulator circuit having the above-described configuration 1-3, in the amplifier circuit, as the transistor pair, a base is connected to an application end of the feedback voltage, and a collector is applied with the input voltage via a first load. A first transistor having an emitter area larger than that of the first transistor and having a base connected to the ground terminal through a first resistor; The first transistor is connected to the base of the first transistor, the collector is connected to the application terminal of the input voltage via the second load, and the emitter is passed through the second resistor having a resistance value smaller than that of the first resistor. And a second transistor connected to the ground terminal via a resistor.

  In the regulator circuit having the above-described configuration 1-4, the amplifier circuit further has a higher breakdown voltage than the first transistor, and the third circuit is inserted between the first transistor and the first load. It is preferable to adopt a configuration (first to fifth configuration) including: a transistor; and a fourth transistor having a higher withstand voltage than the second transistor and inserted between the second transistor and the second load.

  In the regulator circuit having the first to fifth configurations, the control electrodes of the third transistor and the fourth transistor are both connected to the output voltage application terminal (first to sixth configurations). ).

  Of the various technical features disclosed in this specification, the semiconductor device according to the first technical feature includes a regulator circuit having any one of the first to first to sixth configurations. It is set as the structure which it has (1-7th structure).

<Second technical feature>
Among various technical features disclosed in the present specification, a temperature detection circuit according to a second technical feature includes a reference current generation unit that generates a reference current, and a first current corresponding to the reference current. And a current mirror that generates a second current, a first resistor inserted in a path through which the reference current flows downstream of the current mirror, and a path through which the first current flows downstream of the current mirror. A temperature detection signal is generated by comparing the first current and the second current with a second resistor and a diode or a diode-connected transistor inserted in a path through which the second current flows downstream of the current mirror And a current comparison unit (second configuration).

  In the temperature detection circuit having the above-described configuration 2-1, the first resistor and the second resistor may be configured to have the same resistance value (the configuration 2-2).

  The temperature detection circuit having the above-described configuration 2-2 includes a third resistor connected between the first resistor, the second resistor, and a ground terminal, and the third resistor according to the temperature detection signal. It is good to make it the structure (2-3 structure) which further has a switch which conducts / cuts off between both ends of resistance.

  In the temperature detection circuit having the above configuration 2-2 or 2-3, the current mirror has a first electrode connected to a ground terminal via the first resistor, and a second electrode connected to the reference current. A first transistor connected to the generator and having a control electrode connected to its second electrode; the first electrode connected to the ground terminal via the second resistor; and the second electrode connected to the current comparator A second transistor in which a control electrode is connected to a control electrode of the first transistor; a first electrode is connected to a ground terminal via the diode or the diode-connected transistor, and a second electrode is connected to the current comparison unit And a third transistor in which the control electrode is connected to the control electrode of the first transistor (configuration 2-4).

  Further, in the temperature detection circuit having the above configuration 2-4, the current comparison unit includes a first electrode connected to an input voltage application terminal, and a second electrode connected to a second electrode of the second transistor. A fourth transistor having a control electrode connected to its second electrode; a first electrode connected to the input voltage application terminal; a second electrode connected to the second electrode of the third transistor; A fifth transistor connected to the control electrode of the fourth transistor; a first electrode connected to the input terminal of the input voltage; a second electrode connected to an output terminal of the temperature detection signal; A sixth transistor connected to the second electrode of the fifth transistor; and a fourth resistor connected between the output terminal of the temperature detection signal and the ground terminal (a second to fifth structure). Good.

  Of the various technical features disclosed in the present specification, the semiconductor device according to the second technical feature is a temperature detection circuit having any one of the above-described configurations of 2-1 to 2-5. And an internal circuit whose operation is controlled based on the temperature detection signal (configuration 2-6).

<Third technical features>
Among various technical features disclosed in the present specification, a voltage detection circuit according to a third technical feature is a voltage dividing circuit that divides a monitoring target voltage to generate a first voltage and a second voltage. And a comparison circuit that generates a voltage detection signal by comparing the first voltage and the second voltage input to the transistor pair having different emitter areas (configuration 3-1). .

  In the voltage detection circuit having the above configuration 3-1, the comparison circuit includes, as the transistor pair, a first transistor having a base connected to the application terminal of the first voltage; and more than the first transistor; A configuration in which the emitter area is large, the base is connected to the application terminal of the second voltage, and the emitter is connected in common with the emitter of the first transistor (configuration 3-2). Good.

  In the voltage detection circuit having the above configuration 3-2, the comparison circuit further includes a current source that generates emitter currents of the first transistor and the second transistor; and the first transistor or the second transistor. And an output stage for generating the voltage detection signal from the collector voltage of the transistor (configuration 3-3).

  Further, in the voltage detection circuit having any one of the above configurations 3-1 to 3-3, the voltage dividing circuit is a first voltage dividing circuit that generates a reference voltage between the monitored voltage and the ground voltage. And a second voltage dividing circuit that generates the first voltage and the second voltage between the monitoring target voltage and the reference voltage or between the reference voltage and the ground voltage (third to fourth Configuration).

  In the voltage detection circuit having the above configuration 3-4, in the second voltage dividing circuit, a control electrode is connected to an application terminal of the reference voltage, and a second electrode is applied to a ground terminal or the monitoring target voltage. A resistor ladder connected between the second electrode of the third transistor and the application end of the monitoring target voltage or the ground end to generate the first voltage and the second voltage; It is good to make it the structure (3-5 structure) containing these.

  Among various technical features disclosed in the present specification, the semiconductor device according to the third technical feature is a voltage detection circuit having any one of the above-described configurations of 3-1 to 3-5. And an internal circuit whose operation is controlled based on the voltage detection signal (configuration 3-6).

  According to the first technical feature disclosed in the present specification, a separate reference voltage source is not required for setting the output target value, and the output voltage range and the circuit scale are reduced. It is possible to provide a regulator circuit that can be used.

  Further, according to the second technical feature disclosed in the present specification, a separate reference voltage source is not required for setting the detection threshold, and a reduction in circuit scale and a reduction in current consumption are realized. It is possible to provide a temperature detection circuit that can perform this.

  Further, according to the third technical feature disclosed in the present specification, it is possible to provide a voltage detection circuit that does not require a separate reference voltage source for setting the detection threshold.

Block diagram showing one configuration example of series regulator IC Circuit diagram showing first configuration example of series regulator circuit Circuit diagram showing second configuration example of series regulator circuit Circuit diagram showing an example of the configuration of an abnormality protection unit included in a series regulator circuit Circuit diagram showing first configuration example of temperature protection circuit Circuit diagram showing a second configuration example of the temperature protection circuit Circuit diagram showing a first configuration example of an overvoltage protection circuit Circuit diagram showing a second configuration example of the overvoltage protection circuit Diagram for explaining points to consider when sharing the feedback resistance of a series regulator circuit with an overvoltage protection circuit (first configuration example) Diagram for explaining points to consider when sharing the feedback resistance of a series regulator circuit with an overvoltage protection circuit (second configuration example) Circuit diagram showing a conventional example of a series regulator circuit Circuit diagram showing a first conventional example of a temperature protection circuit Circuit diagram showing second conventional example of temperature protection circuit Circuit diagram showing a conventional example of an overvoltage protection circuit

<Overall configuration>
FIG. 1 is a block diagram illustrating a configuration example of a series regulator IC. The series regulator IC 1 of this configuration example is a monolithic semiconductor integrated circuit device in which a series regulator circuit 100, a temperature protection circuit 200, and an overvoltage protection circuit 300 are integrated.

  The series regulator circuit 100 generates a desired output voltage Vout (for example, 5V or 3.3V) from the input voltage Vin (for example, 12V). The series regulator circuit 100 includes an abnormality protection unit that controls whether or not an output operation is possible based on the temperature protection signal S1 and the overvoltage protection signal S2. The configuration and operation of the series regulator circuit 100 will be described in detail later.

  The temperature protection circuit 200 monitors whether the chip temperature Tj of the series regulator IC1 exceeds a predetermined detection threshold value, generates a temperature protection signal S1, and outputs the temperature protection signal S1 to the series regulator circuit 100. The configuration and operation of the temperature protection circuit 200 will be described in detail later.

  The overvoltage protection circuit 300 generates an overvoltage protection signal S2 by monitoring whether or not the output voltage Vout exceeds a predetermined detection threshold, and outputs the overvoltage protection signal S2 to the series regulator circuit 100. The configuration and operation of the overvoltage protection circuit 300 will be described in detail later.

<Series regulator circuit>
FIG. 2 is a circuit diagram illustrating a first configuration example of the series regulator circuit 100. The series regulator circuit 100 of this configuration example includes pnp bipolar transistors 101 to 103, npn bipolar transistors 104 and 105, a P-channel MOS field effect transistor 106, resistors 107 to 111, and a current source 112. Have.

  The emitters of the transistors 101 and 102 are both connected to the application terminal for the input voltage Vin. The bases of the transistors 101 and 102 are both connected to the collector of the transistor 102. The collector of the transistor 101 is connected to the collector of the transistor 104. The collector of the transistor 102 is connected to the collector of the transistor 105. The bases of the transistors 104 and 105 are both connected to the application terminal of the feedback voltage Vfb (a connection node between the resistor 110 and the resistor 111). The emitter of the transistor 105 is connected to the ground terminal via resistors 107 and 108. The emitter of the transistor 104 is connected to a connection node between the resistor 107 and the resistor 108. The emitter area ratio of the transistors 104 and 105 is designed to be 1: N.

  The emitter of the transistor 103 is connected to the application terminal for the input voltage Vin. The collector of the transistor 103 is connected to the ground terminal via the current source 112. The base of the transistor 103 is connected to the collector of the transistor 101. The source of the transistor 106 is connected to the application terminal for the input voltage Vin. The drain of the transistor 106 is connected to the application terminal of the output voltage Vout, and is also connected to the ground terminal via the resistors 110 and 111. The gate of the transistor 106 is connected to the collector of the transistor 103 and is also connected to the application terminal of the input voltage Vin via the resistor 109.

  The transistor 106 corresponds to an output transistor connected between the application terminal of the input voltage Vin and the application terminal of the output voltage Vout. The resistors 110 and 111 correspond to a feedback voltage generation circuit that generates a feedback voltage Vfb (a divided voltage of the output voltage Vout) corresponding to the output voltage Vout. Transistors 101 and 102, transistors 104 and 105, and resistors 107 and 108 are collectors that receive the feedback voltage Vfb at the common base of transistor pairs (transistors 104 and 105) having different emitter areas and flow to the transistors 104 and 105, respectively. This corresponds to an amplifier circuit that outputs a voltage signal V11 according to the magnitude relationship between the currents I11 and I12. The transistor 103, the resistor 109, and the current source 112 amplify the voltage signal V 11 by an amplification stage of a common emitter type (a common source type when a P-channel MOS field effect transistor is used as the transistor 103). This corresponds to a drive signal generation circuit for generating the drive signal V12 (gate voltage).

  The operation of the series regulator circuit 100 having the above configuration will be described. When the feedback voltage Vfb increases as the output voltage Vout increases, the difference between the base-emitter voltages is more dominant between the transistor 104 and the transistor 105 than the difference between the emitter areas. As a result, the collector current I11 becomes larger than the collector current I12, and the voltage signal V11 decreases. When the conductivity of the transistor 103 increases as the voltage signal V11 decreases, the voltage value of the drive signal V12 increases. As a result, the conductivity of the transistor 106 is reduced, and the output voltage Vout is reduced.

  On the other hand, when the feedback voltage Vfb decreases as the output voltage Vout decreases, the difference between the emitter areas of the transistors 104 and 105 becomes more dominant than the difference between the base-emitter voltages. . As a result, the collector current I12 becomes larger than the collector current I11, and the voltage signal V11 increases. When the conductivity of the transistor 103 decreases as the voltage signal V11 increases, the voltage value of the drive signal V12 decreases. As a result, the conductivity of the transistor 106 increases and the output voltage Vout increases.

  Thus, with the series regulator circuit 100 of the first configuration example, a desired output voltage Vout can be generated from the input voltage Vin without requiring a separate reference voltage source for setting the output target value. Note that the target value of the output voltage Vout can be arbitrarily adjusted according to the resistance ratio of the resistors 110 and 111.

  In the case of the series regulator circuit 100 of the first configuration example, a grounded-emitter (or grounded-source) amplifier stage having a circuit scale smaller than that of the booster XB in FIG. 10 can be used. A reduction in current consumption can be realized.

  Further, in the case of the series regulator circuit 100 of the first configuration example, the P-channel (or pnp) transistor 106 can be used as the output transistor, so that the output range of the output voltage Vout is expanded to the vicinity of the input voltage Vin. Is possible.

  In the case of the series regulator circuit 100 of the first configuration example, the amplifier circuit (the transistors 101 and 102, the transistors 104 and 105, and the resistors 107 and 108) cannot operate when the output voltage Vout is not output. However, since the drive signal V12 is pulled down to a low level via the current source 112, the transistor 106 is turned on. As a result, the output voltage Vout increases and the series regulator circuit 100 can be normally started.

  In the case of the series regulator circuit 100 of the first configuration example, the gate of the transistor 106 is pulled up to the application terminal of the input voltage Vin via the resistor 109. Therefore, in a situation where the logic level of the drive signal V12 can be indefinite, the logic level of the drive signal V12 can be fixed to a high level and the transistor 106 can be maintained in an off state.

  If the above configuration is applied, it is possible to realize not only the series regulator circuit 100 but also an amplifier or a comparator used for other purposes.

  FIG. 3 is a circuit diagram illustrating a second configuration example of the series regulator circuit 100. The second configuration example is substantially the same configuration as the first configuration example, in that an N-channel MOS field effect transistor 113 is inserted between the transistor 101 and the transistor 104, and between the transistor 102 and the transistor 105. This is characterized in that an N channel type MOS field effect transistor 114 is inserted into the N channel type MOS field effect transistor 114. As the transistors 113 and 114, an element (for example, NDMOSFET) having a higher breakdown voltage (for example, 50V breakdown voltage) than the transistors 104 and 105 may be used.

  The above connection relationship will be specifically described. The drain of the transistor 113 is connected to the collector of the transistor 101. The source of the transistor 113 is connected to the collector of the transistor 104. The drain of the transistor 114 is connected to the collector of the transistor 102. The source of the transistor 114 is connected to the collector of the transistor 105. The gates of the transistors 113 and 114 are both connected to the application terminal for the output voltage Vout. Note that the gates of the transistors 113 and 114 are other than the application terminal of the output voltage Vout as long as the operation of the amplifier circuit (the transistors 101 and 102, the transistors 104 and 105, the resistors 107 and 108, and the transistors 113 and 114) is possible. It may be connected to the low voltage application terminal.

  Since the transistors 113 and 114 are inserted, only the voltage (= Vout−Vgs) lower than the output voltage Vout by the gate-source voltage Vgs of the transistors 113 and 114 is applied to the collectors of the transistors 104 and 105. Therefore, even if it is difficult to increase the breakdown voltage of the transistors 104 and 105 in the manufacturing process, it is possible to achieve a high breakdown voltage for the entire series regulator circuit 100.

  FIG. 4 is a circuit diagram illustrating a configuration example of the abnormality protection unit included in the series regulator circuit 100. The abnormality protection unit of this configuration example includes a P-channel MOS field effect transistor 115, an N-channel MOS field effect transistor 116, and a negative OR calculator 117.

  The source of the transistor 115 is connected to the application terminal for the input voltage Vin. The drain of the transistor 115 is connected to the gate of the transistor 106. The gate of the transistor 115 is connected to the output terminal of the NOR circuit 117. The first input terminal of the negative OR calculator 117 is connected to the application terminal of the temperature protection signal S1. The second input terminal of the negative OR calculator 117 is connected to the application terminal for the overvoltage protection signal S2. The drain of the transistor 116 is connected to the application terminal for the output voltage Vout. The source of the transistor 116 is connected to the ground terminal. The gate of the transistor 116 is connected to the application end of the temperature protection signal S1.

  When both the temperature protection signal S1 and the overvoltage protection signal S2 are at a low level (normal logic level), both the transistors 115 and 116 are turned off. Therefore, there is no influence on the operation of generating the output voltage Vout. When the temperature protection signal S1 is at a high level (logic level when an abnormal temperature is detected), both the transistors 115 and 116 are turned on. Therefore, the transistor 106 is forcibly turned off, and the output voltage Vout is pulled down to the ground voltage. When the overvoltage protection signal S2 is at a high level (logic level when overvoltage is detected), only the transistor 115 is turned on while the transistor 116 remains off. Therefore, the output voltage Vout is not lowered to the ground voltage, and only the transistor 106 is forcibly turned off.

<Temperature protection circuit (temperature detection circuit)>
FIG. 5 is a circuit diagram illustrating a first configuration example of the temperature protection circuit 200. The temperature protection circuit 200 of the first configuration example includes pnp bipolar transistors 201 to 205, npn bipolar transistors 206 to 211, and resistors 212 to 216.

  The emitters of the transistors 201 and 202 are both connected to the application terminal for the input voltage Vin. The bases of the transistors 201 and 202 are both connected to the collector of the transistor 201. The collector of the transistor 201 is connected to the collector of the transistor 207. The bases of the transistors 206 and 207 are both connected to the collector of the transistor 206. The collector of the transistor 206 is connected to the application terminal of the input voltage Vin via the resistor 212. The emitter of the transistor 206 is connected to the ground terminal. The emitter of the transistor 207 is connected to the ground terminal via the resistor 213. The emitter area ratio of the transistors 206 and 207 is designed to be 1: N.

  The collector of the transistor 202 is connected to the collector of the transistor 208. The bases of the transistors 208 to 210 are all connected to the collector of the transistor 208. The emitter of the transistor 208 is connected to the ground terminal via the resistor 214. The emitter of the transistor 209 is connected to the ground terminal via the resistor 215. The emitter of the transistor 210 is connected to the collector and base of the transistor 211. The emitter of the transistor 211 is connected to the ground terminal.

  The collector of the transistor 209 is connected to the collector of the transistor 203. The collector of the transistor 210 is connected to the collector of the transistor 204 and the base of the transistor 205. The emitters of the transistors 203 and 204 are both connected to the application terminal for the input voltage Vin. The bases of the transistors 203 and 204 are both connected to the collector of the transistor 203. The emitter of the transistor 205 is connected to the application terminal for the input voltage Vin. The collector of the transistor 205 is connected to the output terminal of the temperature protection signal S1, and is also connected to the ground terminal via the resistor 216.

  The transistors 201 and 202, the transistors 206 and 207, and the resistors 212 and 213 correspond to a reference current generation unit that generates the reference current I20. The transistors 208 to 210 correspond to a current mirror that generates the first current I21 and the second current I22 in accordance with the reference current I20. The resistor 214 corresponds to a first resistor inserted in a path through which the reference current I20 flows on the downstream side (ground end side) of the current mirror. The resistor 215 corresponds to a second resistor inserted in a path through which the first current I21 flows on the downstream side of the current mirror. The transistor 211 corresponds to a diode-connected transistor inserted in a path through which the second current I21 flows on the downstream side of the current mirror. Note that the transistor 211 can be replaced with a diode. The transistors 203 to 205 and the resistor 216 correspond to a current comparison unit that compares the first current I21 and the second current I22 to generate the temperature detection signal S1.

The operation of the temperature protection circuit 200 having the above configuration will be described. When the resistance value of the resistor 213 is R21 and the resistance values of the resistors 214 and 215 are both R22, the emitter voltage V21 of the transistor 209 is expressed by the following equation (1). Note that (1) the sign _{V T} in the formula is a thermal voltage (= kT / q) (k : Boltzmann constant, T: absolute temperature, q: electron charge).
V21 = (R22 / R21) V _T lnN (1)

On the other hand, the emitter voltage V22 of the transistor 210 is expressed by the following equation (2). In addition, the code | symbol Vf in (1) Formula is the base-emitter voltage of the diode connection type transistor 211, and has a negative temperature characteristic (-2 mV / degreeC).
V22 = Vf (2)

  The emitter voltages V21 and V22 are reversed in magnitude from each other according to the chip temperature Tj. In the temperature protection circuit 200 of the first configuration example, the chip temperature Tj is detected using this characteristic.

  When the chip temperature Tj is lower than the temperature abnormality detection threshold and the emitter voltage V22 of the transistor 210 is higher than the emitter voltage V21 of the transistor 209, the transistor 210 is turned off and the second current I22 cannot flow. Accordingly, the base voltage V23 of the transistor 205 is at a high level, and the transistor 205 is turned off. As a result, the temperature protection signal S1 becomes low level (normal logic level) via the resistor 216.

  On the other hand, when the chip temperature Tj rises and the emitter voltage V22 of the transistor 210 becomes lower than the emitter voltage V21 of the transistor 209, the transistor 210 is turned on and the second current I22 can flow. . Accordingly, the base voltage V23 of the transistor 205 is at a low level, and the transistor 205 is turned on. As a result, the temperature protection signal S1 becomes high level (logic level when temperature abnormality is detected) via the transistor 205.

  As described above, in the case of the temperature protection circuit 200 of the first configuration example, the temperature protection signal S1 is generated by monitoring the chip temperature Tj without requiring a separate reference voltage source for setting the temperature abnormality detection threshold. Can do. Note that the temperature abnormality detection threshold can be arbitrarily adjusted in accordance with the emitter area ratio of the transistors 206 and 207 and the resistance ratio between the resistor 213 and the resistors 214 and 215.

  Further, with the temperature protection circuit 200 of the first configuration example, it is possible to reduce the circuit scale and the current consumption as compared with the conventional configuration of FIG. 11B.

  Note that the temperature protection circuit 200 of the first configuration example can be mounted not only on the series regulator IC 1 but also on other power ICs.

  FIG. 6 is a circuit diagram showing a second configuration example of the temperature protection circuit 200. The second configuration example is substantially the same configuration as the first configuration example, and is characterized in that a resistor 217, an N-channel MOS field effect transistor 218, and an inverter 219 are added.

  A first end of the resistor 217 is connected to the resistors 214 and 215. A second end of the resistor 217 is connected to the ground end. The drain of the transistor 218 is connected to the first end of the resistor 217. The source of the transistor 218 is connected to the second end of the resistor 217. The gate of the transistor 218 is connected to the output terminal of the inverter 219. The input terminal of the inverter 219 is connected to the output terminal of the temperature protection signal S1.

  The transistor 218 corresponds to a switch that conducts / cuts off between both ends of the resistor 217 in accordance with the temperature detection signal S1.

  When the temperature protection signal S1 is at a low level (normal logic level), the transistor 218 is turned on, and both ends of the resistor 217 are short-circuited. On the other hand, when the temperature protection signal S1 is at the high level (the logic level when the temperature abnormality is detected), the transistor 218 is turned off, and the resistor 217 is inserted between the resistors 214 and 215 and the ground terminal. As a result, the emitter voltage V21 of the transistor 209 is raised, so that it is difficult for the temperature protection signal S1 to return to the low level (normal logic level) even if the chip temperature Tj is lowered.

  Thus, with the temperature protection circuit 200 of the second configuration example, it is possible to impart hysteresis characteristics to the temperature protection signal S1 with a very simple configuration.

<Overvoltage protection circuit (voltage detection circuit)>
FIG. 7 is a circuit diagram showing a first configuration example of the overvoltage protection circuit 300. The overvoltage protection circuit 300 of the first configuration example includes pnp bipolar transistors 301 to 304, npn bipolar transistors 305 and 306, resistors 307 to 312, and a current source 313.

  A first terminal of the resistor 307 is connected to an application terminal for the output voltage Vout. The second end of the resistor 307 is connected to the first end of the resistor 308 and the base of the transistor 303. A second terminal of the resistor 308 is connected to the ground terminal. A first terminal of the resistor 309 is connected to an application terminal for the output voltage Vout. The second end of the resistor 309 is connected to the first end of the resistor 310 and the base of the transistor 305. The second end of the resistor 310 is connected to the first end of the resistor 311 and the base of the transistor 306. A second end of the resistor 311 is connected to the emitter of the transistor 303. The collector of the transistor 303 is connected to the ground terminal.

  The emitters of the transistors 301 and 302 are both connected to the application terminal for the output voltage Vout. The bases of the transistors 301 and 302 are both connected to the collector of the transistor 302. The collector of the transistor 301 is connected to the collector of the transistor 305 and the base of the transistor 304. The collector of the transistor 302 is connected to the collector of the transistor 306. The emitters of the transistors 305 and 306 are both connected to the ground terminal via the current source 313. The emitter of the transistor 304 is connected to the application terminal for the output voltage Vout. The collector of the transistor 304 is connected to the output terminal of the overvoltage protection signal S2, and is also connected to the ground terminal via the resistor 312. The emitter area ratio of the transistors 305 and 306 is designed to be 1: N.

  The transistor 303 and the resistors 307 to 311 correspond to a voltage dividing circuit that divides the output voltage Vout, which is a monitoring target voltage, to generate the first voltage V31 and the second voltage V32. In particular, the resistors 307 and 308 correspond to a first voltage dividing circuit that generates the reference voltage V30 between the output voltage Vout and the ground voltage. The transistor 303 and the resistor ladder (resistors 309 to 311) are connected to the output voltage Vout and the reference voltage V30 (more precisely, the emitter voltage V33 (= V30 + Vf) of the transistor 303 which is higher than the reference voltage V30 by the base-emitter voltage Vf of the transistor 303. )) Corresponds to a second voltage dividing circuit for generating the first voltage V31 and the second voltage V32.

  The transistors 301 and 302, the transistors 304 to 306, the resistor 312, and the current source 313 are overvoltages by comparing the first voltage V31 and the second voltage V32 input to the transistor pair (transistors 305 and 306) having different emitter areas. This corresponds to a comparison circuit that generates the protection signal S2. In particular, the transistor 304 and the resistor 312 correspond to an output stage that generates the overvoltage protection signal S2 from the collector voltage V34 of the transistor 301.

  As described above, the overvoltage protection circuit 300 of the first configuration example gives an input offset to the comparison circuit by giving a difference to the emitter areas of the transistors 305 and 306 forming the input stage of the comparison circuit. It is configured to detect whether or not the output voltage Vout is in an overvoltage state.

  The operation of the overvoltage protection circuit 300 having the above configuration will be described. When the output voltage Vout is lower than the overvoltage detection threshold, the voltage difference between the output voltage Vout and the emitter voltage V33 does not open, so the voltage difference between the first voltage V31 and the second voltage V32 also decreases. Therefore, between the transistors 305 and 306, the difference between the emitter areas is more dominant than the difference between the base-emitter voltages (that is, the voltage difference between the first voltage V31 and the second voltage V32). It becomes. As a result, the collector current I32 of the transistor 306 becomes larger than the collector current I31 of the transistor 305, the base voltage V34 of the transistor 304 becomes high level, and the transistor 304 is turned off. As a result, the overvoltage protection signal S2 becomes a low level (normal logic level) via the resistor 312.

  On the other hand, when the output voltage Vout rises and the voltage difference between the output voltage Vout and the emitter voltage V33 opens, the voltage difference between the first voltage V31 and the second voltage V32 also increases. Therefore, between the transistor 305 and the transistor 306, the difference between the base-emitter voltages (that is, the voltage difference between the first voltage V31 and the second voltage V32) is more dominant than the difference between the emitter areas. It becomes. As a result, the collector current I32 of the transistor 306 becomes smaller than the collector current I31 of the transistor 305, the base voltage V34 of the transistor 304 becomes low level, and the transistor 304 is turned on. As a result, the overvoltage protection signal S2 becomes high level (logic level when overvoltage is detected) via the transistor 304.

  As described above, the overvoltage protection circuit 300 of the first configuration example can monitor the output voltage Vout and generate the overvoltage protection signal S2 without requiring a separate reference voltage source for setting the overvoltage detection threshold. it can. The overvoltage detection threshold can be arbitrarily adjusted according to the emitter area ratio of the transistors 305 and 306 and the voltage dividing ratio of the resistors 307 to 311 (particularly the voltage dividing ratio of the resistors 307 and 308).

Further, in the case of the overvoltage protection circuit 300 of the first configuration example, by appropriately adjusting the resistance values of the resistors 309 to 311, the input offset (= V _T lnN) of the comparison circuit using the transistors 305 and 306 as the input stage can be obtained. It is possible to cancel out the mutual temperature characteristics (both positive temperature characteristics) with the input signals (V31-V32) of the comparison circuit, thereby reducing the temperature dependence of the overvoltage protection signal S2. It becomes possible to do.

  Further, in the case of the overvoltage protection circuit 300 of the first configuration example, the overvoltage protection signal S2 can be generated by monitoring the output voltage Vout using a comparison circuit having higher responsiveness than the series regulator circuit 100. It is possible to appropriately suppress the overshoot of the output voltage Vout caused by the steep fluctuation of the input voltage Vin. Since components with a low breakdown voltage (such as a microcomputer) are often connected to the subsequent stage of the series regulator IC 1, the reliability of the set in which the series regulator IC 1 is mounted can be realized by suppressing the overshoot of the output voltage Vout. It becomes possible to raise.

  Further, with the overvoltage protection circuit 300 of the first configuration example, it is possible to realize a reduction in circuit scale and a reduction in current consumption as compared with the conventional configuration in FIG.

  The overvoltage protection signal S2 generated by the overvoltage protection circuit 300 of the first configuration example can be used not only for the overshoot suppression operation of the output voltage Vout but also for the reset operation of the series regulator IC1 when the overvoltage is abnormal. It is.

  FIG. 8 is a circuit diagram showing a second configuration example of the overvoltage protection circuit 300. In the second configuration example, the polarity is reversed from that of the first configuration example. The npn-type bipolar transistors 321 to 324, the pnp-type bipolar transistors 325 and 326, the resistors 327 to 332, the current source 333, And an inverter 334.

  A first terminal of the resistor 327 is connected to the ground terminal. The second end of the resistor 327 is connected to the first end of the resistor 328 and the base of the transistor 323. A second terminal of the resistor 328 is connected to an application terminal for the output voltage Vout. A first terminal of the resistor 329 is connected to the ground terminal. The second end of the resistor 329 is connected to the first end of the resistor 330 and the base of the transistor 325. The second end of the resistor 330 is connected to the first end of the resistor 331 and the base of the transistor 326. A second terminal of the resistor 331 is connected to the emitter of the transistor 323. The collector of the transistor 323 is connected to the application terminal for the output voltage Vout.

  The emitters of the transistors 321 and 322 are both connected to the ground terminal. The bases of the transistors 321 and 322 are both connected to the collector of the transistor 322. The collector of the transistor 321 is connected to the collector of the transistor 325 and the base of the transistor 324. The collector of the transistor 322 is connected to the collector of the transistor 326. The emitters of the transistors 325 and 326 are both connected to the application terminal of the output voltage Vout via the current source 333. The emitter of the transistor 324 is connected to the ground terminal. The collector of the transistor 324 is connected to the output terminal of the overvoltage protection signal S2 through the inverter 334 and is also connected to the application terminal of the output voltage Vout through the resistor 332. The emitter area ratio of the transistors 325 and 326 is designed to be 1: N.

  The transistor 323 and the resistors 327 to 331 correspond to a voltage dividing circuit that divides the output voltage Vout, which is a monitoring target voltage, to generate the first voltage V31 and the second voltage V32. In particular, the resistors 327 and 328 correspond to a first voltage dividing circuit that generates the reference voltage V30 between the output voltage Vout and the ground voltage. The transistor 323 and the resistor ladder (resistors 329 to 331) are connected to the reference voltage V30 (more precisely, the emitter voltage V33 of the transistor 323 (= V30−Vf) lower than the reference voltage V30 by the base-emitter voltage Vf of the transistor 323). This corresponds to a second voltage dividing circuit that generates the first voltage V31 and the second voltage V32 between the ground voltages.

  The transistors 321 and 322, the transistors 324 to 326, the resistor 332, the current source 333, and the inverter 334 compare the first voltage V31 and the second voltage V32 input to the transistor pair (transistors 325 and 326) having different emitter areas. This corresponds to a comparison circuit that generates the overvoltage protection signal S2. In particular, the transistor 324, the resistor 332, and the inverter 334 correspond to an output stage that generates the overvoltage protection signal S2 from the collector voltage V34 of the transistor 321.

  As described above, the overvoltage protection circuit 300 of the second configuration example gives an input offset to the comparison circuit by giving a difference to the emitter areas of the transistors 325 and 326 forming the input stage of the comparison circuit. It is configured to detect whether or not the output voltage Vout is in an overvoltage state.

  The operation of the overvoltage protection circuit 300 having the above configuration will be described. When the output voltage Vout is lower than the overvoltage detection threshold, the voltage difference between the emitter voltage V33 and the ground voltage does not open, so the voltage difference between the first voltage V31 and the second voltage V32 also decreases. Therefore, between the transistors 325 and 326, the difference between the emitter areas is more dominant than the difference between the base-emitter voltages (that is, the voltage difference between the first voltage V31 and the second voltage V32). It becomes. As a result, the collector current I32 of the transistor 326 is larger than the collector current I31 of the transistor 325, the base voltage V34 of the transistor 324 becomes low level, and the transistor 324 is turned off. As a result, the input signal to the inverter 334 becomes high level via the resistor 312, and the overvoltage detection signal S2 obtained by logically inverting this input signal becomes low level (normal logic level).

  On the other hand, when the output voltage Vout rises and the voltage difference between the emitter voltage V33 and the ground voltage opens, the voltage difference between the first voltage V31 and the second voltage V32 also increases. Therefore, between the transistor 325 and the transistor 326, the difference between the base-emitter voltages (that is, the voltage difference between the first voltage V31 and the second voltage V32) is more dominant than the difference between the emitter areas. It becomes. As a result, the collector current I32 of the transistor 326 becomes smaller than the collector current I31 of the transistor 325, the base voltage V34 of the transistor 324 becomes high level, and the transistor 324 is turned on. As a result, the input signal to the inverter 334 becomes low level via the transistor 324, and the overvoltage protection signal S2 obtained by logically inverting this input signal becomes high level (logic level when overvoltage is detected).

  Thus, the overvoltage protection circuit 300 of the second configuration example can achieve the same effects as those of the first configuration example described above. Note that the overvoltage detection threshold can be arbitrarily adjusted according to the emitter area ratio of the transistors 325 and 326 and the voltage division ratio of the resistors 327 to 331 (particularly, the voltage division ratio of the resistors 327 and 328).

  FIGS. 9A and 9B are diagrams for explaining points to be noted when the feedback resistors 110 and 111 of the series regulator circuit 100 are shared by the overvoltage protection circuit 300, respectively. In FIG. 9A, the overvoltage protection circuit 300 of the first configuration example has a configuration in which the feedback resistors 110 and 111 of the series regulator circuit 100 are shared as the resistors 307 and 308 for generating the reference voltage V30 from the output voltage Vout. It is depicted. 9B, the overvoltage protection circuit 300 of the second configuration example has a configuration in which the feedback resistors 110 and 111 of the series regulator circuit 100 are shared as the resistors 327 and 328 for generating the reference voltage V30 from the output voltage Vout. It is depicted.

  Note that the resistor 110 shown in FIGS. 2 and 3 corresponds to the resistors R1 and R2 shown in FIGS. 9A and 9B. The resistor 111 shown in FIGS. 2 and 3 corresponds to the resistor R3 shown in FIGS. 9A and 9B. The resistor 307 illustrated in FIG. 7 corresponds to the resistor R1 illustrated in FIG. 9A. The resistor 308 illustrated in FIG. 7 corresponds to the resistors R2 and R3 illustrated in FIG. 9A. The resistor 328 illustrated in FIG. 8 corresponds to the resistor R1 illustrated in FIG. 9B. The resistor 327 illustrated in FIG. 8 corresponds to the resistors R2 and R3 illustrated in FIG. 9B.

  In series regulator circuit 100, the bases of npn-type bipolar transistors 104 and 105 are connected to a connection node between resistors R2 and R3. Therefore, the base current IB1 of the transistors 104 and 105 is drawn from the connection node between the resistors R2 and R3. By extracting the base current IB1, there is a slight shift in the resistance voltage dividing ratio of the resistors R1 to R3. Further, when the resistance values of the resistors R1 to R3 are large, the temperature characteristics may be deteriorated by the extraction of the base current IB1.

  Under such circumstances, when the configuration in which the base of the pnp bipolar transistor 303 is connected to the connection node between the resistors R1 and R2 (see FIG. 9A), the connection node between the resistors R1 and R2 is used. Thus, the base current IB2 of the transistor 303 is supplied to the transistor 303, so that the above-described problems associated with the extraction of the base current IB1 can be alleviated.

  On the other hand, when a configuration in which the base of the npn-type bipolar transistor 303 is connected to the connection node between the resistors R1 and R2 (see FIG. 9B), the transistor 323 is further connected from the connection node between the resistors R1 and R2. Since the base current IB2 is extracted, the above-described problem is promoted.

  Therefore, when the overvoltage protection circuit 300 shares the feedback resistors 110 and 111 of the series regulator circuit 100, it can be said that it is desirable to adopt the first configuration example of FIG.

<Other variations>
Various technical features disclosed in the present specification can be variously modified within the scope of the technical creation in addition to the above-described embodiment. For example, the mutual replacement of the bipolar transistor and the MOS field effect transistor and the logic level inversion of various signals are arbitrary within the range in which the operations and functions described in the above embodiments can be realized. That is, the above-described embodiment is an example in all respects and should not be considered as limiting, and the technical scope of the present invention is not the description of the above-described embodiment, but the claims. It should be understood that all modifications that come within the meaning and range of equivalents of the claims are included.

  Various technical features disclosed in the present specification can be used, for example, as a technique for realizing a semiconductor device having no reference voltage source.

1 Series Regulator IC
100 series regulator circuit 101-103 pnp type bipolar transistor 104, 105 npn type bipolar transistor 106 P channel type MOS field effect transistor 107-111 resistance 112 current source 113, 114 N channel type MOS field effect transistor 115 P channel type MOS field effect Transistor 116 N-channel MOS field effect transistor 117 NOR circuit 200 Temperature protection circuit 201 to 205 pnp bipolar transistor 206 to 211 npn bipolar transistor 212 to 217 Resistor 218 N channel MOS field effect transistor 219 Inverter 300 Overvoltage protection Circuits 301 to 304 pnp type bipolar transistors 305 and 306 npn type bipolar transistors Star 307 to 312 Resistor 313 Current source 321 to 324 Npn bipolar transistor 325, 326 Pnp bipolar transistor 327 to 332 Resistor 333 Current source 334 Inverter

Claims (6)

A voltage dividing circuit for dividing the voltage to be monitored to generate the first voltage and the second voltage;
A comparison circuit that generates a voltage detection signal by comparing a differential voltage between the first voltage and the second voltage input to a transistor pair having different emitter areas with a threshold voltage ;
I have a,
The voltage dividing circuit includes:
A first voltage dividing circuit for generating a reference voltage between the monitored voltage and the ground voltage;
A second voltage dividing circuit for generating the first voltage and the second voltage between the monitoring target voltage and the reference voltage or between the reference voltage and the ground voltage;
Including
The first voltage dividing circuit includes a first resistor having a first end connected to an application end of the monitoring target voltage, and a second resistor connected between a second end of the first resistor and a ground end. And adjusting the threshold voltage according to a resistance voltage dividing ratio between the first resistor and the second resistor,
The second voltage dividing circuit cancels the temperature characteristics of the voltage detection signal between the input offset of the comparison circuit and the differential voltage corresponding to the input signal of the comparison circuit, thereby reducing the temperature dependence of the voltage detection signal. A voltage detection circuit characterized by being reduced . The comparison circuit includes the transistor pair as follows:
A first transistor having a base connected to the application terminal of the first voltage;
A second transistor having an emitter area larger than that of the first transistor, a base connected to the application terminal of the second voltage, and an emitter connected in common with the emitter of the first transistor;
The voltage detection circuit according to claim 1, comprising: The comparison circuit further includes:
A current source for generating emitter currents of the first transistor and the second transistor;
An output stage for generating the voltage detection signal from a collector voltage of the first transistor or the second transistor;
The regulator circuit according to claim 2, comprising: The second voltage dividing circuit includes:
A third transistor having a control electrode connected to the application terminal of the reference voltage and a first electrode connected to a ground terminal or the application terminal of the monitored voltage;
A resistance ladder connected between the second electrode of the third transistor and the application end or ground end of the monitoring target voltage to generate the first voltage and the second voltage;
The voltage detection circuit according to claim 1 , wherein the voltage detection circuit includes: The second voltage dividing circuit appropriately adjusts a resistance value of each resistor forming the resistor ladder, thereby mutually connecting between the input offset of the comparison circuit and the differential voltage corresponding to the input signal of the comparison circuit. The voltage detection circuit according to claim 4, wherein the temperature characteristics of the voltage are canceled . The voltage detection circuit according to any one of claims 1 to 5,
An internal circuit whose operation is controlled based on the voltage detection signal;
A semiconductor device comprising:

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