JP5434695B2 - Band gap circuit, low voltage detection circuit and regulator circuit - Google Patents

Description

  The present invention relates to a bandgap circuit that generates a temperature-independent bandgap voltage.

  In general, when a reference voltage that does not depend on temperature and power supply voltage is required, a reference voltage generation circuit called a band gap circuit is widely used. For example, the bandgap circuit adds a forward-biased pn junction voltage and a PTAT (Proportional To Absolute Temperature) voltage proportional to the absolute temperature. It is known that the forward-biased pn junction voltage has a negative linear dependence on absolute temperature (hereinafter also referred to as CTAT (Complementary To Absolute Temperature)) when the pn junction voltage is approximated by a linear expression. It has been. Therefore, a reference voltage independent of temperature, a so-called band gap voltage, is obtained by adding the forward biased pn junction voltage and an appropriate PTAT voltage. For this type of band gap circuit, various circuits have been devised and put into practical use.

US Patent Publication No. 7,193,454 US Patent No. 5,229,710 US Patent No. 5,325,045 US Pat. No. 6,362,612 US Patent No. 6,381,491 US Pat. No. 6,614,209 US Pat. No. 6,657,480 US Pat. No. 6,828,847 US Patent Publication No. 6,891,358 US Patent Publication No. 7,211,993 JP 2004-272346 A JP 2006-519433 A JP-T-2006-512682

R. S. Mao et al., “A New On-Chip Voltage Regulator for High Density CMOS DRAMs,” 1992 Symp. VLSI Circuits Dig. Of Tech. Papers, pp.108-109, 1992.

  However, the band gap circuit has a problem that the band gap voltage fluctuates due to the influence of the offset voltage of the operational amplifier. On the other hand, if the effect of the offset voltage is to be reduced as much as possible, there is a problem that the minimum operating voltage of the bandgap circuit increases.

  The disclosed apparatus includes a first reference voltage generation circuit that generates a first bandgap voltage, and a second reference voltage generation circuit that generates a second bandgap voltage.

  The first reference voltage generation circuit includes a first resistor having one end connected to the emitters of the first and second PNP transistors, the emitter of the second PNP transistor, the emitter of the first PNP transistor, and the first resistor. A first operational amplifier having an end connected to the input; In the first reference voltage generation circuit, the current densities of the first and second PNP transistors are made different from each other, and the potential of the emitter of the first PNP transistor and the first resistance are controlled by negative feedback control of the first operational amplifier. The first bandgap voltage is generated by making the potential at the other end equal.

  In addition to the first and second PNP transistors described above, the second reference voltage generation circuit includes a third PNP transistor whose base is connected to the emitter of the first PNP transistor, and a base connected to the emitter of the second PNP transistor. Has a fourth PNP transistor connected thereto. In the second reference voltage generation circuit, the second resistor having one end connected to the emitter of the fourth PNP transistor, the emitter of the third PNP transistor, and the other end of the second resistor are connected to the input. A second operational amplifier is included. In the second reference voltage generation circuit, the current densities of the third and fourth PNP transistors are made different from each other, and the potential of the emitter of the third PNP transistor and the second resistance are controlled by negative feedback control of the second operational amplifier. The second bandgap voltage is generated by making the potential at the other end equal.

  According to the disclosed apparatus, the first and second band gap voltages can be obtained simultaneously. Here, as compared with the second band gap voltage, the first band gap voltage can be generated even when the power supply voltage is low. On the other hand, the second band gap voltage is more accurate than the first band gap voltage because the influence of the offset voltage in the operational amplifier is small. According to the disclosed apparatus, the first and second PNP transistors are shared by the first reference voltage generation circuit and the second reference voltage generation circuit. Accordingly, the circuit element area can be reduced as compared with the case where the first and second output reference circuits are provided independently.

It is an example of the circuit diagram of a band gap circuit. It is a circuit diagram in consideration of the influence of the offset voltage in the band gap circuit. It is an example of the circuit diagram of a band gap circuit. It is a circuit diagram in consideration of the influence of the offset voltage in the band gap circuit. It is an example of the circuit diagram of a band gap circuit. It is an example of the circuit diagram of the band gap circuit which concerns on embodiment. An example of the circuit diagram of the transistor level of the band gap circuit which concerns on embodiment. An example of the circuit diagram of the transistor level of the band gap circuit which concerns on embodiment. The figure which shows the relationship between the output reference voltage and temperature in a band gap circuit. The figure which shows the relationship between the output reference voltage V and temperature in a band gap circuit. The figure which shows the relationship between the output reference voltage and power supply voltage in a band gap circuit. The enlarged view which shows the relationship between an output reference voltage and a power supply voltage. The enlarged view which shows the relationship between an output reference voltage and a power supply voltage. It is an example of the circuit diagram of the microcontroller to which the band gap circuit which concerns on embodiment is applied. It is an example of the circuit diagram of a low voltage detection circuit. It is an example of the circuit diagram of the transistor level about the comparator in a low voltage detection circuit. An example of the circuit diagram of the low voltage detection circuit which can switch an output reference voltage. It is an example of the circuit diagram of the band gap circuit which concerns on a modification. It is an example of the circuit diagram of the regulator circuit which can switch an output reference voltage.

  Hereinafter, an exemplary embodiment will be described with reference to the drawings.

[Band gap circuit]
First, for comparison, an example of a general bandgap circuit will be described with reference to FIGS.

  FIG. 1 is an example of a circuit diagram of a general band gap circuit. The band gap circuit of FIG. 1 includes resistors R1, R2, and R3, an operational amplifier AMP1, and a PNP transistor (precisely, a PNP bipolar transistor, hereinafter simply referred to as “BJT”) Q1 and Q2. GND indicates a GND terminal, VBGR indicates an output reference voltage, and IM and IP indicate internal nodes. The numerical value attached to the resistor indicates an example of the resistance value. The numerical values (× 1, × 10) attached to BJTQ1 and Q2 show examples of the relative area ratio of BJTQ1 and Q2. VBE1 indicates the base-emitter voltage of BJTQ1, and VBE2 indicates the base-emitter voltage of BJTQ2. In the following, the current flowing through the BJT indicates the emitter current unless otherwise specified.

  In the band gap circuit of FIG. 1, one end of a resistor R1 is connected to the emitter of BJTQ1, and one end of a resistor R3 is connected to the emitter of BJTQ2. The other end of the resistor R3 is connected to the input IM of the operational amplifier AMP1, and the emitter of BJTQ1 is connected to the input IP. The resistor R2 is provided in parallel with the resistor R1, and one end of the resistor R2 is connected to the other end of the resistor R3. The output of the operational amplifier AMP1, the other end of the resistor R1, and the other end of the resistor R2 are connected to a voltage line VL set to the output reference voltage VBGR. Therefore, a negative feedback circuit of the operational amplifier AMP1 is formed by the voltage line VL and the resistor R2.

  If the BJT base-emitter voltage (forward voltage of the pn junction) is VBE, VBE is expressed by the following equation (1).

VBE = Veg−a · T (1)
Here, Veg represents the band gap voltage of silicon, T represents the absolute temperature, and a represents the temperature coefficient of the voltage VBE. Here, it is known that Veg is about 1.2 V, and it is known that the value of a is about 2 mV / ° C. in a practical range. Therefore, for example, when T = 300K, the base-emitter voltage VBE is about 600 mV.

  The relationship between the emitter current IE of the BJT and the voltage VBE is expressed by the following equation (2).

IE = IO · exp {q · VBE / (k · T)} (2)
Here, IO represents a constant proportional to the emitter area of BJT, q represents the charge of electrons, and k represents the Boltzmann constant.

  When the voltage gain of the operational amplifier AMP1 is sufficiently large due to the negative feedback control of the operational amplifier AMP1, the potentials of the inputs IM and IP of the operational amplifier AMP1 are equal to each other and the circuit is stabilized. At this time, as shown in FIG. 1, when the ratio of the resistance values of the resistors R1 and R2 is 100k: 1M = 1: 10, the ratio of the magnitudes of the currents flowing through the BJTQ1 and Q2 is 10: 1. If the current flowing through BJTQ2 is I, the current flowing through BJTQ1 is I × 10. In FIG. 1, “I × 10” and “I” attached to BJTQ1 and Q2 indicate the relative relationship of currents flowing through BJTQ1 and Q2.

  In FIG. 1, it is assumed that the emitter area of BJTQ2 is 10 times the emitter area of BJTQ1. In FIG. 1, “× 1” and “× 10” attached to the BJTQ1 and Q2 indicate the relative relationship of the emitter areas.

  As described above, VBE1 represents the base-emitter voltage of BJTQ1, and VBE2 represents the base-emitter voltage of BJTQ2. Accordingly, the emitter currents of BJTQ1 and Q2 are expressed by the following equations (3) and (4) using equation (2).

10 × I = IO · exp {q · VBE1 / (k · T)} (3)
I = 10 × IO · exp {q · VBE2 / (k · T)} (4)
When equation (3) is divided by equation (4) on both sides, the following equation (5) is obtained.

100 = exp {q · VBE1 / (k · T) −q · VBE2 / (k · T)} (5)
Here, when VBE1−VBE2 = ΔVBE, Expression (5) is expressed by Expression (6) below.

ΔVBE = (k · T / q) ln (100) (6)
From equation (6), the base-emitter voltage difference ΔVBE of BJTQ1 and Q2 is expressed by the product of the logarithm of the current density ratio of BJTQ1 and Q2 ln (100) and the thermal voltage k · T / q. I understand that. As can be seen from FIG. 1, the potential of the input IP is VBE1. Since the potentials of the input IM and IP of the operational amplifier AMP1 are equal to each other, the potential of the input IM is VBE1. Accordingly, the voltage across the resistor R3 is VBE1-VBE2 = ΔVBE, and a current of ΔVBE / R3 flows through the resistors R2 and R3. Therefore, the voltage VR2 across the resistor R2 is expressed by the following equation (7).

VR2 = ΔVBE · R2 / R3 (7)
As described above, since the potentials of the inputs IM and IP of the operational amplifier AMP1 are equal to each other and the potential of the input IM is VBE1, the output reference voltage VBGR is expressed by the following equation (8).

VBGR = VBE1 + VR2
= VBE1 + ΔVBE · R2 / R3 (8)
As shown in the equation (1), the forward voltage VBE1 of the pn junction has a negative temperature dependency characteristic that decreases as the temperature increases. On the other hand, as shown in Equation (6), the difference ΔVBE between the base-emitter voltages of BJTQ1 and Q2 increases in proportion to the temperature. That is, in Expression (8), ΔVBE · R2 / R3 corresponds to the PTAT voltage. Accordingly, by appropriately selecting values such as R2 / R3 and current density, the output reference voltage VBGR becomes a value that does not depend on temperature. The output reference voltage VBGR at that time is about 1.2 V corresponding to the band gap voltage of silicon.

  The band gap circuit of FIG. 1 has an advantage that a reference voltage independent of temperature can be generated with a relatively simple circuit. However, in an actual integrated circuit, the potentials of the inputs IM and IP of the operational amplifier AMP1 do not completely match due to variations in element characteristics of the operational amplifier AMP1. In an ideal integrated circuit, when negative feedback control of the operational amplifier AMP1 is performed and the potentials of the input IM and IP of the operational amplifier AMP1 are equal to each other, the output voltage of the operational amplifier AMP1 is, for example, ½ of the power supply voltage. However, in an actual integrated circuit, since the element characteristics of the operational amplifier AMP1 vary, negative feedback control of the operational amplifier AMP1 is performed, and when the output voltage of the operational amplifier AMP1 is ½ of the power supply voltage, the input IM and IP The potentials are different from each other. The potential difference between the input IM and IP at this time is a so-called “offset voltage”. It is known that a typical offset voltage is about +10 mV to −10 mV. For this reason, the output reference voltage VBGR is affected by the offset voltage of the operational amplifier AMP1 included in the band gap circuit. Hereinafter, how the offset voltage affects the output reference voltage VBGR will be described.

  FIG. 2 is a circuit diagram in consideration of the influence of the offset voltage in the band gap circuit of FIG. 2, the same elements as those described in FIG. 1 are denoted by the same reference numerals, and detailed description thereof will be omitted. In the band gap circuit of FIG. 2, the operational amplifier AMP1 in the band gap circuit of FIG. 1 is replaced with a combination of an ideal operational amplifier IAMP1 and an offset voltage VOFF. The ideal operational amplifier IAMP1 is an ideal operational amplifier with an offset voltage of 0 mV. In the band gap circuit of FIG. 2, the potential of the input IIM of the ideal operational amplifier IAMP1 and the potential of IP coincide with each other, and the potential of the input IIM becomes VBE1. That is, the offset voltage VOFF of the operational amplifier AMP1 is added to the input IIM of the ideal operational amplifier IAMP1, and the potential of the input IM becomes a value obtained by adding the offset voltage VOFF to the potential of the node IIM. That is, the potentials of the inputs IM and IP are shifted by the offset voltage VOFF.

  In the band gap circuit of FIG. 1, when the offset voltage is not taken into consideration, the voltage VR3 across the resistor R3 is expressed by the following equation (9).

VR3 = ΔVBE (9)
On the other hand, in the band gap circuit of FIG. 2 in consideration of the offset voltage, the voltage VR3 ′ across the resistor R3 is expressed by the following equation (10).

VR3 ′ = ΔVBE + VOFF (10)
Therefore, when the offset voltage is taken into consideration, the voltage VR2 ′ across the resistor R2 is expressed by the following equation (11).

VR2 ′ = (ΔVBE + VOFF) · R2 / R3 (11)
Therefore, in the band gap circuit of FIG. 2, the output reference voltage VBGR is expressed by the following equation (8).

VBGR = VBE1 + VOFF + (ΔVBE + VOFF) · R2 / R3 (12)
As can be seen from equation (12), the actual PTAT voltage is (ΔVBE + VOFF) · R2 / R3. Here, since R2: R3 = 1M: 200K, R2 / R3 = 5, and the actual VBGR obtained by Expression (12) is offset from the ideal VBGR obtained by Expression (8). The value is obtained by adding 6 times the voltage VOFF.

  In the band gap circuit of FIGS. 1 and 2, in order to reduce the influence of the offset voltage of the operational amplifier as much as possible, the emitter area of BJTQ2 is set to 10 times the emitter area of BJTQ1 as described above. Further, the current flowing through BJTQ1 is set to 10 times the magnitude I of the current flowing through BJTQ2. Thus, the logarithm of the current density ratio of BJTQ1 and Q2 is expressed by ln (100), and ΔVBE is expressed by the above-described equation (6). Here, when the equation (6) is actually calculated with T = 300K, the following equation (13) is obtained.

ΔVBE = (k · T / q) ln (100)
≈ 26 mV x 4.6 = 120 mV (13)
Thus, ΔVBE is a value that is at least 10 times greater than the offset voltage VOFF (+10 mV to −10 mV). Thereby, the influence of the offset voltage VOFF can be suppressed to some extent. However, even in this case, as shown in the equation (12), in order to obtain the output reference voltage VBGR of 1200 mV by adding the PTAT voltage to the voltage VBE1 of about 600 mV, the value of the equation (13) is multiplied by five. Will be added to VBE1. Therefore, in the bandgap circuit of FIG. 2, the influence of the offset voltage VOFF on the output reference voltage VBGR is amplified about 6 times as shown in the equation (12). That is, as shown in FIG. 2, an error due to the amplified offset voltage VOFF of about +60 mV to −60 mV occurs with respect to the output reference voltage VBGR = 1200 mV.

  As can be seen from the above description, the bandgap circuit of FIG. 1 has the advantage that the bandgap circuit can be configured with a relatively simple circuit configuration, while the accuracy of the output reference voltage VBGR is improved by the offset voltage of the operational amplifier. Has the limit of being limited.

  In order to solve the accuracy limitation of the output reference voltage VBGR due to the offset voltage described above, a circuit has been proposed in which the output reference voltage is not 1,200 mV but 2400 mV or higher. An example of a band gap circuit that outputs an output reference voltage of 2400 mV will be described with reference to FIG.

  FIG. 3 is an example of a circuit diagram of a band gap circuit that outputs an output reference voltage of 2400 mV. In FIG. 3, the same elements as those described in FIG. 1 are denoted by the same reference numerals.

  The band gap circuit of FIG. 3 includes resistors R4, R5, R6, R7, R8, operational amplifiers AMP2, BJTQ3, Q4, Q5, Q6. VBGR24 indicates an output reference voltage of 2.4V, and NODE1 indicates an internal node. The numerical values (× 1, × 10) attached to BJTQ3, Q4, Q5, and Q6 show examples of the ratio of the relative emitter areas of BJTQ3, Q4, Q5, and Q6. VBE3 represents the base-emitter voltage of BJTQ3, and VBE5 represents the base-emitter voltage of BJTQ5.

  In the band gap circuit of FIG. 3, the base of BJTQ4 is connected to the emitter of BJTQ3, and the base of BJTQ6 is connected to the emitter of BJTQ5. One ends of resistors R4, R5, R6, and R8 are connected to the emitters of BJTQ3, Q5, Q4, and Q6, respectively. The other end of the resistor R8 is connected to the input IM of the operational amplifier AMP2, and the emitter of BJTQ4 is connected to the input IP. The resistor R7 is provided in parallel with the resistor R6, and one end of the resistor R7 is connected to the other end of the resistor R8. The other ends of the resistors R4, R5, R6, and R7 and the output of the operational amplifier AMP2 are connected to a voltage line VL24 set to the output reference voltage VBGR24. Accordingly, a negative feedback circuit of the operational amplifier AMP2 is formed by the voltage line VL24 and the resistor R7.

  In the band gap circuit of FIG. 1, the output reference voltage VBGR is represented by the sum of the voltage VBE1 and the PTAT voltage which is a voltage proportional to the absolute temperature T, as shown in the equation (8). On the other hand, in the bandgap circuit of FIG. 3, the output reference voltage VBGR24 is different in that the output reference voltage VBGR24 is indicated by the sum of the value obtained by doubling the base-emitter voltage VBE and the PTAT voltage. This will be specifically described below.

  As described above, the base-emitter voltage VBE of the BJT is expressed by the equation (1), and the relationship between the emitter current IE of the BJT and the voltage VBE is expressed by the equation (2). When the voltage gain of the operational amplifier AMP2 is sufficiently large due to the negative feedback control of the operational amplifier AMP2, the potentials of the inputs IM and IP of the operational amplifier AMP2 are equal to each other and the circuit is stabilized. At this time, as described in FIG. 1, when the ratio of the resistance values of the resistors R6 and R7 is 1:10, the ratio of the magnitudes of the currents flowing through the BJTQ4 and Q6 is 10: 1. Therefore, if the current flowing through BJTQ6 is I, the current flowing through BJTQ4 is I × 10. In FIG. 3, “I × 10” and “I” attached to BJTQ4 and Q6 indicate the relative relationship of currents flowing through BJTQ4 and Q6. In addition, by appropriately designing the relationship between the resistors R4 and R5, the current flowing through the BJTQ3 and Q5 is also designed to be 10: 1. “I × 10” and “I” attached to the BJTQ3 and Q5 indicate a relative relationship between currents flowing through the BJTQ3 and Q5. Further, in FIG. 3, it is assumed that the emitter area of BJTQ6 is 10 times the emitter area of BJTQ4, and the emitter area of BJTQ5 is 10 times the emitter area of BJTQ3. In FIG. 3, “× 1” and “× 10” attached to BJTQ3, Q4, Q5, and Q6 indicate the relative relationship of the emitter areas.

  The emitter currents of BJTQ4 and Q6 are expressed by the following equations (14) and (15) using equation (2), where the base-emitter voltages of BJTQ4 and Q6 are VBE4 and VBE6.

10 × I = IO · exp {q · VBE4 / (k · T)} (14)
I = 10 × IO · exp {q · VBE6 / (k · T)} (15)
When equation (14) is divided by equation (15) on both sides, the following equation (16) is obtained.

100 = exp {q · VBE4 / (k · T) −q · VBE6 / (k · T)} (16)
Here, when VBE4−VBE6 = ΔVBE46, Expression (16) is expressed by Expression (17) below.

ΔVBE46 = (k · T / q) ln (100) (17)
The emitter currents of BJTQ3 and Q5 are expressed by the following equations (18) and (19) using equation (2).

10 × I = IO · exp {q · VBE3 / (k · T)} (18)
I = 10 × IO · exp {q · VBE5 / (k · T)} (19)
When Expression (18) is divided by Expression (19), it is expressed by the following Expressions (20) and (21).

100 = exp {q · VBE3 / (k · T) −q · VBE5 / (k · T)} (20)
ΔVBE35 = (k · T / q) ln (100) (21)
Since the emitter of BJTQ3 is connected to the base of BJTQ4, the potential of the input IP is the sum of VBE3 and VBE4. Further, since the emitter of BJTQ5 is connected to the base of BJTQ6, the potential of NODE1 is the sum of VBE5 and VBE6.

  Since the potentials of the inputs IM and IP are equal, the voltage VR8 across the resistor R8 corresponds to the potential difference between IP and NODE1, and is expressed by the following equation (22).

VR8 = VBE3 + VBE4- (VBE5 + VBE6) (22)
= ΔVBE46 + ΔVBE35
That is, it can be seen that the voltage VR8 is equal to the sum of ΔVBE46 and ΔVBE35, and is proportional to the absolute temperature from the equations (17) and (21). In other words, the voltage VR8 is ln (100) which is the logarithm of the current density ratio of BJTQ4 and Q6 with respect to the product of ln (100) which is the logarithm of the current density ratio of BJTQ3 and Q5 and k · T / q which is the thermal voltage. And the product of the thermal voltage k · T / q.

  A current IR8 flowing through the resistor R8 is expressed by the following equation (23).

IR8 = (ΔVBE35 + ΔVBE46) / R8 (23)
Therefore, the voltage VR7 of the resistor R7 is expressed by the following equation (24).

VR7 = (ΔVBE35 + ΔVBE46) · R7 / R8 (24)
Since the potentials of the inputs IP and IM are both VBE3 + VBE4, the output reference voltage VBGR24 is expressed by the following equation (25).

VBGR24 = VBE3 + VBE4
+ (ΔVBE35 + ΔVBE46) · R7 / R8 (25)
The forward voltages VBE3 and VBE4 of the pn junction have a negative temperature dependence characteristic that decreases as the temperature rises (VBE = Veg−a · T (1)). On the other hand, ΔVBE35 and ΔVBE46 increase in proportion to the temperature. That is, (ΔVBE35 + ΔVBE46) · R7 / R8 corresponds to the PTAT voltage. Therefore, by appropriately selecting values such as R7 / R8 and current density, the output reference voltage VBGR24 becomes a value that does not depend on temperature. The output reference voltage VBGR 24 at that time is about 2.4 V corresponding to twice the band gap voltage of silicon because the value of two VBEs is included in the equation (25).

  The advantage of the bandgap circuit of FIG. 3 is that the influence of the offset voltage of the operational amplifier AMP2 is halved compared to the bandgap circuit of FIG. Hereinafter, the influence of the offset voltage of the operational amplifier AMP2 will be described with reference to FIG.

  FIG. 4 is a circuit diagram in consideration of the influence of the offset voltage in the band gap circuit of FIG. In FIG. 4, the same elements as those described in FIG. 3 are denoted by the same reference numerals, and detailed description thereof will be omitted. In the band gap circuit of FIG. 4, the operational amplifier AMP2 in the band gap circuit of FIG. 3 is replaced with a combination of the ideal operational amplifier IAMP2 and the offset voltage VOFF. The ideal operational amplifier IAMP2 is an ideal operational amplifier with an offset voltage of 0 mV. In the band gap circuit of FIG. 4, the potential of the input IIM of the ideal operational amplifier IAMP2 and the potential of IP coincide with each other, and the potential of the input IIM becomes VBE3 + VBE4. The offset voltage VOFF of the operational amplifier AMP2 is added to the input IIM of the ideal operational amplifier IAMP2. That is, the potentials of the inputs IM and IP are shifted by the offset voltage VOFF.

  In the band gap circuit of FIG. 3, when the offset voltage is not taken into consideration, the voltage VR8 across the resistor R8 is expressed by the following equation (26).

VR8 = ΔVBE35 + ΔVBE46 (26)
On the other hand, in the band gap circuit of FIG. 4 considering the offset voltage, the voltage VR8 ′ across the resistor R8 is expressed by the following equation (10).

VR8 ′ = ΔVBE35 + ΔVBE46 + VOFF (27)
Accordingly, when the offset voltage is taken into consideration, the voltage VR7 ′ across the resistor R7 is expressed by the following equation (28).

VR7 ′ = (ΔVBE35 + ΔVBE46 + VOFF) · R2 / R3 (28)
Therefore, in the band gap circuit of FIG. 5, the output reference voltage VBGR24 is expressed by the following equation (29).

VBGR24 = VBE3 + VBE4 + VOFF
+ (ΔVBE35 + ΔVBE46 + VOFF) ・ R7 / R8 ・ ・ (29)
Here, in the same manner as described in FIG. 4, when R7 / R8 = 5, the actual VBGR 24 obtained by the equation (29) is smaller than the ideal VBGR 24 obtained by the equation (25). Is a value obtained by multiplying by six.

  In the band gap circuit of FIGS. 3 and 4, in order to reduce the influence of the offset voltage of the operational amplifier as much as possible, the emitter area of BJTQ5 is made 10 times the emitter area of BJTQ3, and the emitter area of BJTQ6 is 10 times the emitter area of BJTQ4. It is said. Further, the current flowing through BJTQ3 is 10 times the current flowing through BJTQ5, and the current flowing through BJTQ4 is 10 times the current flowing through BJTQ6. Accordingly, the logarithm of the current density ratio of BJTQ3 and Q5 and the logarithm of the current density ratio of BJTQ4 and Q6 are expressed by ln (100). Here, when T = 300K and ΔVBE35 + ΔVBE46 is actually calculated using the equations (17) and (21), the following equation (30) is obtained.

ΔVBE35 + ΔVBE46 = (k · T / q) ln (100)
+ (K · T / q) ln (100)
≒ 26mV × 4.6 × 2 = 240mV (30)
As described above, ΔVBE35 + ΔVBE46 is twice the value of ΔVBE (120 mV) of the band gap circuit of FIG.

  As described above, in order to obtain the output reference voltage VBGR of 1200 mV in the bandgap circuit of FIG. 1, the value of equation (13) (ΔVBE = 120 mV) is multiplied by 5 and added to VBE1 of 600 mV. It was. At this time, the influence of the offset voltage VOFF on the output reference voltage VBGR was amplified about 6 times (see Expression (12)). On the other hand, in order to obtain the output reference voltage VBGR24 of 2400 mV in the bandgap circuit of FIG. 3, the value of equation (30) (ΔVBE35 + ΔVBE46 = 240 mV) is multiplied by 5 and added to VBE3 + VBE4 of 1200 mV. Also in this case, the influence of the offset voltage VOFF on the output reference voltage VBGR24 is amplified by about 6 times (see Expression (29)).

  However, when both offset voltages VOFF are 10 mV, the value of VBGR in the band gap circuit of FIG. 1 is 1200 mV ± 60 mV, whereas the value of VBGR 24 in the band gap circuit of FIG. 3 is 2400 mV ± 60 mV. Become. That is, the absolute value of the error due to the influence of the offset voltage is the same, but VBGR24 has an output reference voltage that is twice that of VBGR, so that VBGR has an error of 5%, whereas VBGR24 has an error of 2.5%. It becomes an error. As described above, the band gap circuit of FIG. 3 has an advantage that the influence of the offset voltage VOFF can be halved as compared with the band gap circuit of FIG.

  Here, in the band gap circuit of FIG. 3, since the output reference voltage is 2400 mV, the minimum operating voltage is larger than the band gap circuit of FIG. 1 in which the output reference voltage is 1200 mV. Therefore, a bandgap circuit in which the output reference voltage is 1200 mV while maintaining the advantage of reducing the influence of the offset voltage will be described with reference to FIG.

  FIG. 5 shows an example of a bandgap circuit in which the output reference voltage is 1200 mV while maintaining the advantage of reducing the influence of the offset voltage. In FIG. 5, the same elements as those described in FIG. 3 are denoted by the same reference numerals.

  5 includes resistors R8, R9, operational amplifiers AMP3, BJTQ3, Q4, Q5, Q6, Q7, and PMOS transistors PM1, PM2, PM3, PM4, and PM5. In the bandgap circuit of FIG. 5, VDP5 indicates a + 5V power source, for example. Since the PMOS transistors PM1, PM2, PM4, and PM5 have the same gate potential, they function as current mirrors.

  In the band gap circuit of FIG. 3, the elements that supply current to BJTQ3, Q5, Q4, and Q6 are resistors R4, R5, R6, and R7. These resistors R4, R5, R6, and R7 respectively supplied currents corresponding to potential differences between the voltage line VL24 set to the output reference voltage VBGR24 and the emitter of BJTQ3, the emitter of BJTQ5, IP, and NODE1. .

  On the other hand, in the band gap circuit of FIG. 5, the PMOS transistors PM1, PM2, PM3, and PM4 serving as current sources supply a 10: 1 current to the BJTQ3 and Q5 and a 10: 1 current to the BJTQ4 and Q6. To do. The negative feedback control of the operational amplifier AMP3 is performed so that the potentials of the inputs IM and IP are the same as in the band gap circuit of FIG. Therefore, the voltage across the resistor R8 is the same as that in the band gap circuit of FIG. 3, and the current IR8 flowing through the resistor R8 is expressed by equation (23).

IR8 = (ΔVBE35 + ΔVBE46) / R8 (23)
As described above, the PMOS transistors PM1, PM2, PM4, and PM5 function as current mirrors. Therefore, since the current flowing through the resistor R8 and the current flowing through the PMOS transistor PM5 are proportional, the current flowing through the resistors R9 and BJTQ7 is also a current proportional to the absolute temperature (hereinafter, simply “ Referred to as “PTAT current”).

  When a PTAT current is passed through BJTQ7 and resistor R9, the total voltage becomes approximately 1200 mV, which is a voltage that does not depend on temperature, as in the band gap circuit of FIG. That is, the output reference voltage VBGR can be generated by appropriately setting the value of the resistor R9 and the ratio of the currents flowing through the PMOS transistors PM4 and PM5.

  In the bandgap circuit of FIG. 5, like the bandgap circuit of FIG. 3, the voltage across the resistor R8 can be increased to 240 mV, for example, and the influence of the offset voltage on the output reference voltage can be reduced. In the band gap circuit of FIG. 5, the output reference voltage is set to 1200 mV, so that it can operate with a lower power supply voltage than the band gap circuit of FIG.

  However, in the bandgap circuit of FIG. 5, the elements that supply current to the BJTQ3, Q4, Q5, and Q6 are current mirrors including PMOS transistors. Therefore, the characteristic error of these PMOS transistors causes the error of the output reference voltage. There is a risk of becoming. In other words, when comparing the resistance and the PMOS transistor, the PMOS transistor has more parameters to be controlled and is often more disadvantageous than the resistance in terms of matching, which is the degree of matching of the element characteristics. In other words, although the bandgap circuit of FIG. 5 has succeeded in improving the minimum operating voltage, the error of the output reference voltage is larger than that of the bandgap circuit of FIG. 3 due to the characteristic error of the PMOS transistor. There is a possibility.

  Therefore, in the bandgap circuit according to the embodiment, by combining the bandgap circuit of FIG. 1 and the bandgap of FIG. 3, the output reference voltage VBGR24 in which the accuracy of the output reference voltage is improved and the minimum operating voltage is suppressed. The obtained output reference voltage VBGR is obtained. Hereinafter, this will be specifically described with reference to FIG.

  FIG. 6 is an example of a circuit diagram of the bandgap circuit according to the embodiment. In FIG. 6, the same elements as those described in FIG. Specifically, in FIG. 6, Qn (n is an integer) indicates BJT, Rn (n is an integer) indicates a resistance and a resistance value, GND indicates a GND terminal, and AMPn (n is an integer) indicates an operational amplifier. Show. Further, IM, IP, IM2, IP2, NODE1 indicate internal nodes, VBGR indicates an output reference voltage of 1.2V, and VBGR24 indicates an output reference voltage of 2.4V. VBEn (n is an integer) indicates the base-emitter voltage of BJTQn. The numerical values (× 1, × 10) attached to BJTQn show examples of the ratio of the relative emitter area of BJTQn.

  The band gap circuit of FIG. 6 includes a circuit CC1 and a circuit CC2. The circuit CC1 is a circuit part that outputs the output reference voltage VBGR, and the circuit CC2 is a circuit part that outputs the output reference voltage VBGR24. The output reference voltage VBGR corresponds to the first band gap voltage, and the output reference voltage VBGR 24 corresponds to the second band gap voltage. Therefore, the circuit CC1 functions as a first reference voltage generation circuit, and the circuit CC2 functions as a second reference voltage generation circuit.

  The circuit CC1 includes BJTQ3 and Q5, resistors R10, R11 and R12, and an operational amplifier AMP4. One end of a resistor R10 is connected to the emitter of BJTQ3, and one end of a resistor R12 is connected to the emitter of BJTQ5. The other end of the resistor R12 is connected to the input IM2 of the operational amplifier AMP4, and the emitter of BJTQ3 is connected to the input IP2. The resistor R11 is provided in parallel with the resistor R10, and one end of the resistor R11 is connected to the other end of the resistor R12. That is, one end of the resistors R10 and R11 is also connected to the input of the operational amplifier AMP4. The output of the operational amplifier AMP4 and the other ends of the resistors R10 and R11 are connected to a voltage line VL set to the output reference voltage VBGR. Therefore, a negative feedback circuit of the operational amplifier AMP4 is formed by the voltage line VL and the resistor R11. BJTQ3 functions as a first PNP transistor, and BJTQ5 functions as a second PNP transistor. The resistor R12 functions as a first resistor, and the operational amplifier AMP4 functions as a first operational amplifier. Further, the resistor R10 functions as a third resistor, and the resistor R11 functions as a fourth resistor.

  The circuit CC2 includes BJTQ3, Q5, Q4, Q6, resistors R6, R7, R8, and an operational amplifier AMP2. The base of BJTQ4 is connected to the emitter of BJTQ3, and the base of BJTQ6 is connected to the emitter of BJTQ5. One end of a resistor R6 is connected to the emitter of BJTQ4, and one end of a resistor R8 is connected to the emitter of BJTQ6. The other end of the resistor R8 is connected to the input IM of the operational amplifier AMP2, and the emitter of BJTQ6 is connected to the input IP. The resistor R7 is provided in parallel with the resistor R8, and one end of the resistor R7 is connected to the other end of the resistor R8. The output of the operational amplifier AMP2 and the other ends of the resistors R6 and R7 are connected to a voltage line VL24 set to the output reference voltage VBGR24. Accordingly, a negative feedback circuit of the operational amplifier AMP2 is formed by the voltage line VL24 and the resistor R7. BJTQ4 functions as a third PNP transistor, and BJTQ6 functions as a fourth PNP transistor. Further, the resistor R8 functions as a second resistor, and the operational amplifier AMP2 functions as a second operational amplifier.

  Hereinafter, the operation of the band gap circuit of FIG. 6 will be described separately for the circuits CC1 and CC2.

  First, the operation of the circuit CC1 will be described. As described above, the circuit CC1 including the BJTQ3 and Q5, the resistors R10, R11, and R12, and the operational amplifier AMP4 is a bandgap circuit that outputs the output reference voltage VGBR of 1.2 V, similarly to the bandgap circuit of FIG. Function as. The difference between the bandgap circuit of FIG. 1 and the circuit CC1 is that in the circuit CC1, the emitter of BJTQ3 is connected to the base of BJTQ4, and the emitter of BJTQ5 is connected to the base of BJTQ6.

  When the voltage gain of the operational amplifier AMP4 is sufficiently large due to the negative feedback control of the operational amplifier AMP4, the potentials of the inputs IM2 and IP2 of the operational amplifier AMP4 are equal to each other and the circuit is stabilized. That is, the potential of the input IM2 is VBE3, and the voltage across the resistor R12 is ΔVBE35 that is the difference between VBE3 and VBE5. Therefore, in the case of R7 / R8 = 5, in order to obtain the output reference voltage VBGR of 1200 mV by adding the PTAT voltage to VBE3 of about 600 mV, the value of equation (13) is multiplied by 5 and added to VBE1. become. That is, as described in FIG. 1, when ΔVBE35 is set to 120 mV, the output reference voltage VBGR becomes 1200 mV.

  That is, as described in FIG. 1, by setting the current flowing through BJTQ3 to 10 times the current flowing through BJTQ5 and setting the emitter area of BJTQ5 to 10 times the emitter area of BJTQ3, ΔVBE35 can be expressed by the following equation (31). It is represented by Here, if T = 300K, ΔVBE = about 120 mV.

ΔVBE35 = (kT / q) ln (100)
= 26 mV × 4.6 = 120 mV (31)
Here, unlike the band gap circuit of FIG. 1, the base currents of BJTQ4 and Q6 flow in BJTQ3 and Q5 in addition to the current flowing from resistors R10 and R11. Therefore, in order to make the current flowing through BJTQ3 10 times the current flowing through BJTQ5, the ratio of the resistance values of resistors R10 and R11 is 1:10, and the ratio of the base currents of BJTQ4 and Q6 is 10: 1. . Specifically, since the current amplification factors hFE of BJTQ4 and Q6 are the same, the ratio of the emitter currents of BJTQ4 and Q6 is set to 10: 1. As a result, the base current ratio of BJTQ4 and Q6 is also 10: 1. By doing so, an output reference voltage VBGR of 1200 mV can be generated as in FIG.

  Next, the operation of the circuit CC2 will be described. As described above, the circuit CC2 having the BJTQ3, Q5, Q4, Q6, the resistors R6, R7, R8, and the operational amplifier AMP2 outputs the 2.4 V output reference voltage VGBR24 as in the band gap circuit of FIG. Functions as a band gap circuit.

  Since the base of BJTQ4 is connected to the emitter of BJTQ3, the potential of IP becomes VBE3 + VBE4. On the other hand, since the base of BJTQ6 is connected to the emitter of BJTQ5, the potential of NODE1 is VBE5 + VBE6. When the voltage gain of the operational amplifier AMP2 is sufficiently large due to the negative feedback control of the operational amplifier AMP2, the potentials of the inputs IM and IP of the operational amplifier AMP2 are equal to each other and the circuit is stabilized. Further, the current flowing through BJTQ4 is set to 10 times the current flowing through BJTQ6, and the emitter area of BJTQ6 is set to 10 times the emitter area of BJTQ4. At this time, the voltage across the resistor R8 is expressed by the following equation (32), assuming that T = 300K, as described in FIG.

ΔVBE35 + ΔVBE46 = (k · T / q) ln (100)
+ (K · T / q) ln (100)
≈ 26 mV x 4.6 x 2 = 240 mV (32)
In addition, by setting the resistance value ratio of the resistors R6 and R7 to 1:10, the emitter current ratio of the BJTQ4 and Q6 can be set to 10: 1. Here, the ratio of the emitter currents of BJTQ4 and Q6 coincides with the ratio of current flowing from resistors R10 and R11. In this way, even if the base current of BJTQ4 and Q6 is not negligible relative to the current flowing from resistors R10 and R11, the ratio of the emitter currents of BJTQ3 and Q5 is as described above. Can be set to a predetermined value (10: 1 in the example of FIG. 6).

  If R7 / R8 = 5, the output reference voltage VBGR24 is obtained by adding the PTAT voltage (ΔVBE35 + ΔVBE46) · R7 / R8 to the IP voltage VBE3 + VBE4, as described in FIG. Therefore, as described in FIG. 3, the circuit CC2 functions as a band gap circuit that outputs the output reference voltage VBGR24 of 2.4V.

  As can be seen from the above description, in the bandgap circuit according to the embodiment shown in FIG. 6, the output reference voltage VBGR of the bandgap circuit of FIG. 1 and the output reference voltage VBGR2 of the bandgap circuit of FIG. It is possible to obtain. Further, in the band gap circuit of FIG. 6, the same effect as described in FIG. 1 can be obtained for the circuit CC1, and the same effect as described in FIG. 3 can be obtained for the circuit CC2.

  In the band gap circuit of FIG. 6, BJTQ3 and Q5 are shared by the circuit CC1 which is the VBGR output circuit portion and the circuit CC2 which is the VBGR24 output circuit portion. Therefore, compared with the case where two band gap circuits of FIG. 1 and two band gap circuits of FIG. 3 are provided independently, the circuit element area can be reduced in the band gap circuit of FIG.

  Compared with the band gap circuit of FIG. 3 alone, in the band gap circuit of FIG. 6, the circuit element area is hardly increased or the circuit element area can be reduced in some cases. More specifically, the resistance value of the band gap circuit of FIG. 6 can be reduced compared with the band gap circuit of FIG. 3, so that the circuit element area does not need to be increased. Hereinafter, this reason will be described.

  Compare the values of the resistors R4 and R5 in the band gap circuit of FIG. 3 with the values of the resistors R10 and R11 in the band gap circuit of FIG. Here, in the description of FIG. 3, the ratio of the currents flowing through the BJTQ3 and Q5 is 10: 1. In this case, VBE3 is 60 mV greater than VBE5 at T = 300K, and the voltage across resistor R5 is 60 mV greater than the voltage across resistor R4. However, since the potential difference between VBE3 and VBE5 is small, it is assumed here that the voltages at both ends of the resistors R4 and R5 are substantially equal for simplicity.

  In the band gap circuit of FIG. 3, the other ends of the resistors R4 and R5 are connected to a voltage line VL24 set to the output reference voltage VBGR24. Therefore, when the output reference voltage VBGR24 is 2.4V and VBE3 is 0.6V, the voltage across the resistors R4 and R5 is 1.8V. If the current flowing through BJTQ3 is 6 μA and the current flowing through BJTQ5 is 0.6 μA, the resistance value of resistor R4 is 1.8 V / 6 μA = 300 kΩ, and the resistance value of resistor R5 is 3000 kΩ, which is 10 times this value. It becomes.

  On the other hand, in the bandgap circuit of FIG. 6, the other ends of the resistors R10 and R11 are connected to a voltage line VL set to the output reference voltage VBGR. Therefore, when the output reference voltage VBGR is 1.2V and VBE3 is 0.6V, the voltage across the resistors R10 and R11 is 0.6V. That is, compared to the voltage across the resistors R4 and R5 in the bandgap circuit of FIG. 3, the voltage across the resistors R10 and R11 is 1/3. Here, if the current flowing through the BJTQ10 is 6 μA and the current flowing through the BJTQ11 is 0.6 μA, the resistance value of the resistor R10 is 100 kΩ, and the resistance value of the resistance value R11 is 1000 kΩ. That is, compared to the resistors R4 and R5 in the band gap circuit of FIG. 3, the resistance values of the resistors R10 and R11 in the band gap circuit of FIG. Can be.

  In summary, the voltages applied to the resistors R4 and R5 in the band gap circuit of FIG. 3 are VBGR24-VBE3 and VBGR24-VBE5. In contrast, the voltages applied to the resistors R10 and R11 in the band gap circuit of FIG. 6 are as small as VBGR-VBE3 and VBGR-VBE5. Therefore, compared to the resistance values of the resistors R4 and R5 in the bandgap circuit of FIG. 3, the resistance values of the resistors R10 and R11 in the bandgap circuit of FIG. can do.

  In a band gap circuit mounted on a microcontroller, it is often desired that the power consumption is small, so that the resistance value tends to increase. Therefore, in such a band gap circuit, the ratio of the resistance to the entire circuit area increases. In this regard, in the band gap circuit of FIG. 6, although the number of operational amplifiers and the like is increased as compared with the band gap circuit of FIG. 3, the resistance value is small as described above. Therefore, compared with the circuit element area of the band gap circuit of FIG. 3, the circuit element area of the band gap circuit of FIG. Rather, in the band gap circuit of FIG. 6, when the resistance per unit square of the resistance element, that is, the sheet resistance is small, the effect of reducing the circuit element area can be obtained.

  Compared with the band gap circuit of FIG. 3, the band gap circuit of FIG. 6 can suppress the fluctuation of the output reference voltage VBGR24 with respect to the fluctuation of the current amplification factor hFE of BJT. Hereinafter, this reason will be described.

  As described above, in the band gap circuit of FIG. 3, when the ratio of the currents flowing through BJTQ3 and Q5 is 10: 1, VBE3 is 60 mV larger than VBE5 at 300 K, and the voltage across resistor R5 Is 60 mV greater than the voltage across resistor R4. Since the difference ΔVBE35 between VBE3 and VBE5 is proportional to the absolute temperature, when the temperature changes, the voltage across the resistors R4 and R5 changes to such an extent that ΔVBE35 changes. That is, since the ratio of the resistance values of the resistors R4 and R5 is constant at 1:10, the voltage at both ends of the resistor R5 becomes larger than the voltage at both ends of the resistor R4 as the temperature increases. Therefore, even if the ratio of the values of the currents flowing through the BJTQ3 and Q5 is designed to be 10: 1 at room temperature, it changes from 10: 1 when the temperature rises. Since ΔVBE35 is determined so as to balance with the ratio of the current values, the actual ΔVBE35 is not proportional to the temperature and is smaller than the value of ΔVBE35 in the equation (30).

  ΔVBE35 is also affected by the base currents of BJTQ4 and Q6. That is, the value of the emitter current of BJTQ3 and Q5 is indicated by the sum of the value of the current flowing through resistors R4 and R5 and the value of the base current of BJTQ4 and Q6. In the band gap circuit of FIG. 3, as described above, it is difficult to accurately design the ratio of the values of the currents flowing through the resistors R4 and R5 to 10: 1. Therefore, it is also affected by fluctuations in the BJT current amplification factor hFE. Here, the relationship between the base currents of the BJTQ4 and Q5 and the current amplification factor hFE is represented by the current flowing through the resistors R6 and R7 = the base current of the BJTQ4 and Q5 × (1 + hFE). Therefore, if the ratio of the resistance values of the resistors R6 and R7 is designed to be 1:10, the base current value of the BJTQ4 and Q5 can be designed to be 10: 1. However, in the bandgap circuit of FIG. 3, as described above, even when the base current values of the BJTQ4 and Q5 are designed to be 10: 1, the ratio of the current values flowing through the resistors R4 and R5 is accurately 10: 1. Difficult to design. Therefore, the ratio of the emitter current values of BJTQ3 and Q5 varies under the influence of the current amplification factor hFE of BJTQ4 and Q5. As a result, ΔVBE35 varies and the output reference voltage VBGR24 also varies.

  On the other hand, in the band gap circuit of FIG. 6, one end of the resistors R10 and R11 is connected to the input of the operational amplifier AMP4, and the other end of the resistors R10 and R11 is connected to the voltage line VL set to the output reference voltage VBGR. Has been. Due to the negative feedback control of the operational amplifier AMP4, the potential of the input IM2 becomes VBE3. Therefore, both of the voltages across the resistors R10 and R11 can be set to the output reference voltage VBGR-VBE3. Therefore, by setting the ratio of the resistance values of the resistors R10 and R11 to 1:10, the ratio of the values of the currents flowing through the resistors R10 and R11 can be set to 10: 1 regardless of the temperature. Therefore, if the ratio of the emitter current values of BJTQ4 and Q6 is set to 10: 1, the ratio of the emitter currents of BJTQ3 and Q5 will change from 10: 1 even if the current amplification factor hFE of BJT varies. There is no. That is, in the band gap circuit of FIG. 6, the ratio of the values of the currents flowing through the resistors R10 and R11 can be 10: 1 regardless of the temperature, so that the fluctuation of the output reference voltage VBGR24 with respect to the fluctuation of the current amplification factor hFE It has the advantage that can be suppressed.

  Further, compared with the band gap circuit of FIG. 5, the band gap circuit of FIG. 6 has an advantage that the accuracy of the output reference voltage can be improved. This will be specifically described below.

  In the band gap circuit of FIG. 5, in order to solve the problem that it is difficult to accurately design the ratio of the values of the currents flowing through the resistors R4 and R5 in the band gap circuit of FIG. 3, current is supplied to the BJTQ3 and Q5. The element was a PMOS transistor current mirror. Further, unlike the bandgap circuit of FIG. 3, the bandgap circuit of FIG. 5 eliminates the voltage drop due to the resistance by supplying current to the BJTQ4 and Q6 by the PMOS transistors PM3 and PM4. Was possible. However, as described above, in the bandgap circuit of FIG. 5, the ratio of the values of the currents flowing through the PMOS transistors depends on the degree of matching of the characteristics of the PMOS transistors. There are increasing sources of error. Since the degree of matching of the resistance characteristics is usually better than the degree of matching of the PMOS transistors, the bandgap circuit of FIG. 6 is more current in the PMOS transistor current mirror than the bandgap circuit of FIG. This is advantageous in terms of accuracy because no error occurs due to. That is, compared with the bandgap circuit of FIG. 5 that requires matching of the characteristics of the PMOS transistors, the bandgap circuit of FIG. 6 can design the output reference voltage VBGR24 only by the resistance ratio. Accuracy can be improved.

  In the description of the bandgap circuit of FIG. 6, the ratio of the current values of BJTQ3 and Q5 and the ratio of the current values of BJTQ4 and Q6 are described as 10: 1 as an example, but the present invention is not limited to this. And can be designed freely. In addition, the ratio of the emitter areas of BJTQ3 and Q5 and the ratio of the emitter areas of BJTQ4 and Q6 have been described as 1:10 as an example, but the present invention is not limited to this, and the design can be made with any ratio.

  Next, a transistor level circuit example of the band gap circuit of FIG. 6 will be described with reference to FIGS. Circuit simulation results of the bandgap circuit of FIG. 6 having the circuits shown in FIGS. 7 and 8 will be described with reference to FIGS.

  7 and 8 show an example of a transistor level circuit diagram of the bandgap circuit of FIG. 7 and 8, in the band gap circuit of FIG. 6, the operational amplifier is described at the transistor level, and an example of the bias circuit is described. In FIGS. 7 and 8, some circuits such as a start-up circuit are omitted to simplify the drawing.

  7 and 8, Qn (n is an integer) indicates a PNP transistor (BJT), Rn and RBn (n is an integer) indicates a resistance and its resistance value, and PMBn (n is an integer) indicates a PMOS transistor. , NMBn (n is an integer) indicates an NMOS transistor. GND indicates a GND terminal, and VDPn (n is an integer) indicates a positive power supply voltage of nV, for example. In the example shown in FIGS. 7 and 8, since it is VDP5, the power supply voltage is + 5V. BPB and BNB indicate bias potentials, CB1 indicates a phase compensation capacitance, VOFF and VOFF24 indicate offset voltages, and VBE3, VBE5, IM2, IP2, IP, and IM indicate internal nodes. The circuit elements and nodes corresponding to FIG. 6 are given the same element names and node names. 7 and 8, the offset voltages are shown as voltage sources VOFF and VOFF24, but this is for confirming the influence of the offset voltage on the circuit simulation. In practice, this voltage source VOFF Needless to say, VOFF24 does not exist.

  The operation of the bias circuit and the operation of each operational amplifier circuit will be briefly described below.

  FIG. 7 will be described. FIG. 7 shows an example of a transistor level circuit diagram for the bias circuit and the operational amplifier AMP4.

  In FIG. 7, a circuit BCR having transistors PMB1, PMB2, NMB1, NMB2, and a resistor RB1 functions as a bias circuit for generating bias potentials BNB and BPB. Note that the startup circuit is omitted. BPB and BNB are used as bias potentials for the operational amplifier circuit.

  The transistors PMB3, PMB4, PMB5, PMB6, NMB3, NMB4, and NMB5 operate as a general two-stage operational amplifier circuit and function as the operational amplifier AMP4 in FIG. With such an operational amplifier circuit, it is possible to generate a band gap voltage VBGR of 1.2V. The transistors PMB3, PMB5, PMB6, NMB3, and NMB4 are first-stage differential circuits of a two-stage operational amplifier circuit, and play a role of amplifying the gate potential difference between the transistors PMB5 and PMB6. PMB4 and NMB5 are second-stage source grounded amplifier circuits. The phase compensation capacitor CB1 is provided to limit the band of the first-stage differential circuit, and maintains the stability of feedback. Using the bias potential BPB generated in the bias circuit BCR, the currents of the transistors PMB3 and PMB4 serving as current sources are controlled. Accordingly, the transistors PMB3 to PMB6 and the transistors NMB3 to NMB5 can be operated as operational amplifier circuits. As described above, in the actual circuit, VOFF is short-circuited, and the operational amplifier circuit performs negative feedback control so that the potentials of the inputs IM2 and IP2 coincide with each other. Therefore, the gate potential of the transistor PMB6 becomes VBE3. . Since the potential of IP2 is relatively low, for example, about 0.6 V, FIG. 7 shows a circuit example using an operational amplifier circuit having a differential input stage with a PMOS transistor. Note that the startup circuit is omitted.

  FIG. 8 will be described. FIG. 8 shows an example of a transistor level circuit diagram of the operational amplifier AMP2.

  In FIG. 8, transistors PMB7, PMB8, PMB9, NMB6, NMB7, NMB8, and NMB9 operate as a general two-stage operational amplifier circuit and function as the operational amplifier AMP2 in FIG. Such an operational amplifier circuit can generate a band gap voltage VBGR24 of 2.4V. The transistors PMB7, PMB8, NMB6, NMB7, and NMB8 are first-stage differential circuits of a two-stage operational amplifier and play a role of amplifying the gate potential difference between the transistors NMB6 and NMB7. The transistors PMB9 and NMB9 are grounded source amplifier circuits. Using the bias potential BNB generated in the bias circuit BCR, the currents of the transistors NMB8 and NMB9 serving as current sources are controlled. Accordingly, the transistors PMB7 to PMB9 and the transistors NMB6 to NMB9 can be operated as operational amplifier circuits. As described above, in the actual circuit, VOFF 24 is short-circuited, and the operational amplifier circuit performs negative feedback control so that the potentials of IM and IP coincide with each other. Therefore, the gate potential of the transistor NMB7 is equal to the potential of IP. It will be the same. Since the IP potential is relatively high, for example, about 1.2 V, FIG. 8 shows an example of a circuit using an operational amplifier circuit having a differential input stage with NMOS transistors. Note that the startup circuit is omitted. The start-up circuit can be realized, for example, with a circuit configuration that supplies current to the IP so that the potential of the IP rises only when the potential of the VBGR 24 is at a potential near GND.

  As described above, the bandgap circuit of FIG. 6 can be realized by the circuit example described with reference to FIGS. 7 and 8 show circuit examples as an example, but the circuit configuration at the transistor level can be variously modified, and the startup circuit can be realized in various ways as long as it can fulfill the purpose of the startup circuit. It goes without saying that there is a way.

  Next, a circuit simulation result of the band gap circuit of FIG. 6 having the circuits shown in FIGS. 7 and 8 will be described with reference to FIGS.

  FIG. 9 shows the relationship between the output reference voltage VBGR and the temperature in the bandgap circuit of FIG. In FIG. 9, the vertical axis indicates the output reference voltage VBGR, and the horizontal axis indicates the temperature.

  FIG. 9 shows, as parameters, a graph in which the offset voltage VOFF is +10 mV, 0 mV, and −10 mV, and the offset voltage VOFF24 is +10 mV, 0 mV, and −10 mV. Referring to FIG. 9, in the ideal state, that is, when the offset voltages VOFF and VOFF24 are 0 mV, the output reference voltage VBGR is about 1.2 V regardless of the temperature. On the other hand, when the offset voltage VOFF is +10 mV, the output reference voltage VBGR increases, and when the offset voltage VOFF is −10 mV, the output reference voltage VBGR decreases. This is expected from what has been described with reference to FIGS.

  Here, the influence of the offset voltage VOFF24 on the output reference voltage VBGR will be described. As the offset voltage VOFF24 varies, the value of the emitter current flowing through the BJTQ4 and Q6 changes, and the value of the base current flowing through the BJTQ4 and Q6 also changes. Therefore, it is expected that the values of VBE3 and VBE5 will fluctuate and the output reference voltage VBGR will fluctuate. However, as can be seen from FIG. 9, even when the offset voltage VOFF24 varies, the output reference voltage VBGR hardly varies. For example, looking at the graph when the offset voltage VOFF = + 10 mV, the value of the output reference voltage VBGR is only about 0.005 V, regardless of whether the offset voltage VOFF24 = + 10 mV or −10 mV. It turns out that there is no big difference. That is, in the band gap circuit of FIG. 6, the influence of the offset voltage VOFF24 on the output reference voltage VBGR is relatively small, and the relationship between the output reference voltage VBGR and the offset voltage VOFF is the same as in the band gap circuit of FIG. It can be said.

  FIG. 10 shows the relationship between the output reference voltage VBGR24 and the temperature in the bandgap circuit of FIG. In FIG. 9, the vertical axis indicates the output reference voltage VBGR24, and the horizontal axis indicates the temperature.

  Also in FIG. 10, as parameters, a graph is shown in which the offset voltage VOFF is +10 mV, 0 mV, and −10 mV, and the offset voltage VOFF24 is +10 mV, 0 mV, and −10 mV. Referring to FIG. 10, in an ideal state, that is, when the offset voltages VOFF and VOFF24 are 0 mV, the output reference voltage VBGR24 is about 2.4 V regardless of the temperature. On the other hand, when the offset voltage VOFF24 is +10 mV, the output reference voltage VBGR24 increases, and when the offset voltage VOFF24 is −10 mV, the output reference voltage VBGR24 decreases. This is expected from what has been described with reference to FIGS.

  Here, the influence of the offset voltage VOFF on the output reference voltage VBGR24 will be described. As the offset voltage VOFF varies, the output reference voltage VBGR varies, and the emitter current flowing through the BJTQ3 and Q5 also varies. As a result, the values of VBE3 and VBE5 vary, and the output reference voltage VBGR24 is also expected to vary. However, as can be seen from FIG. 9, even when the offset voltage VOFF varies, the output reference voltage VBGR24 hardly varies. For example, looking at the graph when the offset voltage VOFF24 = + 10 mV, the difference in the value of the output reference voltage VBGR24 is only about 0.01 V, regardless of whether the offset voltage VOFF = + 10 mV or −10 mV. It turns out that there is no big difference. That is, in the band gap circuit of FIG. 6, the influence of the offset voltage VOFF on the output reference voltage VBGR24 is relatively small, and the relationship between the output reference voltage VBGR24 and the offset voltage VOFF24 is the same as that of the band gap circuit of FIG. It can be said.

  FIG. 11 shows the relationship between the output reference voltages VBGR and VBGR24 and the power supply voltage VDP5 in the band gap circuit of FIG. In FIG. 11, the vertical axis represents the output reference voltages VBGR and VBGR24, and the horizontal axis represents the power supply voltage VDP5. In FIG. 11, the offset voltages VOFF and VOFF24 are assumed to be zero.

  As can be seen from FIG. 11, when the power supply voltage VDP5 exceeds 1.2V, the output reference voltage VBGR is maintained at 1.2V. Therefore, it can be seen that the minimum operating voltage of the circuit CC1 that generates the output reference voltage VBGR in the bandgap circuit of FIG. 6 is 1.2V. When the power supply voltage VDP5 exceeds 2.4V, the output reference voltage VBGR24 is kept at 2.4V. Therefore, it can be seen that the minimum operating voltage of the circuit CC2 that generates the output reference voltage VBGR24 in the bandgap circuit of FIG. 6 is 2.4V.

  12 is a diagram showing the relationship between the output reference voltage VBGR and the power supply voltage VDP5 in the bandgap circuit of FIG. 6, and is an enlarged view of the vertical axis of FIG.

  As can be seen from FIG. 12, when the power supply voltage VDP5 exceeds 1.2V, the output reference voltage VBGR becomes 1.2V. To be precise, the output reference voltage VBGR gradually increases until the power supply voltage VDP5 changes from 1.2V to 2.4V, and when the power supply voltage VDP5 exceeds 2.4V, the output reference voltage VBGR is It becomes constant. That is, since the output reference voltage VBGR24 is not stable until the power supply voltage VDP5 exceeds 2.4V (see FIG. 11), the current flowing through the BJTQ4 and Q6 while the power supply voltage VDP5 changes from 1.2V to 2.4V. The value increases as the power supply voltage VDP5 increases. For this reason, while the power supply voltage VDP5 changes from 1.2V to 2.4V, VBE3 and VBE5 moderately depend on the power supply voltage VDP5, and the output reference voltage VBGR also shows some power supply dependency. However, it can be seen from FIG. 12 that the change in the output reference voltage VBGR at this time is slight, and the output reference voltage VBGR can be regarded as stable when the power supply voltage VDP5 exceeds 1.2V.

  FIG. 13 is a diagram illustrating the relationship between the output reference voltage VBGR24 and the power supply voltage VDP5 in the bandgap circuit of FIG. 6, and is an enlarged view of the vertical axis of FIG.

  As can be seen from FIG. 13, when the power supply voltage VDP5 exceeds 2.4V, the output reference voltage VBGR24 becomes 2.4V. Thereafter, even when the power supply voltage VDP5 rises, the output reference voltage VBGR24 is It is held constant at 2.4V.

  From FIG. 11 to FIG. 13, in the band gap circuit of FIG. 6, when the power supply voltage VDP5 exceeds 1.2V, the output reference voltage VBGR can be obtained, and when the power supply voltage VDP5 exceeds 2.4V. It can be seen that the output reference voltage VBGR24 can be obtained. In this manner, in the band gap circuit of FIG. 6, it is possible to obtain the output reference voltage VBGR that can suppress the minimum operating voltage and the output reference voltage VBGR24 that can improve accuracy.

[Application examples of band gap circuits]
Next, application examples of the band gap circuit according to the embodiment will be described.

  FIG. 14 is a circuit diagram of a microcontroller MCU1 as an example of a microcontroller to which the bandgap circuit according to the embodiment is applied.

  The microcontroller MCU1 includes a band gap circuit BGR1, a regulator circuit REG1, low voltage detection circuits LVDH1 and LVDL1, and a logic circuit LOGIC1 according to the embodiment shown in FIG. In FIG. 14, VBGR indicates an output reference voltage of 1.2V, and VBGR24 indicates an output reference voltage of 2.4V. VDP5 indicates a 5V + power supply, and GND indicates a 0V potential.

  The regulator circuit REG1 is a circuit that generates a constant voltage VDD from the power supply voltage VDP5. The logic circuit LOGIC1 is a circuit that operates using the voltage VDD generated by the regulator circuit REG1 as a power source. The low voltage detection circuit LVDH1 is a low voltage detection circuit that monitors the power supply voltage VDP5, and the low voltage detection circuit LVDL1 is a low voltage detection circuit that monitors the voltage VDD.

  In the microcontroller MCU1, it is desirable that the regulator circuit REG1 operates up to the lowest possible input voltage. Therefore, the 1.2V output reference voltage VBGR is used as the reference voltage of the regulator circuit REG1 and the reference voltage of the low voltage detection circuit LVDL1 that monitors the voltage VDD generated in the regulator circuit REG1. On the other hand, the output reference voltage VBGR24 of 2.4V is used as the reference voltage of the low voltage detection circuit LVDH1 that monitors the power supply voltage VDP5. This is because, for the low voltage detection circuit LVDH1, there is no problem even if the minimum operating voltage of the circuit that generates the reference voltage is larger than the output reference voltage VBGR24 of 2.4V. In this way, the reference voltage of the regulator circuit REG1 and the low voltage detection circuit LVDL1 is made as small as possible, and the low voltage detection circuit LVDH1 uses the output reference voltage VBGR24 to improve the accuracy of the reference voltage. it can. Hereinafter, the operation of the circuit of each unit will be described in detail.

  The regulator circuit REG1 generates a constant voltage VDD from the power supply voltage VDP5 and supplies the generated voltage VDD to the logic circuit LOGIC1. The regulator circuit REG1 includes a PMOS transistor PMO1, an error amplifier EAMP1, and resistors RR1 and RR2.

  In the regulator circuit REG1, the resistors RR1 and RR2 function as a voltage dividing circuit. VDIV1 indicates a divided voltage of the voltage dividing circuit. One end of the resistor RR1 is connected to the drain of the PMOS transistor PMO1, and is set to the output voltage VDD of the regulator circuit REG1. The error amplifier EAMP1 receives the output reference voltage VBGR from the band gap circuit BGR1 and the divided voltage VDIV. The output of the error amplifier EAMP1 is connected to the gate of the PMOS transistor PMO1.

  In the regulator circuit REG1, the error amplifier EAMP1, the PMOS transistor PMO1, and the resistor R1 operate as a feedback circuit that matches the output reference voltage VBGR and the divided voltage VDIV1. Since the divided voltage VDIV1 matches the output reference voltage VBGR, if the ratio of the values of the resistors RR1 and RR2 is designed to be 1: 2, for example, the voltage VDD is kept constant at 1.8V. Note that CO1 functions as a capacitor for stabilizing the voltage VDD provided outside the microcontroller MCU1.

  Such a microcontroller is often desirable to operate over a wide range of power supply voltages. For example, one example of a microcontroller system that is desired to operate over a wide range of power supply voltages is a system that operates on a battery. In such a case, the regulator circuit is also desired to keep the voltage to be generated constant within the widest possible range of the power supply voltage. That is, it is desirable to make the minimum operating voltage of the regulator circuit as low as possible. Therefore, it is desirable that the reference voltage of the regulator circuit is as low as possible. Therefore, in the microcontroller MCU1, the output reference voltage VBGR of the band gap circuit BGR1 is used as the reference voltage of the regulator circuit REG1.

  The low voltage detection circuit LVDL1 functions as a low voltage detection circuit for monitoring the voltage VDD. Specifically, the low voltage detection circuit LVDL1 detects whether or not the voltage VDD is lower than a predetermined voltage. As described above, since the lowest operating voltage of the regulator circuit REG1 is desired to be as low as possible, the reference voltage of the low voltage detection circuit LVDL1 that monitors the output voltage VDD is also desired to be as low as possible. For this purpose, in the microcontroller MCU1, the output reference voltage VBGR of the band gap circuit BGR1 is used as the reference voltage of the low voltage detection circuit LVDL1.

  The low voltage detection circuit LVDL1 includes a comparator CMP2 and resistors RL3 and RL4. LVDLOX1 indicates the output of LVDL1.

  In the low voltage detection circuit LVDL1, the resistors RL3 and RL4 operate as a voltage dividing circuit and divide the voltage VDD. VDIV3 indicates a divided voltage divided by the resistors RL3 and RL4. The divided voltage VDIV3 and the output reference voltage VBGR are input to the comparator CMP2. The comparator CMP2 outputs the comparison result as LVDLOX1. The low voltage detection circuit LVDL1 divides the voltage VDD by the resistors RL3 and RL4, and detects whether or not the voltage VDD is lower than a predetermined voltage by detecting whether or not the divided voltage VDIV3 is lower than the output reference voltage VBGR. To detect. For example, if the ratio of the values of the resistors RL3 and RL4 is designed to be 1: 3, the divided voltage VDIV3 becomes 3/4 of the voltage VDD. Therefore, the low voltage detection circuit LVDL1 detects whether or not the voltage VDD is lower than 1.6V by detecting whether or not the divided voltage VDIV3 is lower than the reference voltage using the output reference voltage VBGR as a reference voltage. It becomes possible to detect. Whether or not the divided voltage VDIV3 is lower than the reference voltage is determined by the comparator CMP2. In the comparator CMP2, when the divided voltage VDIV3 is lower than the output reference voltage VBGR, the output LVDLOX1 is “1”, and when the divided voltage VDIV3 is equal to or higher than the output reference voltage VBGR, the output LVDLOX1 is “0”. It becomes. That is, whether or not the voltage VDD is lower than 1.6V can be known from the value of the output LVDLOX1. When the low voltage detection circuit LVDL1 detects that the voltage VDD has become lower than a specified value (for example, 1.6 V), for example, an interrupt can be generated or a reset can be generated. .

  The low voltage detection circuit LVDH1 functions as a low voltage detection circuit for monitoring the power supply voltage VDP5. Specifically, the low voltage detection circuit LVDH1 detects whether or not the voltage VDP5 is lower than a predetermined voltage. For example, the low voltage detection circuit LVDH1 is used when an AD conversion circuit that is desirably operated at a power supply voltage of 3.6 V or more is mounted and the power supply voltage of a 5 V power supply is monitored for that purpose. .

  The low voltage detection circuit LVDH1 includes a comparator CMP1 and resistors RL1 and RL2. LVDHOX1 indicates the output of LVDH1.

  In the low voltage detection circuit LVDH1, the resistors RL1 and RL2 operate as a voltage dividing circuit and divide the power supply voltage VDP5. VDIV2 indicates a divided voltage divided by the resistors RL1 and RL2. The divided voltage VDIV2 and the output reference voltage VBGR24 are input to the comparator CMP1. The comparator CMP1 outputs the comparison result as LVDHOX1.

  The low voltage detection circuit LVDH1 divides the power supply voltage VDP5 by the resistors RL1 and RL2, and detects whether the divided voltage VDIV2 is lower than the output reference voltage VBGR24, whereby the power supply voltage VDP5 is lower than a predetermined voltage. Whether or not is detected. For example, if the ratio of the values of the resistors RL1 and RL2 is designed to be 1: 2, the divided voltage VDIV2 is 2/3 of the voltage VDP5. Therefore, the low voltage detection circuit LVDH1 uses the output reference voltage VBGR24 as a reference voltage to detect whether the divided voltage VDIV2 is lower than the reference voltage, thereby determining whether the power supply voltage VDP5 is lower than 3.6V. Can be detected. Whether or not the divided voltage VDIV2 is lower than the reference voltage is determined by the comparator CMP1. In the comparator CMP1, when the divided voltage VDIV2 is lower than the output reference voltage VBGR24, the output LVDHOX1 is “1”, and when the divided voltage VDIV2 is equal to or higher than the output reference voltage VBGR24, the output LVDHOX1 is “0”. It becomes. That is, whether or not the power supply voltage VDP5 is lower than 3.6 V can be known from the value of the output LVDHOX1. Similar to the low voltage detection circuit LVDL1, the low voltage detection circuit LVDH1 detects that the power supply voltage VDP5 has become lower than a specified value (eg, 3.6V). It is possible to generate a reset.

  When determining whether or not the power supply voltage VDP5 is lower than 3.6V, even if the minimum operating voltage of the circuit for generating the reference voltage is somewhat large, for example, about 2.7V, a voltage of 3.6V In many cases, there is no inconvenience. In such a case, rather, it is often desirable that the reference voltage for determining 3.6V has a high accuracy of the reference voltage rather than a low minimum operating voltage.

  For example, 5% of 3V is 150 mV, and 5% of 4V is 200 mV. If the absolute value of the power supply voltage to be determined is large and the error of the reference voltage is large, the absolute value of the error may be unacceptably large.

  Assuming that the accuracy of voltage division by the resistors RL1 and RL2 is sufficiently good, it is mainly the accuracy of the reference voltage that determines the determination accuracy of the power supply voltage VDP5. As an example, consider a case where the power supply voltage VDP5 is determined by comparing the divided voltage obtained by dividing the power supply voltage VDP5 by 1/3 with the output reference voltage VBGR. In this case, for example, if the error of the output reference voltage VBGR is 1.2V ± 5%, the accuracy when determining 3.6V is 3.6V ± 5%, that is, 3.6V ± 180 mV.

  On the other hand, consider the case where the potential of the voltage VDP5 is determined by comparing the divided voltage obtained by dividing the power supply voltage VDP5 by 2/3 with the output reference voltage VBGR24. In this case, for example, if the error of the output reference voltage VBGR24 is 2.4V ± 2.5%, the accuracy when determining 3.6V is 3.6V ± 2.5%, that is, 3.6V ± 90 mV.

  In this way, when the minimum operating voltage may be slightly higher in the low voltage detection circuit, the accuracy of the low voltage detection circuit is improved by using the output reference voltage VBGR24 as shown in FIG. The effect that can be obtained.

  Further, when the operation and stop of the AD conversion circuit is controlled by the low voltage detection circuit LVDH1, it is possible to relax the operation voltage requirement for the AD conversion circuit by improving the accuracy of the low voltage detection circuit. This will be specifically described below.

  In the low voltage detection circuit LVDH1, for example, in order to determine a voltage of 3.6V using the output reference voltage VBGR, the detection width of 3.6V is actually from 3.6V-180 mV to 3.V. 6V + 180mV. Accordingly, the operation of the AD converter circuit can be reliably stopped at 3.42V, and the voltage that can be reliably used by the AD converter circuit is higher than 3.78V.

  On the other hand, in the low voltage detection circuit LVDH1, for example, in order to determine the voltage of 3.6V using the output reference voltage VBGR24, the detection width of 3.6V is actually 3.6V-90mV. To 3.6V + 90 mV. Therefore, the operation of the AD converter circuit can be surely stopped at 3.51 V, and the voltage that can be reliably used by the AD converter circuit is higher than 3.69 V.

  That is, when 3.6 V is determined using the output reference voltage VBGR, the minimum voltage for determination is 3.42 V and the maximum voltage is 3.78 V. For this reason, when used to use the AD converter circuit, the AD converter circuit needs to operate at a minimum voltage of 3.42V, and may not be used unless the power supply voltage exceeds 3.78V. On the other hand, when 3.6 V is determined using the output reference voltage VBGR24, the minimum voltage for determination is 3.51 V and the maximum is 3.69 V, so that the AD converter circuit is operated at a lower voltage than necessary. No need to design. That is, it can be used from a voltage close to the minimum operating voltage of the AD converter circuit.

  In this way, by improving the voltage detection accuracy of the low voltage detection circuit using the output reference voltage VBGR24, it becomes possible to relax the operating voltage requirement for the target circuit to be controlled.

  In the microcontroller MCU1 of FIG. 14, the bandgap circuit according to the embodiment shown in FIG. 6 is used as the bandgap circuit BGR1, but the present invention is not limited to this. Instead of using the band gap circuit BGR1, the output reference voltages VBGR and VBGR24 may be generated using the band gap circuit of FIG. 1 and the band gap circuit of FIG. This also makes it possible to both maintain the minimum operating voltage in the regulator circuit REG1 and the low voltage detection circuit LVDL1 as described above and improve the accuracy of the power supply voltage in the low voltage detection circuit LVDH1.

  In the microcontroller MCU1 of FIG. 14, the example in which the output reference voltage VBGR24 is used as the reference voltage of the low voltage detection circuit LVDH1 is shown, but the present invention is not limited to this. Instead of doing this, the output reference voltage VBGR24 and the output reference voltage VBGR may be switched and used. Moreover, although the example using output reference voltage VBGR as a reference voltage of regulator circuit REG1 was shown, it is not restricted to this. Instead of doing this, the output reference voltage VBGR24 and the output reference voltage VBGR may be switched and used. Hereinafter, a circuit example of a low voltage detection circuit capable of switching the output reference voltage will be described.

  First, a circuit example of a low voltage detection circuit capable of switching the output reference voltage can be described.

  FIG. 15 is an example of a circuit diagram of a low voltage detection circuit for monitoring the power supply voltage VDP5 capable of switching the output reference voltages VBGR and VBGR24.

  In FIG. 15, the same elements as those described in FIG. 14 are denoted by the same reference numerals. In FIG. 15, RLHn (n is an integer) indicates a resistance and a resistance value, SWn (n is an integer) indicates a switch, and CMPH1 indicates a comparator. The power supply voltage VDP5 is divided by a voltage dividing circuit having resistors RLH1, RLH2, RLH3, and RLH4. VDIVH1 and VDIVH2 indicate divided voltages divided by the voltage dividing circuit. Any one of the divided voltages VDIVH1 and VDIVH2 is selected by the switches SW1 and SW2. VTL indicates a divided voltage selected by the switches SW1 and SW2. The output reference voltages VBGR and VBGR24 are selected by the switches SW3 and SW4. VRRF indicates a reference voltage selected by the switches SW3 and SW4. The voltages VTL and VRFF are input to the comparator CMPH1. Accordingly, the switches SW1 and SW2 function as a first switch circuit, and the switches SW3 and SW4 function as a second switch circuit. LVDHOX1 is the output of the comparator CMPH1.

  The low voltage detection circuit of FIG. 15 switches between a case where it is detected whether the power supply voltage VDP5 is lower than 2.4V and a case where it is detected whether the power supply voltage VDP5 is lower than 3.6V. It is a possible circuit. This will be specifically described below.

  In the low voltage detection circuit of FIG. 15, the values of the resistors RLH1, RLH2, RLH3, and RLH3 are 100 kΩ, 50 kΩ, 50 kΩ, and 100 kΩ. The divided voltage VDIVH1 is a voltage divided by the resistor RLH1 and the sum of the resistors RLH2, RLH3, and RLH4. That is, the divided voltage VDIVH1 is a voltage obtained by dividing the power supply voltage VDP5 by 2/3. Therefore, when the switch SW1 is turned on and the switch SW2 is turned off, the voltage VTL becomes a value obtained by dividing the power supply voltage VDP5 by 2/3. Here, when the switch SW3 is turned off and the switch SW4 is turned on, the voltage VREF becomes the output reference voltage VBGR24. By comparing the output reference voltage VBGR24 of 2.4V and the voltage VTL obtained by dividing the power supply voltage VDP5 by 2/3 by the comparator CMPH1, it is determined whether or not the power supply voltage VDP5 is lower than 3.6V. Can do.

  The divided voltage VDIVH2 is a voltage divided by the sum of the resistors RLH1 and RLH2 and the sum of the resistors RLH3 and RLH4. That is, the divided voltage VDIVH2 is a voltage obtained by dividing the power supply voltage VDP5 by half. Therefore, when the switch SW1 is turned off and the switch SW2 is turned on, the voltage VTL becomes a value obtained by dividing the power supply voltage VDP5 by half. Here, when the switch SW3 is turned on and the switch SW4 is turned off, the voltage VREF becomes the output reference voltage VBGR. Comparing the output reference voltage VBGR of 1.2V and the voltage VTL obtained by dividing the power supply voltage VDP5 by 1/2 with the comparator CMPH1 determines whether or not the power supply voltage VDP5 is lower than 2.4V. Can do.

  According to the low voltage detection circuit of FIG. 15, when the power supply voltage VDP5 is relatively large so as to determine whether or not the power supply voltage VDP5 is lower than 3.6V, the output reference voltage VBGR24 with high accuracy is used. Can be obtained. On the other hand, when the power supply voltage VDP5 is relatively small so as to determine whether or not the power supply voltage VDP5 is lower than 2.4V, the power supply voltage VDP5 can be operated in a wider power supply voltage range than the output reference voltage VBGR24. It becomes possible. In addition, since the output reference voltage VBGR is stabilized earlier than the output reference voltage VBGR24, the time required for the determination can be shortened in this way. Here, “power supply voltage VDP5 is relatively large” indicates, for example, that power supply voltage VDP5 is equal to or higher than output reference voltage VBGR24, and “power supply voltage VDP5 is relatively small” means, for example, power supply voltage It indicates that VDP5 is less than the output reference voltage VBGR24.

  Here, the comparator CMPH1 will be described with reference to FIG. FIG. 16 shows an example of a transistor level circuit diagram of the comparator CMPH1.

  In FIG. 16, PMCn (n is an integer) indicates a PMOS transistor, NMCn (n is an integer) indicates an NMOS transistor, GND indicates a GND terminal, and VDP5 indicates, for example, a 5V + power supply. CIM and CIP indicate inputs of the comparator circuit, CMPO indicates an output, and BNB indicates a bias potential. The circuit elements and nodes corresponding to FIGS. 7 and 15 are given the same element names and node names.

  FIG. 16 shows an example in which the comparator is operated using the bias potential BNB generated by the bias circuit BCR shown in FIG. The transistor NMC5 biases a constant current using the bias potential BNB generated by the bias circuit BCR. Since the transistors NMC1 and NMC2 have the same source potential, the transistors NMC1 and NMC2 function to pass a current corresponding to the difference between the gate potentials. That is, the transistors NMC1 and NMC2 share a constant current of the transistor NMC5 according to the gate potential. The transistors PMC1 and PMC2 function as a load for the differential pair. The transistors PMC1 and PMC3 operate as current mirrors, and the transistors PMC2 and PMC4 also operate as current mirrors. Therefore, the transistor PMC3 mirrors the drain current of the transistor PMC1, and the transistor PMC4 mirrors the drain current of the transistor PMC2. Transistors NMC3 and NMC4 also operate as current mirrors, and the current in transistor NMC3 is mirrored to transistor NMC4. Eventually, in the output CMPO, the currents of the transistors NMC1 and NMC2 are added. In this way, the circuit of FIG. 16 functions as a comparator that compares the potentials of the inputs CIM and CIP. For example, the circuit of FIG. 16 can realize the function of the comparator CMPH1 of FIG. 15 and can be used in combination with the circuits of FIGS.

  FIG. 17 is another example of a circuit diagram of a low voltage detection circuit for monitoring the power supply voltage VDP5 that can switch the output reference voltages VBGR and VBGR24. FIG. 15 shows a circuit example in which the reference voltage is switched by a switch and used as a low voltage detection circuit for monitoring the power supply voltage. On the other hand, FIG. 17 shows a circuit example in which a plurality of comparator circuits are prepared.

  In FIG. 17, the same elements as those described in FIG. 15 are denoted by the same reference numerals. CMPH2 and CMPH3 represent comparators, and OR1 represents a logical sum (OR) circuit. The divided voltage VDIVH1 and the output reference voltage VBGR24 are input to the comparator CMPH2. CMPH2O indicates the output of the comparator CMPH2. The divided voltage VDIVH2 and the output reference voltage VBGR are input to the comparator CMPH3. LVDHO3 indicates the output of the comparator CMPH3. CMPH2O and LVDHO3 are input to the OR circuit OR1. LVDHO2 indicates the output of the OR circuit OR1. Note that LVDHO2 and LVDHO3 are also outputs of the low voltage detection circuit. Note that the comparator CMPH2 functions as a first comparison circuit, and the comparator CMPH3 functions as a second comparison circuit. This will be specifically described below.

  As described in FIG. 15, the divided voltage VDIVH1 is a value obtained by dividing the power supply voltage VDP5 by 2/3. It is possible to determine whether or not the power supply voltage VDP5 is lower than 3.6V by comparing the output reference voltage VBGR24 with the divided voltage VDIVH1 obtained by dividing the power supply voltage VDP5 by 2/3 by the comparator CMPH2. it can.

  The divided voltage VDIVH2 is a value obtained by dividing the power supply voltage VDP5 by half. It is possible to determine whether or not the power supply voltage VDP5 is lower than 2.4V by comparing the output reference voltage VBGR and the divided voltage VDIVH2 obtained by dividing the power supply voltage VDP5 by 1/2 by the comparator CMPH3. it can.

  The LVDHO3 becomes “H” (High) when the power supply voltage VDP5 is less than 2.4V, and becomes “L” (Low) when the power supply voltage VDP5 is 2.4V or more. CMPH2O becomes “H” when the power supply voltage VDP5 is less than 3.6V, and becomes “L” when the power supply voltage VDP5 is 3.6V or more. Since the minimum operating voltage of the circuit that generates the output reference voltage VBGR24 is 2.4 V or more, when the power supply voltage VDP5 is less than 2.4 V, the output reference voltage VBGR24 is not stable, and the determination result CMPH2O of the comparator CMPH2 is correct. There is no possibility. In such a case, the low voltage detection circuit of FIG. 17 can supplement the determination result based on the output reference voltage VBGR24 by using the determination result of the circuit operating at a lower power supply voltage.

  For example, when the power supply voltage VDP5 is 3.6 V or more, CMPH2O becomes “L” and LVDHO3 becomes “L”, so that LVDHO2 becomes “L”. When the power supply voltage VDP5 is 2.4 V or more and less than 3.6 V, CMPH2O becomes “H” and LVDHO3 becomes “L”, so that LVDHO2 becomes “H”. When the power supply voltage VDP5 is less than 2.4V, the value of CMPH2O is not reliable. However, in this case, since LVDHO3 becomes “H”, LVDHO2 becomes “H” regardless of the value of CMPH2O.

  That is, when LVDHO3 is “L” and LVDHO2 is “L”, it can be determined that the power supply voltage VDP5 is 3.6 V or higher. When the LVDHO3 is “L” and the LVDHO2 is “H”, it can be determined that the power supply voltage VDP5 is 2.4 V or more and less than 3.6 V. When LVDHO3 is “H” and LVDHO2 is “H”, it can be determined that the power supply voltage VDP5 is less than 2.4V.

  According to the low voltage detection circuit of FIG. 17, even when the power supply voltage VDP5 is less than 2.4 V and the comparator CMPH2 shows an incorrect result, it is possible to prevent erroneous determination of the power supply voltage VDP5. It is possible to accurately determine the voltage VDP5.

  Note that the output reference voltage VBGR is not stable when the power supply voltage VDP5 is less than 1.2 V, and an error may occur in the determination result of the comparator CMPH3. However, also in this case, it is possible to prevent erroneous determination by using the same method as in the case of the output reference voltage VBGR24 described above. In the low voltage detection circuit of FIG. 17, an OR circuit is used. However, the present invention is not limited to this, and a logical product (AND) circuit may be used instead.

  As described above, the low voltage detection circuit using the output reference voltages VBGR24 and VBGR can also be realized by the low voltage detection circuit of FIG. 17, and the same effect as the low voltage detection circuit of FIG. 15 can be obtained. Can do.

  In the low voltage detection circuit of FIGS. 15 and 17, the case where there are two voltage values to be determined is shown, but the present invention is not limited to this. Instead of doing this, the circuit concept of the invention can be applied to a circuit having an arbitrary number of determination voltages. Needless to say, the low voltage detection circuit of FIG. 15 and the low voltage detection circuit of FIG. 17 may be combined.

[Modification]
Next, a modified example of the band gap circuit will be described.

  FIG. 18 is an example of a circuit diagram of a band gap circuit according to a modification. In FIG. 18, the circuit elements and nodes corresponding to FIG. 6 are given the same element names and node names. The numerical value attached to the resistance indicates an example of the resistance value. In FIG. 6, RBDn (n is an integer) indicates a resistance, and VBGR2 indicates an output reference voltage.

  In the band gap circuit of FIG. 6, the band gap circuit that outputs the output reference voltages VBGR24 and VBGR is shown. However, in addition to this, it is also possible to generate the output reference voltage VBGR2 of 1.2 V by dividing the potential of the output reference voltage VBGR24 by ½. For example, when the output reference voltage VBGR24 is used as the reference voltage of the regulator circuit, it is more convenient to use the output reference voltage VBGR24 after processing it as a reference potential of 1.2V, which is the same as the output reference voltage VBGR, rather than using the output reference voltage VBGR24 as it is. May be good. In this case, a 1.2 V output reference voltage VBGR2 generated based on the output reference voltage VBGR24 may be used by using the band gap circuit of FIG. Since the output reference voltage VBGR2 is generated from the output reference voltage VBGR24, the voltage is not stable unless the power supply voltage reaches 2.4V or more, but the error caused by the offset voltage of the operational amplifier is smaller than the output reference voltage VBGR. The advantage of 1/2 is obtained.

  The difference between the circuit of FIG. 6 and the circuit of FIG. 18 is the resistors RBD1 and RBD2 that function as voltage dividing circuits, so the operation of this part will be described.

  As is apparent from FIG. 18, the divided voltage of the voltage dividing circuit having the resistors RBD1 and RBD2 is assumed to be VBGR2. If the resistance values of the resistors are designed to be equal, an output reference voltage VBGR2 of 1.2V can be obtained. Since the relative accuracy of the resistors can generally be expected to have an error of less than 0.5%, the error of the voltage dividing circuit itself is sufficiently small relative to the error of the VBGR 24.

  By adopting such a configuration, it is possible to use the 1.2V output reference voltage VBGR2 with higher accuracy than the output reference voltage VBGR. The output reference voltage VBGR2 functions as a third band gap voltage.

  In the regulator circuit and the low voltage detection circuit shown in FIG. 14 and the low voltage detection circuit shown in FIGS. 15 and 17, the output reference voltage VBGR2 may be used as the reference voltage instead of the output reference voltage VBGR. In this case, since the value of the reference voltage itself is 1.2 V for both VBGR and VBGR2, it is only necessary to switch the output reference voltage VBGR to the output reference voltage VBGR2. Further, in the low voltage detection circuit of FIGS. 15 and 17, an example in which the low voltage detection circuit is operated by using the output reference voltage VBGR24 is shown for the principle explanation. The voltage VBGR2 may be used. This also provides an effect that the high accuracy of the reference voltage can be used as in the case where the output reference voltage VBGR24 is used. Further, in the low voltage detection circuit of FIGS. 15 and 17, a circuit using the output reference voltage VBGR24 and a circuit using the output reference voltages VBGR and VBGR2 can be mixed and combined. For example, in the low voltage detection circuit of FIGS. 15 and 17, instead of inputting the output reference voltage VBGR, the output voltage of the switch circuit that switches and outputs the output reference voltages VBGR and VBGR2 may be input. Note that a configuration for switching the output reference voltages VBGR and VBGR2 in the regulator circuit will be described with reference to FIG.

  As described above, by preparing the output reference voltage VBGR2 obtained by dividing the output reference voltage VBGR24, the effect of improving the accuracy of the reference voltage can be obtained.

  Next, a circuit example of a regulator circuit capable of switching the output reference voltages VBGR and VBGR2 will be described.

  FIG. 19 is an example of a circuit diagram of a regulator circuit that is connected to a band gap circuit according to a modification and can switch the output reference voltages VBGR and VBGR2.

  The circuit elements and nodes corresponding to FIG. 14 are given the same element names and node names. SWn (n is an integer or the like) indicates a switch, and CTRL1 indicates a control signal for the switches SW5 and SW6.

  In the regulator circuit of FIG. 19, unlike the regulator circuit REG1 in the microcontroller of FIG. 14, the output reference voltage VBGR selected by the switches SW5 and SW6 or the output reference voltage VBGR2 is input to the negative input of EAMP1.

  For example, when the power supply voltage VDP5 is less than 2.8 V, the switch SW5 is turned on and the switch SW6 is turned off. On the other hand, when the power supply voltage VDP5 is 2.8 V or higher, the switch SW6 is turned on and the switch SW5 is turned off. By adopting such a configuration, when the power supply voltage VDP5 is relatively high, a highly accurate VBGR2 is used, and when the power supply voltage VDP5 is relatively low, it is generated in a circuit that suppresses the minimum operating voltage. It becomes possible to use VBGR. As a result, as in the circuits of FIGS. 15 and 17, when the power supply voltage is relatively small, the characteristics of operating in a wide power supply voltage range using a circuit that suppresses the minimum operating voltage and the power supply voltage are relatively high. In such a case, it is possible to achieve both characteristics that allow the use of a highly accurate reference voltage.

  In addition, as shown in FIG. 19, the low voltage detection circuit LVDH1 may determine the power supply voltage of VDP5 and use the control signal CTRL1 to control the on / off of the switches SW5 and SW6. Needless to say. FIG. 19 shows an example in which the output reference voltages VBGR and VBGR2 are selectively used with a power supply voltage of 2.8 V as a boundary, but the present invention is not limited to this. Instead of doing this, it goes without saying that the output reference voltage VBGR2 is stable and can be designed to various voltages as required in consideration of the range of voltage determination.

  By preparing the output reference voltage VBGR2 as shown in FIG. 18 and using it as in the circuit of FIG. 19, there is an advantage that the regulator circuit REG1 itself need not be changed. Both of the output reference voltages VBGR and VBGR2 are 1.2 V reference voltages, and the difference is only in accuracy and the usable power supply voltage range.

  The embodiments are not limited to the above-described embodiments, and can be appropriately changed without departing from the spirit or concept of the invention that can be read from the claims and the entire specification.

Q3, Q4, Q5, Q6 PNP transistors R6, R7, R8, R10, R11, R12 Resistors AMP2, AMP4 Operational amplifier

Claims (7)

The first and second PNP transistors, the first resistor having one end connected to the emitter of the second PNP transistor, and the emitter of the first PNP transistor and the other end of the first resistor are input. A first reference voltage generation circuit having a first operational amplifier connected thereto;
A base is connected to an emitter of the first PNP transistor, a second PNP transistor, a third PNP transistor whose base is connected to an emitter of the first PNP transistor, and an emitter of the second PNP transistor. The fourth PNP transistor, the second resistor having one end connected to the emitter of the fourth PNP transistor, and the emitter of the third PNP transistor and the other end of the second resistor are connected to the input. A second reference voltage generation circuit having a second operational amplifier,
The first reference voltage generation circuit makes the current densities of the first and second PNP transistors different from each other and controls the potential of the emitter of the first PNP transistor by negative feedback control of the first operational amplifier. And the potential of the other end of the first resistor are made equal to generate a first band gap voltage,
The second reference voltage generation circuit makes the current densities in the third and fourth PNP transistors different from each other and controls the potential of the emitter of the third PNP transistor by negative feedback control of the second operational amplifier. And a potential of the other end of the second resistor are made equal to each other to generate a second bandgap voltage. The first reference voltage generation circuit includes third and fourth resistors having different resistance values,
One ends of the third and fourth resistors are connected to the input of the first operational amplifier,
2. The bandgap circuit according to claim 1, wherein the other ends of the third and fourth resistors are connected to a voltage line set to the first bandgap voltage.   2. A voltage dividing circuit that divides the second band gap voltage to generate a divided voltage equal to the first band gap voltage and outputs the divided voltage as the third band gap voltage. A band gap circuit according to claim 1. A low voltage detection circuit that is connected to the band gap circuit according to claim 1 and that monitors a power supply voltage supplied from the outside,
A voltage dividing circuit that divides the power supply voltage to generate and output first and second divided voltages;
A first switch circuit that selects and outputs a predetermined divided voltage from the first and second divided voltages;
A second switch circuit that selects and outputs a predetermined reference voltage from the first and second band gap voltages;
A low voltage detection circuit comprising: a comparison circuit that compares the output of the first switch circuit and the output of the second switch circuit. A low voltage detection circuit that is connected to the band gap circuit according to claim 1 and that monitors a power supply voltage supplied from the outside,
A voltage dividing circuit that divides the power supply voltage to generate and output first and second divided voltages;
A first comparison circuit for comparing the first divided voltage and the first band gap voltage;
A second comparison circuit for comparing the second divided voltage and the second band gap voltage;
A low voltage detection circuit comprising: a logic circuit that takes a logical sum or a logical product of an output of the first comparison circuit and an output of the second comparison circuit. A regulator circuit connected to the band gap circuit according to claim 3 and generating a predetermined constant voltage from a power supply voltage supplied from the outside,
A regulator circuit having a switch circuit that selects one of the first and third band gap voltages as a reference voltage. A microcontroller having a band gap circuit according to claim 1, a low voltage detection circuit for monitoring a power supply voltage supplied from the outside, and a regulator circuit for generating a predetermined constant voltage from the power supply voltage,
The regulator circuit uses the first band gap voltage as a reference voltage,
The microcontroller, wherein the low voltage detection circuit uses the second band gap voltage as a reference voltage.

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