JP4710259B2 - Motor control device and control method thereof - Google Patents

Description

  The present invention provides a motor control device that feeds an alternating current to a three-phase AC motor by driving an inverter using a pulse width modulation method, and performs feedback control so that the current value of the three-phase AC motor follows a current command value, and the control thereof Regarding the method.

Conventionally, an AC current is supplied to a three-phase AC motor (hereinafter abbreviated as a motor) by switching on / off of a power element constituting an inverter by a PWM (Pulse Wide Modulation) method, and the motor There is known a motor control device that performs proportional integral (PI) control so that the current value of the current follows the current command value. In such a motor control device, the total loss per pulse of the power element is represented by the sum of the switching loss and the steady loss, and if the total loss is large, the power element may generate heat and may be thermally damaged. There is. From such a background, a method of reducing the carrier (switching) frequency has been proposed in order to reduce the total loss of the power element (see, for example, Patent Document 1).
JP 2000-134990 A

  However, in general, when the carrier period is synchronized with the control period, the upper limit value of the current feedback gain in the PI control also decreases as the carrier frequency decreases, so the conventional method of simply reducing the carrier frequency is used. Therefore, when the carrier frequency is switched, the current behavior of the motor may become unstable. Furthermore, when the carrier frequency is switched, the rotation angle of the motor that advances in one control cycle changes. Therefore, according to the conventional method, the predicted rotation angle of the motor in the PI control is actually Therefore, the switching timing of the power element may be deviated, resulting in disturbance of the current waveform and torque shock.

  The present invention has been made to solve the above-described problems, and an object of the present invention is to provide a motor control device capable of reducing the total loss of a power element without causing disturbance of a current waveform or torque shock, and It is in providing the control method.

In order to solve the above-described problems, the motor control device and the control method thereof according to the present invention change the feedback gain used in the feedback control according to the carrier frequency when switching the carrier frequency, so that the carrier frequency is low. Indeed, when switching the carrier frequency by lowering the proportional gain, the integral gain is determined by the crossing angular frequency and the calculation cycle. When the carrier frequency is switched from high frequency to low frequency, the integral gain at low frequency is high frequency. is greater than the integral gain when, after switching the carrier frequency from a high frequency to a low frequency, to change the integral gain from the integral gain at a high frequency to the integral gain at a low frequency.

According to the motor control device and the control method thereof according to the present invention, when switching the carrier frequency, the feedback gain that changes with the change of the carrier frequency is changed according to the carrier frequency, and the proportional gain decreases as the carrier frequency decreases. When the carrier frequency is switched, the integral gain is obtained from the crossing angular frequency and the calculation cycle. When the carrier frequency is switched from the high frequency to the low frequency, the integral gain at the low frequency is larger than the integral gain at the high frequency. In this case, after switching the carrier frequency from high frequency to low frequency, the integral gain is changed from the integral gain at the high frequency to the integral gain at the low frequency, so there is no disturbance of the current waveform or torque shock. Total loss can be reduced.

  Hereinafter, the configuration and operation of a motor control device according to an embodiment of the present invention will be described with reference to the drawings.

[Configuration of motor controller]
As shown in FIG. 1, a motor control device 1 according to an embodiment of the present invention includes a three-phase AC motor (motor) 2 that drives a vehicle, an inverter (INV) 3, a θ, ω calculation unit 4, and 3 A phase-to-two-phase conversion unit 5, a current PI control unit 6, a two-phase to three-phase conversion unit 7, a duty (DUTY) command unit 8, and a control unit 9 are provided as main components.

  The inverter 3 includes a plurality of power elements that select a positive electrode or a negative electrode of a DC voltage source such as a battery and connect the selected electrodes to the U-phase, V-phase, and W-phase electrodes of the motor 2. Then, the inverter 3 converts the DC voltage into the AC voltage by switching on / off of the plurality of power elements by the PWM method according to the duty commands Du *, Dv *, Dw * input from the duty command unit 8, A three-phase alternating current is supplied to the motor 2.

  The θ, ω calculation unit 4 calculates the current rotation angle θ of the motor 2 and the estimated rotation angle θ ′ of the motor 2 in the next calculation cycle for each calculation cycle in accordance with an instruction from the control unit 9 and the calculated rotation. The rotational speed ω of the motor 2 is calculated using the angle θ. Then, the θ, ω calculation unit 4 calculates the calculated rotation angle θ, estimated rotation angle θ ′, and rotation speed ω, respectively, as a three-phase → two-phase conversion unit 5, a two-phase → three-phase conversion unit 7, and a current PI control unit. 6 The three-phase → two-phase conversion unit 5 converts the actual currents Iu, Iv, Iw of the motor 2 into the actual currents of the d axis and the q axis based on the rotation angle θ of the motor 2 input from the θ, ω calculation unit 4. Id and Iq are converted and input to the current PI controller 6.

  As shown in FIG. 2, the current PI control unit 6 includes a difference calculation unit 11 that calculates the difference between the actual currents Id and Iq and the current command values Id * and Iq *, and a proportional gain adjustment unit according to the control of the control unit 9. The proportional term calculation unit 13 that calculates the proportional term in PI control using the proportional gain instructed by 12 and the integral term in PI control using the integral gain instructed by the integral gain adjusting unit 14 according to the control of the control unit 9 And an integral term calculation unit 15 for calculating the D-axis and the PI-axis so that the actual currents Id and Iq of the d-axis and the q-axis coincide with the d-axis current command value Id * and the q-axis current command value Iq *, respectively. To generate voltage command values Vd * and Vq * for the d-axis and the q-axis. The current PI control unit 6 inputs the generated d-axis and q-axis voltage command values Vd * and Vq * to the two-phase → three-phase conversion unit 7.

  The two-phase → three-phase conversion unit 7 converts the voltage command values Vd * and Vq * into the three-phase voltage command values Vu * and Vv * based on the estimated rotation angle θ ′ of the motor calculated by the θ and ω calculation unit 4. , Vw * and the three-phase voltage command values Vu *, Vv *, Vw * are input to the duty command unit 8. The control unit 9 generates a d-axis current command value Id * and a q-axis current command value Iq * in response to a torque request from the vehicle, and inputs the d-axis current command value Iq * to the current PI control unit 6. As will be described in detail later, the control unit 9 controls the proportional gain adjustment unit 12 and the integral gain adjustment unit 14 in response to an instruction to switch the carrier frequency, thereby controlling the proportional gain and integration in the current PI control. The magnitude of the gain is switched in stages, and the calculation method of the estimated rotation angle θ ′ is switched in stages by controlling the θ, ω calculator 4.

  The motor control device 1 having such a configuration prevents the current waveform from being disturbed and the occurrence of a torque shock when the carrier frequency is switched by the control unit 9 operating as described below. Hereinafter, the operation of the control unit 9 at the time of switching the carrier frequency will be described in detail with reference to the timing chart shown in FIG.

[Carrier frequency switching processing]
In general, the proportional gain used in PI control is represented by the product of the coil inductance and the crossing angular frequency, and the integral gain is represented by the product of the winding resistance, the crossing angular frequency, and one calculation cycle (time). . Moreover, the upper limit value of the controllable crossing angular frequency is reduced as the carrier frequency is reduced. Therefore, when switching the carrier frequency, it is necessary to change the proportional gain and the integral gain as the current feedback gain. Further, since the timing at which the duty command value calculated in the current calculation cycle is reflected in the PWM pulse is the next calculation cycle, the θ, ω calculation unit 4 calculates the next calculation from the current rotation angle θ of the motor 2. The estimated rotation angle θ ′ in the cycle must be calculated. However, when the carrier frequency changes, the rotation angle θ of the motor 2 that advances in one calculation cycle (that is, the increase amount Δθ of the rotation angle) changes. Therefore, in the normal calculation method, the estimated rotation angle θ ′ is accurately set. It cannot be calculated. Therefore, in the motor control device 1 according to the embodiment of the present invention, the control unit 9 operates as described below in accordance with the switching of the carrier frequency, thereby switching the magnitudes of the proportional gain and the integral gain in stages. At the same time, the calculation method of the estimated rotation angle θ ′ is switched stepwise. Hereinafter, the operation of the control unit 9 will be described in detail in the order of the operation at the time of gain switching and the operation at the time of switching the calculation method.

[Operation at gain switching]
When the carrier frequency is switched from a high frequency to a low frequency, the proportional gain decreases as the crossing angular frequency decreases, but the integral gain is determined by the product of the crossing angular frequency and the calculation period when switching the carrier frequency. Therefore, when the carrier frequency is switched from a high frequency to a low frequency, the control unit 9 determines the magnitude relationship between the integral gain at the time of the low frequency carrier and the integral gain at the time of the high frequency carrier, and is shown below according to the magnitude relationship. Switch the gain so that In this embodiment, only the operation at the time of switching the carrier frequency from the high frequency to the low frequency will be described, but it is of course applicable to the case where the carrier frequency is switched from the low frequency to the high frequency by performing the reverse procedure. .

(1) When the integral gain at the time of the low frequency carrier is larger than the integral gain at the time of the high frequency carrier The control unit 9 sends the carrier switching mode signal (I) to the proportional gain adjusting unit 12 before switching the carrier frequency from the high frequency to the low frequency. The proportional gain is changed to a low gain value by inputting. Then, the control unit 9 changes the integral gain to a high gain value by inputting the carrier switching mode signal (II) to the integral gain adjusting unit 14 in accordance with switching the carrier frequency from a high frequency to a low frequency.

(2) When the integral gain at the time of the low frequency carrier is smaller than the integral gain at the time of the high frequency carrier The control unit 9 transfers the carrier frequency to the proportional gain adjustment unit 12 and the integral gain adjustment unit 14 before switching the carrier frequency from the high frequency to the low frequency. By inputting the switching mode signal (I), the proportional gain and the integral gain are changed to low gain values. Then, after changing the proportional gain and the integral gain to the low gain value, the control unit 9 switches the carrier frequency from the high frequency to the low frequency.

[Operation when switching calculation method]
FIG. 3 shows a case where the carrier frequency is reduced from 10 [kHz] to 2.5 [kHz] (time T = t2) and a case where the carrier frequency is increased from 2.5 [kHz] to 10 [kHz] (time). It is a timing chart for demonstrating operation | movement of the control part 9 of T = t8).

  When the carrier frequency is 10 [kHz], the timing at which the duty command value calculated this time is reflected in the PWM pulse is the next calculation timing (when the duty value is calculated at time T = t1 shown in FIG. 3). Is time T = t2) after 100 [μs], and its pulse center is 150 [μs] after time T = t1. For this reason, the θ, ω calculation unit 4 uses an angle 150 [μs] after the rotation angle θ of the motor 2 detected this time at the current calculation timing. Specifically, the control unit 9 calculates the deviation Δθ between the previous rotation angle θz and the current rotation angle θ, the deviation Δθz (deviation between the previous rotation angle and the previous rotation angle) and the current time at the previous calculation timing. The average value of the deviation Δ at the calculation timing is calculated. Then, the control unit 9 uses an angle obtained by adding 1.5 times the calculated average value to the current rotation angle θ as an estimated rotation angle θ ′ at the current calculation timing.

  Similarly, when the carrier frequency is 2.5 [kHz], the timing at which the duty command value calculated this time is reflected in the PWM pulse is the next calculation timing (the duty value at time T = t4 shown in FIG. 3). Is calculated at time T = t6) after 400 [μs], and the pulse center is 600 [μs] after time T = t4. For this reason, the θ, ω calculation unit 4 uses an angle after 600 [μs] of the rotation angle θ of the motor 2 detected this time as an estimated rotation angle θ ′ at the current calculation timing.

  On the other hand, when switching the carrier frequency from 10 [kHz] to 2.5 [kHz], the control unit 9 sends a carrier switching mode signal to the θ and ω calculation unit 4 before switching the carrier frequency (time T = t1). By inputting (I), a value (300 [μs]) obtained by adding the half width (200 [μs]) of the pulse after switching the carrier frequency to the current pulse width (100 [μs]) is θ-corrected. The θ and ω calculation unit 4 is controlled so as to calculate the time (I) and use the angle after the θ correction time (I) (time T = t3) as the estimated rotation angle θ ′ at the current calculation timing.

  Next, in response to the carrier frequency switching from 10 [kHz] to 2.5 [kHz] (time T = t2), the controller 9 sends a carrier switching mode signal (II) to the θ, ω calculator 4. By inputting, a value (600 [μs]) obtained by adding the half width (200 [μs]) of the pulse after switching the carrier frequency to the current pulse width (400 [μs]) is θ correction time (II) The θ and ω calculation unit 4 is controlled so that the angle after θ correction time (II) (time T = t5) is used as the estimated rotation angle θ ′ at the current calculation timing.

  Finally, after the carrier frequency is switched to 2.5 [kHz] (time T = t4), the control unit 9 inputs the carrier switching mode signal (III) to the θ, ω calculation unit 4 to thereby A value (600 [μs]) obtained by adding the half width (200 [μs]) of the pulse after switching the carrier frequency to the pulse width (400 [μs]) is calculated as θ correction time (III), and θ correction time (III) The θ, ω calculation unit 4 is controlled so that the angle after (time T = t7) is used as the estimated rotation angle θ ′ at the current calculation timing.

  Similarly, when the carrier frequency is switched from 2.5 [kHz] to 10 [kHz], the control unit 9 similarly switches the carrier frequency from 2.5 [kHz] to 10 [kHz] (time T = t6). ) By inputting the carrier switching mode signal (I) to the θ, ω calculation unit 4, and the half width (50 [μs]) of the pulse after switching the carrier frequency to the current pulse width (400 [μs]). (450 [μs]) is added as the θ correction time (I), and the angle after the θ correction time (I) (time T = t9) is used as the estimated rotation angle θ ′ at the current calculation timing. The θ, ω calculation unit 4 is controlled.

  Next, in response to the carrier frequency switching from 2.5 [kHz] to 10 [kHz] (time T = t8), the controller 9 sends the carrier switching mode signal (II) to the θ, ω calculator 4. By inputting, a value (150 [μs]) obtained by adding the half width (50 [μs]) of the pulse after switching the carrier frequency to the current pulse width (100 [μs]) is θ correction time (II) The θ and ω calculation unit 4 is controlled so that the angle after θ correction time (II) (time T = t11) is used as the estimated rotation angle θ ′ at the current calculation timing.

  Finally, after the carrier frequency is switched to 10 [kHz] (time T = t10), the control unit 9 inputs the carrier switching mode signal (III) to the θ, ω calculation unit 4 to thereby obtain the current pulse width. A value (150 [μs]) obtained by adding the half width (50 [μs]) of the pulse after switching the carrier frequency to (100 [μs]) is calculated as θ correction time (III), and θ correction time (III ) The θ, ω calculation unit 4 is controlled so that the angle after (time T = t13) is used as the estimated rotation angle θ ′ at the current calculation timing.

  As is apparent from the above description, according to the motor control device 1 according to the embodiment of the present invention, when the carrier frequency is switched, the control unit 9 changes the proportional gain and the integral gain. At this time, it is possible to prevent the current behavior of the motor 2 from becoming unstable.

  Further, according to the motor control device 1 according to the embodiment of the present invention, the control unit 9 switches the calculation method of the estimated rotation angle θ ′ of the motor 2 step by step according to the carrier frequency before and after switching the carrier frequency. Therefore, when the rotation angle of the motor 2 is different from the actual rotation angle, it is possible to prevent a deviation in the switching timing of the power element, thereby causing a current waveform disturbance or a torque shock.

  As mentioned above, although the embodiment to which the invention made by the present inventors was applied has been described, the present invention is not limited by the description and the drawings that form part of the disclosure of the present invention according to this embodiment. That is, it should be added that other embodiments, examples, operation techniques, and the like made by those skilled in the art based on the above embodiments are all included in the scope of the present invention.

It is a block diagram which shows the structure of the motor control apparatus used as embodiment of this invention. It is a block diagram which shows the internal structure of the electric current PI control part shown in FIG. It is a timing chart for demonstrating operation | movement of the control part at the time of reducing and increasing a carrier frequency.

Explanation of symbols

1: Motor control device 2: Motor 3: Inverter 4: θ, ω calculation unit 5: 3 phase → 2 phase conversion unit 6: Current PI control unit 7: 2 phase → 3 phase conversion unit 8: Duty (DUTY) command unit 9: Control unit 11: Difference calculation unit 12: Proportional gain adjustment unit 13: Proportional term calculation unit 14: Integral gain adjustment unit 15: Integral term calculation unit

Claims (4)

By switching on / off of a plurality of power elements constituting the inverter by pulse width modulation processing, AC current is supplied to the three-phase AC motor, and feedback control is performed so that the current value supplied to the motor follows the current command value A motor control device,
When switching the carrier frequency, comprising a control unit that changes the feedback gain used in the feedback control according to the carrier frequency,
The feedback gain has a proportional gain and an integral gain,
The lower the carrier frequency, the lower the proportional gain,
When the carrier frequency is Ru is switched from high frequency to low frequency, the integral gain is determined by calculation cycle and the frequency crossover, further integral gain when the integral gain and a high frequency when the carrier frequency is low frequency The size is decided ,
If the integral gain at low frequency is larger than the integral gain at high frequency, switch the carrier frequency from high frequency to low frequency and then change the integral gain from high frequency integral gain to low frequency integral gain. A motor control device. The motor control device according to claim 1,
The proportional gain is represented by the product of the coil inductance and the crossing angular frequency,
The motor control apparatus according to claim 1, wherein the integral gain is represented by a product of winding resistance, crossing angular frequency, and calculation cycle. The motor control device according to claim 1 or 2,
The controller is configured to switch the calculation method of the rotation angle of the three-phase AC motor stepwise in accordance with the carrier frequency before and after switching before and after switching the carrier frequency. By switching on / off of a plurality of power elements constituting the inverter by pulse width modulation processing, AC current is supplied to the three-phase AC motor, and feedback control is performed so that the current value supplied to the motor follows the current command value A control method for a motor control device, comprising:
When switching the carrier frequency, the step of changing the feedback gain used in the feedback control according to the carrier frequency,
The feedback gain has a proportional gain and an integral gain,
The lower the carrier frequency, the lower the proportional gain,
When the carrier frequency is Ru is switched from high frequency to low frequency,
Obtaining the integral gain by a cross angular frequency and a calculation period ;
Determining the integral gain when the carrier frequency is low and the integral gain when the carrier frequency is high;
When the integral gain at low frequency is larger than the integral gain at high frequency, the step of switching the carrier frequency from high frequency to low frequency, the step of changing the integral gain from the integral gain at high frequency to the integral gain at low frequency, the method of the motor control device characterized by having a.

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