JP4697016B2 - Control device for multiphase rotating electrical machine - Google Patents

Description

  The present invention operates the switching element of the inverter based on the magnitudes of the upper and lower limits of a predetermined hysteresis region determined by the command current for the multiphase rotating electrical machine and the actual current flowing through the multiphase rotating electrical machine, thereby operating the actual current. Is controlled to a required current value for generating the required torque.

  In general, an inverter that controls the output torque of a motor by causing a sinusoidal current to flow in each phase of a three-phase motor performs PWM control using a triangular wave-shaped carrier wave. However, since this method has a low voltage utilization rate defined by the effective value of the interphase voltage of the motor with respect to the input voltage of the inverter, it is difficult to achieve a high rotational speed and a large output torque.

  On the other hand, as a control method for theoretically maximizing the voltage utilization factor, so-called one-pulse control is known in which the switching element of the inverter is operated with a drive pulse that matches the half cycle of the required current for generating the required torque. ing. According to this one-pulse control, since the voltage utilization rate can be maximized, it is possible to realize control with high rotational speed and large output torque.

  However, since 1-pulse control is open-loop control, the current flowing through the motor may exceed the maximum rated current and become excessively large. Furthermore, since the voltage utilization factor is fixed, an increase in power loss becomes a problem in the low rotation speed / low output torque region.

  Therefore, conventionally, as can be seen in Patent Document 1 below, for example, a so-called instantaneous current value for operating the switching element of the inverter based on the upper and lower limits of the hysteresis region determined by the command current for the motor and the actual current flowing through the motor. Control has also been proposed. Since the instantaneous current value control can continuously increase the voltage utilization rate up to a voltage utilization rate higher than that of the PWM control, it is possible to appropriately perform a control with a higher rotation speed and a larger output torque than the PWM control. .

By the way, when the rotational speed is increased by the instantaneous current value control, it is necessary to use the voltage utilization rate by the one-pulse control in order to generate the required torque. However, in the instantaneous current value control, since the switching element is operated so as to follow the sinusoidal command current, there is a tendency that a plurality of switching operations are performed during one cycle of the required current. The utilization rate decreases. And when a voltage utilization factor falls, there exists a possibility that a required torque cannot be produced | generated.
Japanese Patent Laid-Open No. 10-174453

  The present invention has been made in order to solve the above-described problems. The purpose of the present invention is to improve the controllability of a multi-phase rotating electrical machine by instantaneous current value control even in a region where the rotational speed of the multi-phase rotating electrical machine is high. An object of the present invention is to provide a control device for a multi-phase rotating electrical machine that can be maintained.

  Hereinafter, means for solving the above-described problems and the operation and effects thereof will be described.

According to a fourth aspect of the present invention, the switching element other than the vicinity of the inversion timing of the driving pulse assumed when the required torque is realized by one-pulse control using a driving pulse that matches a half cycle of the required current value. It is characterized by comprising limiting means for limiting the operation of the switching element by the feedback control by fixing the operation state.
  According to a fifth aspect of the present invention, in the invention according to the fourth aspect, the limiting means is a sine wave-shaped phase assumed when the required current is supplied to the multi-phase rotating electrical machine in a region other than the region where the limiting is performed. When the voltage decreases, only switching that allows the negative electrode potential side of the inverter and the multiphase rotating electrical machine side to conduct is allowed. When the phase voltage increases, the positive potential side of the inverter and the multiphase rotating electrical machine side are connected. Only switching for conducting is allowed.
  According to a sixth aspect of the present invention, the feedback control is performed except in the vicinity of the inversion timing of the drive pulse assumed when the required torque is realized by one-pulse control using a drive pulse that matches a half cycle of the required current value. In order to limit the operation of the switching element according to the above, there is provided limiting means for increasing the absolute value of the command current more than the required current value.

  In the above-described configuration, since the limit is made except in the vicinity of the inversion timing of the drive pulse by the one-pulse control, switching of the switching operation state is suppressed outside the vicinity. For this reason, even in a high rotation speed and large torque region, control that approximates to one-pulse control that provides the theoretically maximum voltage utilization rate can be performed by feedback control of current.

The invention according to an eighth aspect is the invention according to any one of the first to seventh aspects, wherein the limiting means has a sinusoidal shape assumed when a current of the required current value is passed through the multiphase rotating electrical machine. The limitation is performed when a value twice the amplitude of the phase voltage exceeds the input voltage of the inverter.

  In the above configuration, when the value of twice the amplitude of the sinusoidal phase voltage assumed when the required current is passed exceeds the input voltage of the inverter, switching was performed so as to follow the sinusoidal command current. However, the followability of the actual current to the command current is reduced. At this time, an unnecessary switching operation is performed, which may reduce the voltage utilization rate. In this regard, in the above-described configuration, unnecessary switching operation can be suppressed by performing the above limitation when the value of twice the amplitude of the phase voltage exceeds the input voltage of the inverter, and thus the voltage utilization rate is reduced. Can be suppressed.

According to the first aspect of the present invention, a phase in which a sine wave-shaped phase voltage assumed when the required current is supplied to the multiphase rotating electrical machine exceeds the positive electrode potential of the inverter, The phase rotating electric machine side is limited to be in a conductive state, and the phase of the sinusoidal phase voltage assumed when the required current is supplied to the multiphase rotating electric machine is less than the negative potential of the inverter. There is provided a limiting means for restricting the positive electrode potential side and the multiphase rotating electrical machine side from being brought into conduction.

  In the above configuration, when the phase voltage exceeds the positive electrode potential of the inverter, it is not possible to generate a sinusoidal voltage for following the sinusoidal command current. For this reason, if normal instantaneous current value control is performed at this time, an unnecessary switching operation may be performed, which may reduce the voltage utilization factor. Under such circumstances, in the above configuration, it is possible to limit the unnecessary switching operation by limiting the negative electrode potential side of the inverter and the multiphase rotating electrical machine side from being in a conductive state, and thus the voltage utilization rate. Can be suppressed.

  In the above configuration, when the phase voltage is lower than the negative potential of the inverter, it is not possible to generate a sinusoidal voltage for following the sinusoidal command current. For this reason, if normal instantaneous current value control is performed at this time, an unnecessary switching operation may be performed, which may reduce the voltage utilization factor. Under such circumstances, in the above configuration, by restricting the positive electrode potential side and the multiphase rotating electrical machine side of the inverter from being in a conductive state, it is possible to limit the unnecessary switching operation, and thus the voltage utilization rate. Can be suppressed.

According to a second aspect of the present invention, the phase voltage is positive in a range other than a predetermined phase range centered on a zero-cross point of a sinusoidal phase voltage assumed when the required current is supplied to the multiphase rotating electrical machine. The phase of the negative electrode side of the inverter and the side of the multi-phase rotating electrical machine are limited to be in a conductive state, and the phase of the phase voltage of the negative phase side of the inverter and the multi-phase rotating electrical machine of the inverter It is characterized by comprising limiting means for limiting the connection with the side.

  In the above-described configuration, a point where the sine wave-shaped phase voltage is at its maximum and minimum is included outside the predetermined phase range centered on the zero-cross point of the phase voltage, and the actual current is added to the sine-wave shaped command current. If a switching operation is performed to follow, this may become an inappropriate switching operation. Under such circumstances, in the above configuration, unnecessary switching operation is performed by restricting that the negative electrode potential side of the inverter and the multiphase rotating electrical machine side are in a conductive state for the phase in which the phase voltage is positive. It is possible to suppress a decrease in voltage utilization rate due to. In addition, with respect to the phase in which the phase voltage is negative, by limiting the positive electrode potential side of the inverter and the multiphase rotating electrical machine side from being in a conductive state, the voltage utilization rate is reduced due to unnecessary switching operations. Can be suppressed.

According to a third aspect of the present invention, in the second aspect of the invention, the multi-phase rotating electrical machine is a three-phase rotating electrical machine, and the predetermined phase range is “± 30 °” or more.

  In the above configuration, by setting the predetermined phase range to “± 30 °” or more, at least one of the three phases can always be in a state where the switching operation is not restricted. For this reason, in spite of the limitation, it is possible to appropriately cope with this change by feedback control when the operating state of the multiphase rotating electrical machine changes suddenly.

The invention according to claim 9 is the invention according to any one of claims 1 to 8 , wherein the limiting means has a sinusoidal shape for causing the command current to flow to the multiphase rotating electrical machine based on the command current. A means for calculating a phase voltage is provided, and the restriction is performed based on the calculated phase voltage.

  In the above configuration, in order to calculate the phase voltage based on the command current, whether or not the control for causing the actual current flowing through the multiphase rotating electrical machine to follow the command current can be appropriately performed by the instantaneous current value control. Judgment can be made. On the other hand, for example, when calculating the phase voltage based on the actual current of the multiphase rotating electrical machine, the phase voltage becomes inappropriate depending on the degree to which the actual current is separated from the required current of the sine wave shape. As a result, the reliability of the above judgment is lowered.

According to a seventh aspect of the invention, in the sixth aspect of the invention, the limiting means sets the absolute value of the command current to be equal to or less than the maximum rated current of the inverter when performing the limiting.

  In the above configuration, when a current exceeding the maximum rated current flows through the inverter, the switching element is operated by feedback control regardless of whether or not the current is limited. Can be suppressed.

  Hereinafter, an embodiment in which a control device for a multi-phase rotating electrical machine according to the present invention is applied to a control device for a three-phase rotating electrical machine mounted on a hybrid vehicle will be described with reference to the drawings.

  FIG. 1 shows the overall configuration of the three-phase rotating electrical machine and its control device.

  As illustrated, an inverter 10 is connected to three phases (U phase, V phase, and W phase) of a DC brushless motor (motor 2) that is a three-phase rotating electrical machine. The inverter 10 is a three-phase inverter and includes a parallel connection body of switching elements 12 and 14, switching elements 16 and 18, and switching elements 20 and 22 corresponding to the three phases. Furthermore, the inverter 10 includes diodes 24 to 34 connected in parallel to the switching elements 12 to 22. A connection point where the switching element 12 and the switching element 14 are connected in series is connected to the U phase of the motor 2. A connection point for connecting the switching element 16 and the switching element 18 in series is connected to the V phase of the motor 2. Furthermore, the connection point for connecting the switching element 20 and the switching element 22 in series is connected to the W phase of the motor 2. Incidentally, these switching elements 12-22 are comprised by the insulated gate bipolar transistor (IGBT) in this embodiment.

  The voltage of the power supply 42 is connected to both ends (positive potential side and negative potential side) of each pair of switching elements 12 and 14, switching elements 16 and 18, and switching elements 20 and 22 of the inverter 10 via a smoothing capacitor 40. Applied.

  On the other hand, the microcomputer 50 includes a position sensor 52 that detects the rotation angle of the output shaft of the motor 2, current sensors 54 and 56 that detect current flowing in the U phase and the V phase, and a voltage sensor 58 that detects the voltage of the power source 42. Capture the detection results. The microcomputer 50 calculates the current flowing through the W phase from the current flowing through the U phase and the current flowing through the V phase based on Kirchhoff's law. The microcomputer 50 operates the switching elements 12 to 22 via the gate drive circuits 60 to 70 based on the rotation angle of the output shaft of the motor 2 and the currents flowing through the three phases.

  FIG. 2 shows a functional block diagram of processing related to generation of drive pulses for operating the switching elements 12 to 22 among processing performed by the microcomputer 50.

  In the present embodiment, the switching elements 12 to 22 are operated by instantaneous current value control. In the following, first, among the processes shown in FIG. 2, a process related to normal instantaneous current value control will be described.

  The command current generation unit 80 determines the command currents iqc, idc according to the required torque of the motor 2 calculated by another logic (not shown), the rotation speed as a time differential value of the rotation angle detected by the position sensor 52, and the like. Is the part that generates The command currents iqc and idc are command currents on the dq axis.

  The command current conversion unit 82 is a part that converts the command currents iqc, idc on the dq axis into three-phase command currents iuc, ivc, iwc. These command currents iuc, ivc, and iwc are taken into hysteresis comparators 84 to 88, respectively. In the hysteresis comparators 84 to 88, the current actually flowing through each phase of the motor 2 (actual currents iu, iv, iw) is smaller when the upper limit of a predetermined hysteresis region determined by the command currents iuc, ivc, iwc is larger. Output signals gu, gv, and gw whose values are inverted at the same time.

  The output signals gu, gv, and gw are taken into the drive pulse limiting unit 90. The drive pulse limiting unit 90 directly outputs output signals gu, gv, and gw when the motor 2 is operated at a low rotational speed and a small output torque. Then, these output signals gu, gv, gw and their inverted signals by the inverters 102, 104, 106 are taken into the deadtime generator 108. The Deadtime generation unit 108 shapes the waveform of the selected signals and the inverted signal corresponding to the selected signals so as to avoid the timing overlap between these edge portions. The waveform-shaped signal includes a drive pulse gup for operating the U-phase switching element 12, a drive pulse gun for operating the U-phase switching element 14, a drive pulse gvp for operating the V-phase switching element 16, and a V-phase. Drive pulse gvn for operating the switching element 18, drive pulse gwp for operating the W-phase switching element 20, and drive pulse gwn for operating the W-phase switching element 22.

  According to such a configuration, based on the magnitudes of the actual currents iu, iv, iw and the command currents iuc, ivc, iwc (more precisely, based on the magnitudes of the upper limit and the lower limit of the hysteresis region), the switching elements 12 to 22 operations are performed. FIG. 3 shows an aspect of instantaneous current value control in which the switching elements 12 to 22 are operated according to such an aspect. In FIG. 3, for the U phase, the actual current iu is indicated by a solid line, and the command current iuc is indicated by a two-dot chain line. As shown in the figure, a region between a value larger by “½” of the hysteresis width hys than the command current iuc and a value smaller by “½” of the hysteresis width hys than the command current iuc (hysteresis region). The actual current iu is controlled so as to fall within the range. Thereby, the current flowing through each phase of the motor 2 can be controlled to the required current for generating the required torque. That is, the actual currents iu, iv, and iw can be set as the required currents by performing the instantaneous current value control while setting the command currents iuc, ivc, and iwc as the required currents.

  By the way, in the high rotation speed and large required torque region of the motor 2, the voltage utilization rate necessary for flowing the required current to the motor 2 increases. The maximum value of the voltage utilization rate can theoretically be realized by the one-pulse control shown in FIG. A two-dot chain line in FIG. 4A indicates a U-phase command current iuc, and a solid line indicates a U-phase actual current iu. FIG. 4B shows a driving pulse by one-pulse control that generates the actual current iu. As shown in the figure, in the one-pulse control, a drive pulse (a logic “H” state and a logic “L” state within one cycle of the request current are changed in the pulse width and the half cycle of the request current (command current iu)). The switching elements 12 to 22 are operated by a driving pulse that is set once.

  In the high rotation speed and large required torque region of the motor 2, it is desired that the instantaneous current value control be controlled by a drive pulse similar to the drive pulse shown in FIG. However, in the instantaneous current value control, control is performed so as to follow the required current (command current iuc) having a sine wave shape, so that the drive pulses for operating the switching elements 12 to 22 are as shown in FIG. It will be different. FIG. 4C shows an image of a drive pulse when the command current iuc shown in FIG. 4A is followed by instantaneous current value control. Thus, the drive pulse by the instantaneous current value control is different from that shown in FIG.

  Therefore, in the present embodiment, processing for limiting the operation of the switching elements 12 to 22 by instantaneous current value control is performed under a predetermined condition. When the predetermined condition is satisfied, twice the amplitude of the sinusoidal phase voltage (applied voltages vu, vv, vw of each phase) assumed when the required current is supplied to the motor 2 is the input voltage of the inverter 10 ( It is assumed that the voltage Ve) of the power source 42 is exceeded. Hereinafter, of the processes shown in FIG. 2 described above, a process related to restriction will be described.

  The current-voltage conversion unit 110 is a part that converts the command currents idc, iqc into command voltages vdc, vqc. On the other hand, the command voltage conversion unit 112 is a part that converts the command voltages vdc, vqc into three-phase command voltages vuc, vvc, vwc. The switching restriction unit 114 is a part that operates the drive pulse restriction unit 90 to perform restriction based on the command voltages vuc, vvc, vwc and the power supply voltage Ve. FIG. 5 shows a mode of operation restriction of the switching elements 12 to 22 by the switching restriction unit 114 and the drive pulse restriction unit 90.

  FIG. 5A shows a case where the fluctuation range of the U-phase command voltage vuc is within the input voltage of the inverter 10 (the voltage Ve of the power supply 42). At this time, normal instantaneous current value control is performed.

  On the other hand, FIG. 5B and FIG. 5C show when the fluctuation range of the U-phase command voltage vuc protrudes from the input voltage of the inverter 10 (the voltage Ve of the power source 42).

  In this case, when the command voltage vuc is larger than “½” of the power supply voltage Ve, the switching element 12 is fixed in the on state and the switching element 14 is fixed in the off state. In other words, the positive electrode potential side of the inverter 10 and the U phase of the motor 2 are fixed in a conductive state. Here, when the command voltage vuc is larger than “½” of the power supply voltage Ve, the sine wave-shaped command voltage vuc cannot be generated, and appropriate feedback control cannot be performed depending on the instantaneous current value control. . In this region, according to one-pulse control, the positive potential side of the inverter 10 and the U phase of the motor 2 are usually fixed in a conductive state. Therefore, in the present embodiment, in this region, the switching elements 12 to 22 are fixed in the same manner as in the one-pulse control.

  When the command voltage vuc is smaller than “−½” of the power supply voltage Ve, the switching element 12 is fixed in the off state and the switching element 14 is fixed in the on state. In other words, the negative electrode potential side of the inverter 10 and the U phase of the motor 2 are fixed in a conductive state. Here, when the command voltage vuc is smaller than “−½” of the power supply voltage Ve, the sine wave-shaped command voltage vuc cannot be generated, and appropriate feedback control may be performed depending on the instantaneous current value control. Can not. In this region, according to one-pulse control, the negative electrode potential side of the inverter 10 and the U phase of the motor 2 are usually fixed in a conductive state. Therefore, in the present embodiment, in this region, the switching elements 12 to 22 are fixed in the same manner as in the one-pulse control.

  In order to fix the switching elements 12 to 22 in the above manner, the drive pulse limiting unit 90 is as shown in FIG. That is, in the drive pulse limiting unit 90, logical sum signals of the outputs of the hysteresis comparators 84 to 88 and the H fixing signals gu1, gv1, and gw1 are generated by the logical sum generation units 92u, 92v, and 92w. Further, the inverted signal generation units 94u, 94v, and 94w generate a logically inverted signal of the L fixing signal gu0. Further, logical product signals of the outputs of the logical sum generation units 92u, 92v, 92w and the outputs of the inverted signal generation units 94u, 94v, 94w are generated by the logical product generation units 96u, 96v, 96w.

  FIG. 7 shows a processing procedure relating to the generation of the H fixing signal gu1 and the L fixing signal gu0 by the switching restriction unit 114. Note that the processes related to the generation of the H fixing signals gv1, gw1 and the L fixing signals gv0, gw0 are the same as the processes shown in FIG. 7, and therefore the description thereof is omitted here.

  As shown in the figure, when the command voltage vuc is not more than “½” of the power supply voltage Ve (step S10: NO) and is not less than “−1/2” of the power supply voltage Ve (step S16: NO). The H fixing signal gu1 and the L fixing signal gu0 are set to logic “L” (steps S14 and S20).

  On the other hand, when the command voltage vuc is higher than “½” of the power supply voltage Ve, the H fixing signal gu1 is set to logic “H” (step S12). When the command voltage vuc is lower than “−½” of the power supply voltage Ve, the L fixing signal gu0 is set to logic “H” (step S18).

  This makes it possible to limit the operation of the switching elements 12 and 14 in the manner shown in FIG.

  According to the embodiment described in detail above, the following effects can be obtained.

  (1) The switching limiting unit 114 and the drive pulse limiting unit 90 are provided to limit the operation of the switching elements 12 to 22 by feedback control. Thereby, the fall of the voltage utilization factor by switching operation more than necessary can be suppressed suitably, and the controllability of the motor 2 can be maintained high by extension.

  (2) Limitation was performed except in the vicinity of the inversion timing of the drive pulse assumed when the required torque was realized by one-pulse control. As a result, even in the high rotation speed and large torque region, control that approximates the one-pulse control that provides the theoretically maximum voltage utilization rate can be performed by current feedback control.

  (3) Limitation was performed when a value twice the amplitude of the sinusoidal phase voltage assumed when the current of the required current value flows through the motor 2 exceeds the input voltage of the inverter 10. As a result, the switching is performed so as to follow the sinusoidal command currents iuc, ivc, iwc, so that the tracking of the actual currents iu, iv, iw to the command currents iuc, ivc, iwc is reduced. It can be performed.

  (4) For the phase in which the phase voltage (potential) of the sine wave assumed when the required current is supplied to the motor 2 exceeds the positive potential of the inverter 10, the negative potential side of the inverter 10 and the motor 2 side are in a conductive state. For the phase in which the phase voltage (potential) of the sine wave assumed when the required current flows through the motor 2 is lower than the negative potential of the inverter 10, the positive potential side of the inverter 10 and the motor 2 side are Restricted to be in a conductive state. Thereby, it can restrict | limit that an unnecessary switching operation is made, and can suppress the fall of a voltage utilization factor by extension.

  (5) Based on the command currents idc and iqc, a sinusoidal phase voltage for flowing the command currents iuc, ivc and iwc to the motor 2 was calculated, and the limitation was performed based on this. As a result, it is possible to appropriately determine whether or not the control for causing the actual currents iu, iv, iw flowing through the motor 2 to follow the command currents iuc, ivc, iwc can be appropriately performed by the instantaneous current value control.

  (6) Restriction was performed by fixing the operating state of the switching elements 12-22. Thereby, unnecessary operation of the switching elements 12 to 22 can be prohibited.

(Second Embodiment)
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 8 shows a mode of operation restriction of the switching elements 12 to 22 according to the present embodiment. FIG. 8 corresponds to FIG.

  As shown in the figure, also in this embodiment, switching is limited when the fluctuation range of the command voltage vuc exceeds the voltage range of the power supply 42. However, in the present embodiment, switching is limited except within a predetermined phase φ centered on the zero-cross point of the command voltage vuc. Specifically, when it is out of the above range and the command voltage vuc is positive, the switching element 12 is fixed in the on state and the switching element 14 is fixed in the off state. In other words, the positive electrode potential side of the inverter 10 and the motor 2 side are fixed in a conductive state. On the other hand, when it is out of the range and the command voltage vuc is negative, the switching element 12 is fixed in the OFF state and the switching element 14 is fixed in the ON state. In other words, the negative electrode potential side of the inverter 10 and the motor 2 side are fixed in a conductive state.

  FIG. 9 shows a functional block diagram of processing related to generation of drive pulses for operating the switching elements 12 to 22 among the processing performed by the microcomputer 50. In FIG. 9, blocks having the same functions as those of the block shown in FIG.

As illustrated, the present embodiment includes a voltage phase calculation unit 116. The voltage phase calculation unit 116 calculates phases θvu, θvv, θvw of the three-phase command voltages vuc, vvc, vwc based on the command voltages vdc, vqc. This becomes the following formulas (c1) to (c3).
θvu = θvdq + π / 2 (c1)
θvv = θvdq + π / 2− (2/3) π (c2)
θvw = θvdq + π / 2 + (2/3) π (c3)
However, the phase θvdq is expressed by the following equation (c4) using the initial value θre.
θvdq = arctan (vqc / vdc) + θre (c4)
On the other hand, the switching restriction unit 114 restricts the switching operation based on the phases θvu, θvv, and θvw. However, since this restriction is performed only when the command voltages vuc, vvc, and vwc exceed the power supply voltage Ve, the switching restriction unit 114 captures the command voltages vuc, vvc, and vwc also in this embodiment. FIG. 10 shows a procedure of processing related to generation of the H fixing signal gu1 and the L fixing signal gu0 by the switching restriction unit 114.

  According to this series of processing, when the phase θvu is equal to or smaller than the phase φ (step S30: NO), when larger than the phase “π−φ” and equal to or smaller than “π + φ” (step S38: NO), or when the phase “ When larger than 2π−φ ”(step S40: NO), the H fixing signal gu1 is set to logic“ L ”(step S36) and the L fixing signal gu0 is set to logic“ L ”(step S36). S44). On the other hand, when the phase θvu is larger than the phase φ and smaller than the phase “π−φ” (step S32: YES), the H fixing signal gu1 is set to the logic “H” (step S34). When the phase θvu is larger than the phase “π + φ” and smaller than the phase “2π−φ” (step S40: YES), the L fixing signal gu0 is fixed to the logic “H” (step S42).

  The phase φ is preferably “30 °” or more. As a result, at least one of the three phases can always be in a state where the switching operation is not restricted. For this reason, this change can be appropriately dealt with by feedback control when the operating state of the motor 2 is suddenly changed despite the limitation. In view of the fact that the operation mode of the switching elements 12 to 22 can be approximated to the operation mode of the switching elements 12 to 22 by one-pulse control as the unrestricted region is narrowed, it is desirable to make the phase φ as small as possible. Therefore, from the above two viewpoints, it is particularly desirable that the phase φ is “30 °” in order to make it possible to always perform the feedback control and approximate the one-pulse control as much as possible.

  According to the present embodiment described above, in addition to the effects (1) to (3), (5), and (6) of the first embodiment, the following effects can be obtained. .

  (7) For the phases in which the command voltages vuc, vvc, vwc are positive outside the range of the predetermined phase φ centered on the zero cross point of the command voltages vuc, vvc, vwc, the negative potential side of the inverter 10 and the motor 2 side And the positive voltage side of the inverter 10 and the motor 2 side are restricted to be in a conductive state for the phase in which the command voltages vuc, vvc, and vwc are negative. Thereby, the fall of the voltage utilization factor by performing unnecessary switching operation can be suppressed.

  (8) The predetermined phase φ is set to “30 °” or more. Thereby, even when the operating state of the motor 2 is suddenly changed, this change can be appropriately dealt with by feedback control.

(Third embodiment)
Hereinafter, the third embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  In this embodiment, as shown in FIG. 11, the operation of the switching elements 12 to 22 is limited by increasing the absolute value of the command current iuc. That is, as shown in the figure, when the command voltage vuc is higher than “½” of the power supply voltage Ve, the command current iuc is set to the upper limit value Imax of the output current. Thereby, it is suppressed that the actual current iu exceeds the command current iuc, and the negative electrode potential side of the inverter 10 and the motor 2 side are restricted from being brought into conduction. Further, when the command voltage vuc is lower than “−½” of the power supply voltage Ve, the command current iuc is set to the lower limit value Imin of the output current. As a result, the actual current iu is suppressed from falling below the command current iuc, and the positive electrode potential side of the inverter 10 and the motor 2 side are restricted from being brought into conduction.

  FIG. 12 shows a configuration of the drive pulse limiting unit 90 according to the present embodiment. In FIG. 12, blocks having the same functions as those shown in FIG. 6 are given the same reference numerals for the sake of convenience.

  As shown in the figure, in the present embodiment, the drive pulse limiter 90 outputs a signal to the hysteresis comparators 84, 86, 88. The drive pulse limiting unit 90 includes command current switching units 98u, 98v, and 98w corresponding to the respective phases. The command current switching units 98u, 98v, and 98w take in the output of the command current conversion unit 82, and take in the H fixing signal gu1 and the L fixing signal gu0, and based on this, the command currents iuc, ivc, iwc Is generated.

  FIG. 13 shows a procedure of processing performed particularly by the command current switching unit 98u among the command current switching units 98u, 98v, 98w.

  As shown in the figure, when both the H fixing signal gu1 and the L fixing signal gu0 are logic “L” (step S54: NO), the command current iuc output from the command current conversion unit 82 of FIG. Is used as it is (step S58). On the other hand, when the H fixing signal gu1 is logic “H” (step S50: YES), the command current iuc is set to the upper limit value Imax (step S52). When the L fixing signal gu0 is logic “H” (step S54: YES), the command current iuc is set to the lower limit value Imin (= −Imax) (step S56).

  Note that the upper limit value Imax is equal to or less than the maximum rated current IM of the inverter 10. Thereby, when the absolute value of the current flowing through the inverter 10 exceeds the maximum rated current, feedback control can be performed so as to decrease the absolute value.

  According to this embodiment described above, in addition to the effects (1) to (5) of the first embodiment, the following effects can be obtained.

  (9) The switching operation is limited by increasing the absolute values of the command currents iuc, ivc, iwc from the required current value. Thereby, unnecessary operation of the switching element can be suitably suppressed.

  (10) When limiting the switching operation, the absolute values of the command currents iuc, ivc, iwc are set to be equal to or less than the maximum rated current of the inverter 10. As a result, when a current higher than the maximum rated current flows through the inverter 10, the switching element is operated by feedback control regardless of the limitation of the switching operation, so that the current exceeding the maximum rated current is prevented from flowing through the inverter 10. be able to.

(Fourth embodiment)
Hereinafter, the fourth embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  In the present embodiment, in a region where the operation state of the switching elements 12 to 22 is not fixed, when the command voltages vuc, vvc, and vwc are reduced, only switching that makes the negative potential side of the inverter 10 and the motor 2 side conductive is permitted. When the command voltages vuc, vvc, and vwc rise, only switching that allows the positive electrode potential side of the inverter 10 and the motor 2 side to conduct is allowed. As a result, switching is allowed only once in a region other than the region where the restriction is performed.

  FIG. 14 shows a configuration of the drive pulse limiting unit 90 according to the present embodiment. In FIG. 14, blocks having the same functions as those of the block shown in FIG.

  As shown in the figure, the drive pulse limiting unit 90 of the present embodiment is the same as that of the first embodiment (FIG. 6). However, in the present embodiment, the H fixing signals Gu1, Gv1, Gw1 are input to the logical sum generation units 92u, 92v, 92w instead of the H fixing signals gu1, gv1, gw1. Further, the L fixing signals Gu0, Gv0, Gw0 are input to the inverted signal generating units 94u, 94v, 94w instead of the L fixing signals gu0, gv0, gw0. The H fixing signals Gu1, Gv1, Gw1 and the L fixing signals Gu0, Gv0, Gw0 are switched based on the H fixing signals gu1, gv1, gw1 and the L fixing signals gu0, gv0, gw0. It is generated by the restriction unit 114 by the process shown in FIG.

  In the series of processing shown in FIG. 15, when the command voltage vuc exceeds “½” times the power supply voltage Ve (see FIG. 5), the H fixing signal gu1 is logic “H”, so step S60. In step S62, an affirmative determination is made. In step S62, the H fixing history signal gu1old is set to logic “H”, the H fixing signal Gu1 is set to logic “H”, and the L fixing signal Gu0 is set to logic “L”. Therefore, when the H fixing signal gu1 is set to logic “H”, the switching element 12 is fixed in the on state and the switching element 14 is fixed in the off state, as in the first embodiment.

  When the command voltage vuc falls within “½” of the power supply voltage Ve, the H fixing signal gu1 is inverted to logic “L”. For this reason, a negative determination is made in step S60. At this time, since the L fixing signal gu0 is also logic “L”, a negative determination is also made in step S64. When it is determined in step S68 that the output signal gu of the hysteresis comparator 84 is logic “H”, the process proceeds to step S70. However, at this time, since the L fixing history signal gu0old is not the logic “L”, a negative determination is made in step S70, and this series of processes is temporarily ended. On the other hand, when the output signal gu is inverted to the logic “L”, a negative determination is made in step S68, and the process proceeds to step S74. Here, since the H fixing history signal gu1old becomes logic “H”, the process proceeds to step S76. In step S76, the H fixing history signal gu1old is set to logic “L”, and the L fixing signal Gu0 is set to logic “H”. Thereby, the switching element 12 is switched to an OFF state, and the switching element 14 is switched to an ON state.

  If this process is entered again in this state, a negative determination will be made in both S70 and S74 while both the H fixing signal gu1 and the L fixing signal gu0 are logic "L". The switching state is not switched as described above.

  Thereafter, when the command voltage vuc falls below “−½” times the power supply voltage Ve, the L fixing signal gu0 is inverted to the logic “H”, so that an affirmative determination is made in step S64. Accordingly, in step S66, the L fixing history signal gu0old is set to logic “H”, the L fixing signal Gu0 is set to logic “H”, and the H fixing signal Gu1 is set to logic “L”. Therefore, when the L fixing signal gu0 is set to logic “H”, the switching element 12 is fixed in the off state and the switching element 14 is fixed in the on state, as in the first embodiment.

  When the command voltage vuc falls within “½” of the power supply voltage Ve, the L fixing signal gu0 is inverted to logic “L”. For this reason, a negative determination is made in step S64. When it is determined in step S68 that the output signal gu of the hysteresis comparator 84 is logic “L”, the process proceeds to step S74. However, at this time, since the H fixing history signal gu1old is logic “L”, a negative determination is made in step S74, and this series of processes is temporarily ended. On the other hand, when the output signal gu is inverted to logic “H”, an affirmative determination is made in step S68, and the process proceeds to step S70. Here, since the L fixing history signal gu0old is logic “H”, an affirmative determination is made, and the routine proceeds to step S72. In step S72, the L fixing history signal gu0old is set to logic “L”, and the H fixing signal Gu1 is set to logic “H”. Thereby, the switching element 12 is switched to an on state, and the switching element 14 is switched to an off state.

  If this process is entered again in this state, a negative determination will be made in both S70 and S74 while both the H fixing signal gu1 and the L fixing signal gu0 are logic "L". The switching state is not switched as described above.

  According to this embodiment described above, the following effects can be obtained in addition to the effects (1) to (6) of the first embodiment.

  (11) In a region where the operating state of the switching elements 12 to 22 is not fixed, when the command voltages vuc, vvc, and vwc decrease, only switching that allows the negative electrode potential side of the inverter 10 and the motor 2 side to conduct is allowed. When vuc, vvc, and vwc rise, only switching that connects the positive electrode potential side of the inverter 10 and the motor 2 side is allowed. Thereby, 1-pulse control can be reliably performed by allowing switching of the operation state of the switching elements 12 to 22 only once.

(Fifth embodiment)
Hereinafter, a fifth embodiment will be described with reference to the drawings, focusing on differences from the first embodiment.

  FIG. 16 is a functional block diagram of processing related to generation of drive pulses for operating the switching elements 12 to 22 among the processing performed by the microcomputer 50. In FIG. 16, blocks having the same functions as those shown in FIG. 2 are given the same reference numerals for the sake of convenience.

  As shown in the figure, the present embodiment includes a current-voltage converter 120 that converts actual currents iu, iv, and iw into three-phase voltages vu, vv, and vw. Here, the voltages vu, vv, vw are sinusoidal phase voltages assumed when the actual currents iu, iv, iw are passed. The voltages vu, vv, vw are not actual phase voltages of the motor 2, but using the voltages vu, vv, vw, the H fixing signals gu1, gv1, gw1 and the L fixing signals gu0, gv0, gw0 Can be generated.

  Also according to this embodiment described above, the effects (1) to (4) and (6) of the first embodiment can be obtained.

(Sixth embodiment)
Hereinafter, the sixth embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 17 shows a functional block diagram of processing related to generation of drive pulses for operating the switching elements 12 to 22 among the processing performed by the microcomputer 50. In FIG. 17, blocks having the same functions as those shown in FIG. 2 are given the same reference numerals for the sake of convenience.

  As shown in the figure, in the present embodiment, the switching restriction unit 114 generates the H fixing signals gu1, gv1, gw1 and the L fixing signals gu0, gv0, gw0 based on the required torque and the rotation speed. Here, the H fixing signals gu1, gv1, gw1 and the L fixing signals gu0, gv0, gw0 are map-calculated based on the required torque and the rotational speed. This map obtains in advance a one-pulse control drive pulse that is assumed when the required current is supplied to the motor 2, and the H-fixing signals gu1, gv1 so as to limit switching except in the vicinity of the inversion timing of the drive pulse. , Gw1 and the L fixing signals gu0, gv0, gw0. Then, after creating this map, it is stored in the switching restriction unit 114.

  Also according to this embodiment described above, the effects (1) to (4) and (6) of the first embodiment can be obtained.

(Other embodiments)
Each of the above embodiments may be modified as follows.

  In the first to fourth embodiments, the switching limitation is performed based on the command currents idc and iqc calculated from the required torque and the rotation speed. However, the switching limitation method based on the command current is not limited to this. . For example, the final command currents idc and iqc are calculated from the basic values of the command currents idc and iqc calculated from the required torque and the rotational speed and the two-phase actual currents id and iq, and the limit is performed based on the calculated command currents idc and iqc. Good.

  In the second embodiment, the predetermined phase φ may be variably set according to the required torque and the rotation speed. That is, as the rotational speed increases and the required torque increases, a higher voltage utilization rate is required, so it is desirable to approximate one-pulse control. For this reason, the phase φ may be set to a smaller value as the rotational speed is higher and the required torque is larger. As a result, as the rotational speed increases and the required torque increases, it is possible to perform control that approximates one-pulse control by instantaneous current value control.

  In the third embodiment, the upper limit value Imax and the lower limit value Imin are variable according to the required torque and rotational speed, or according to the command currents idc, iqc, or the command currents iuc, ivc, iwc in three phases. Also good. For example, the actual currents iu, iv, iw are suitable for the command currents iuc, ivc, iwc by increasing the absolute values of the upper limit value Imax and the lower limit value Imin as the amplitude of the command currents iuc, ivc, iwc in three phases increases. Can be followed. Even in this case, it is desirable that the upper limit value Imax and the lower limit value Imin are not more than the maximum rated current of the inverter 10.

  In each of the above embodiments, it is assumed that the present invention is applied to the motor 2. However, the present invention is not limited to this, and the present invention may be applied to a generator. In this case, the case where the output torque of the motor 2 is large may be read as the case where the torque of the generator is negative and large (when the load torque is large).

  In each of the above embodiments, only the instantaneous current value control is used as the switching method of the inverter 10, but the present invention is not limited to this. For example, in the low rotation speed and small torque region of the motor 2, PWM control using a triangular wave as a carrier wave may be employed.

  The control device for the multi-phase rotating electrical machine is not limited to that mounted on the hybrid vehicle, but may be mounted on an electric vehicle, for example.

  In the above embodiments, the control processing is not limited to the microcomputer. For example, a hard device such as an FPGA or a dedicated LSI may be used.

The figure which shows the structure of the motor, inverter, and microcomputer concerning 1st Embodiment. The functional block diagram which shows the process in the microcomputer concerning the embodiment. The time chart which shows the aspect of the instantaneous electric current value control in the embodiment. The time chart which shows the aspect of 1 pulse control. The time chart which shows the restriction | limiting aspect of switching in the said embodiment. The figure which shows the structure of the drive pulse restriction | limiting part in the embodiment. The flowchart which shows the process sequence of the restriction | limiting of switching in the embodiment. The time chart which shows the aspect of the instantaneous electric current value control in 2nd Embodiment. The functional block diagram which shows the process in the microcomputer concerning 2nd Embodiment. The flowchart which shows the process sequence of the restriction | limiting of switching in the embodiment. The time chart which shows the aspect of the instantaneous electric current value control in 3rd Embodiment. The figure which shows the structure of the drive pulse restriction | limiting part in the embodiment. The flowchart which shows the process sequence of the restriction | limiting of switching in the embodiment. The figure which shows the structure of the drive pulse restriction | limiting part in 4th Embodiment. The flowchart which shows the process sequence of the restriction | limiting of switching in the embodiment. The functional block diagram which shows the process in the microcomputer concerning 5th Embodiment. The functional block diagram which shows the process in the microcomputer concerning 6th Embodiment.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 2 ... Motor (one embodiment of a multiphase rotating electrical machine), 10 ... Inverter, 12-22 ... Switching element, 50 ... Microcomputer (One embodiment of a multiphase rotating electrical machine).

Claims (9)

Based on the magnitude of the upper and lower limits of the predetermined hysteresis region determined by the command current for the multiphase rotating electrical machine and the actual current flowing through the multiphase rotating electrical machine, the actual current is obtained as the required torque by operating the switching element of the inverter. In a control device for a multiphase rotating electrical machine that performs feedback control to a required current value for generating
For a phase in which a sine wave-shaped phase voltage assumed when the required current is supplied to the multiphase rotating electrical machine exceeds the positive potential of the inverter, the negative potential side of the inverter and the multiphase rotating electrical machine side are in a conductive state. For the phase in which the phase voltage of the sine wave assumed when the required current is supplied to the multiphase rotating electrical machine is lower than the negative potential of the inverter, the positive potential side of the inverter and the multiphase A control device for a multiphase rotating electrical machine, comprising limiting means for restricting the rotating electrical machine side from being in a conductive state. Based on the magnitude of the upper and lower limits of the predetermined hysteresis region determined by the command current for the multiphase rotating electrical machine and the actual current flowing through the multiphase rotating electrical machine, the actual current is obtained as the required torque by operating the switching element of the inverter. In a control device for a multiphase rotating electrical machine that performs feedback control to a required current value for generating
Outside the predetermined phase range centered on the zero-cross point of the sinusoidal phase voltage assumed when the required current is supplied to the multiphase rotating electrical machine, the negative phase of the inverter for the phase in which the phase voltage is positive The potential side and the multiphase rotating electrical machine side are restricted from being in a conducting state, and the positive potential side of the inverter and the multiphase rotating electrical machine side are in a conducting state for a phase in which the phase voltage is negative. A control device for a multi-phase rotating electrical machine, comprising limiting means for limiting The multi-phase rotating electrical machine is a 3-phase rotating electrical machine;
3. The control device for a multi-phase rotating electrical machine according to claim 2, wherein the predetermined phase range is “± 30 °” or more. Based on the magnitude of the upper and lower limits of the predetermined hysteresis region determined by the command current for the multiphase rotating electrical machine and the actual current flowing through the multiphase rotating electrical machine, the actual current is obtained as the required torque by operating the switching element of the inverter. In a control device for a multiphase rotating electrical machine that performs feedback control to a required current value for generating
By fixing the operation state of the switching element except in the vicinity of the inversion timing of the drive pulse assumed when the required torque is realized by one-pulse control using a drive pulse that matches the half cycle of the required current value. A control device for a multiphase rotating electrical machine, comprising limiting means for limiting the operation of the switching element by the feedback control. The limiting means is configured such that when a phase voltage having a sinusoidal shape assumed when the required current is supplied to the multiphase rotating electric machine is reduced in a region other than the region where the limitation is performed, the inverter negative electrode potential side and the multiphase rotation only allow switching to conduct the electrical side, according to claim 4, wherein when said phase voltage increases, characterized in that the allowing only switching to conduction between the multiphase rotary electric machine side and the positive potential side of the inverter Multiphase rotating electrical machine control device. Based on the magnitude of the upper and lower limits of the predetermined hysteresis region determined by the command current for the multiphase rotating electrical machine and the actual current flowing through the multiphase rotating electrical machine, the actual current is obtained as the required torque by operating the switching element of the inverter. In a control device for a multiphase rotating electrical machine that performs feedback control to a required current value for generating
The operation of the switching element by the feedback control is limited except in the vicinity of the inversion timing of the drive pulse assumed when the required torque is realized by one-pulse control using a drive pulse that matches the half cycle of the required current value. Therefore, a control device for a multi-phase rotating electrical machine, comprising limiting means for increasing the absolute value of the command current above the required current value. The control device for a multi-phase rotating electrical machine according to claim 6 , wherein the limiting means sets the absolute value of the command current to be equal to or less than a maximum rated current of the inverter at the time of the limitation. The restricting means restricts the restriction when a value that is twice the amplitude of a sinusoidal phase voltage that is assumed when the current of the required current value flows through the multiphase rotating electrical machine exceeds the input voltage of the inverter. The control device for a multi-phase rotating electrical machine according to any one of claims 1 to 7, wherein the control device is a control device. The limiting means includes means for calculating a sinusoidal phase voltage for flowing the command current to the multiphase rotating electrical machine based on the command current, and performs the limitation based on the calculated phase voltage. The control apparatus for a multi-phase rotating electrical machine according to any one of claims 1 to 8 .

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