JP4527101B2 - Burst radio signal transmission system, radio transmitter, radio receiver and burst radio signal transmission method - Google Patents

Description

  The present invention relates to a burst wireless signal transmission system, a wireless transmission device, a wireless reception device, and a burst wireless signal transmission method for wirelessly transmitting a burst signal.

  In order to wirelessly transmit the burst signal, the radio reception station side needs reception synchronization functions of burst detection, frequency synchronization, symbol synchronization, and frame synchronization. Examples of conventional burst radio signal transmission systems include satellite communication systems Dynet and PHS (for example, see Non-Patent Documents 1 and 2).

  FIGS. 35A and 35B are conceptual diagrams showing a frame configuration of a burst radio signal transmission system (hereinafter referred to as a first conventional system) described in Non-Patent Documents 1 and 2. FIG. FIG. 36 is a block diagram showing a functional configuration of reception synchronization processing of the first conventional system. The first conventional system has a frame configuration in which, in order to realize the above-described reception synchronization function, in the Dynanet, burst detection, carrier recovery (CR) symbol for frequency synchronization, bit timing recovery (BTR) symbol for symbol synchronization, frame Three types of training signals for unique word (UW) symbols for synchronization, PHS is composed of preamble (PR) symbols for burst detection, frequency synchronization and symbol synchronization, UW symbols for frame synchronization, and two types of training signals. ing.

  In the first conventional system described above, since frame synchronization is established by comparing the UW reception reproduction bit string with the original UW sequence, training for burst detection, frequency synchronization, and symbol synchronization necessary for bit reproduction is performed. It is necessary to separate the signal and the training signal (UW) dedicated to frame synchronization, and a plurality of training signals as shown in FIGS. For this reason, the overhead due to the training signal increases, and the transmission efficiency deteriorates particularly in a short packet with a short payload portion.

  As shown in FIG. 36, the first conventional system performs these four reception synchronization processes by separate calculation processes, and performs heavy calculations such as autocorrelation and DFT (Discrete Fourier Transform) for each. Therefore, there is a problem that the reception processing circuit scale and power consumption increase. Furthermore, since initial pull-in of frequency synchronization cannot be performed in one burst, there is a problem that it takes time for initial synchronization.

As described above, the first conventional system has the following three problems.
(1) Since a UW dedicated to frame synchronization is required, the overhead increases, and the transmission efficiency deteriorates particularly when the payload portion is short.
(2) Since each reception synchronization process is performed as a separate process, the amount of calculation is large, and the circuit scale and power consumption of the receiving station increase.
(3) The initial acquisition of frequency synchronization cannot be performed in one burst.

  Among these, as a conventional burst radio signal transmission system that solves the above (3), there is IEEE 802.11a (hereinafter referred to as a second conventional system) (for example, see Non-Patent Document 3).

  FIG. 37 is a conceptual diagram showing the frame configuration of the second conventional system. In the second conventional system, in order to realize the reception synchronization function, a frame configuration in which two types of training signals, a short preamble for frame synchronization, symbol synchronization and frequency synchronization, and a long preamble for channel estimation, are arranged at the head of a burst. I take the. FIGS. 38A and 38B are functional block diagrams for explaining reception synchronization processing of the second conventional system. A sliding cross-correlation operation is performed on the short preamble portion (23-1), and by detecting the peak position of the cross-correlation value (23-2), frame synchronization and symbol synchronization are established (23-3), Further, the autocorrelation value calculation is performed on the short preamble (23-4), and the phase synchronization of the autocorrelation value is detected (23-5), thereby establishing frequency synchronization (23-6: frequency synchronization). May also use long preambles).

  Note that since frame synchronization can be established directly from the detection of the peak position of the cross-correlation value, that is, the burst head position can be acquired, it is not necessary to perform burst detection before each synchronization process as in the first conventional system. As a result, since each synchronization can be pulled in with one burst, the above (3), which is a problem of the initial synchronization time found in the first conventional system, is solved.

  However, since IEEE802.11a is an OFDM (Orthogonal Frequency Division Multiplexing) synchronous detection method, channel estimation of each subcarrier is necessary, and a training signal in which uniform amplitude is assigned to all subcarriers is necessary. The short preamble needs to have a time period shorter than one OFDM symbol in consideration of the frequency synchronization pull-in range due to autocorrelation, and it is difficult to make such a training signal pattern compatible.

  Therefore, in addition to the short preamble, a long preamble that is a training signal sequence for channel estimation is required. Therefore, the problem (1) that is a problem when the overhead seen in the first conventional system becomes large cannot be solved. Further, since the second conventional system also requires two calculation processes, an autocorrelation value calculation and a cross-correlation value calculation, for the reception synchronization process, the amount of calculation of the receiving station found in the first conventional system The problem (2) that there are many is not solved. Furthermore, since a short preamble having a short time period does not become a pseudo-random signal sequence (PN (Psue Noise) sequence), the autocorrelation as orthogonal as the PN sequence cannot be obtained.

  Accordingly, the peak detection accuracy of the cross-correlation value is lower than that of the PN sequence, and there is a problem that the symbol synchronization accuracy and the frame synchronization accuracy are deteriorated as compared with the PN sequence. In addition, for low-speed communications where the amount of phase rotation within the short preamble due to carrier frequency offset is not negligible, each term has a different phase amount during product-sum when calculating the cross-correlation value, so each term can cancel out There is a possibility that the cross-correlation value does not peak at a desired time position. Therefore, there has been a problem that the symbol synchronization accuracy and the frame synchronization accuracy deteriorate even during low-speed communication.

Thus, although the problem (3) of the first conventional system has been solved, the second conventional system has the following problems.
(1) Since a training signal for channel estimation is required in addition to the training signal for each synchronization, overhead is increased, and transmission efficiency is deteriorated particularly when the payload portion is short.
(2) Since the cross-correlation calculation and autocorrelation calculation are performed in each reception synchronization process, the circuit scale and power consumption of the receiving station increase.
(4) Since the short preamble does not become a PN sequence, the symbol synchronization and frame synchronization accuracy using the short preamble deteriorates more than the synchronization accuracy using the PN sequence.
(5) In low-speed communication, since the phase rotates within the short preamble, symbol synchronization and frame synchronization accuracy deteriorate.

  Among these, HiSWANA (hereinafter referred to as third conventional system) is known as a conventional burst radio signal transmission system that solves (2) and (5) (for example, Non-Patent Document 4).

  FIG. 39 is a conceptual diagram showing the frame configuration of the third conventional system. In order to realize the reception synchronization function, in the third conventional system, a frame in which two types of training signals of preambles A and B for frame synchronization, symbol synchronization and frequency synchronization and preamble C for channel estimation are arranged at the head of a burst. Take composition.

  FIG. 40 is a functional block diagram for explaining the reception synchronization processing of the third conventional system. A sliding autocorrelation calculation is performed on the preambles A and B (24-1), and the peak position of the autocorrelation value is detected (24-3), thereby establishing frame synchronization and symbol synchronization (24-3). In addition, by detecting the phase component of the autocorrelation value (24-4), frequency synchronization is established (24-5: frequency synchronization may also use preamble C).

  As in the second conventional system, since the head of the burst can be detected by detecting the peak position of the autocorrelation value, it is necessary to perform burst detection before each reception synchronization process as in the first conventional system. Frame synchronization can be established.

  This solves the problem (2) that the computation amount of the receiving station increases in the first and second conventional systems because the computation processing required for the reception synchronization processing is only autocorrelation computation. Yes. In addition, since autocorrelation calculation is used for symbol synchronization and frame synchronization, even if the phase is rotated due to the carrier frequency offset in preambles A and B, at the time of product-sum calculation of autocorrelation calculation, Since the phases are integrated and integrated, an assumed autocorrelation value can be obtained. Therefore, the problem (5) of deterioration of symbol synchronization and frame synchronization accuracy during low-speed communication, which is seen in the second conventional system, is solved.

  However, since the third conventional system is also an OFDM synchronous detection method, as in the second conventional system, a training signal (preamble C) dedicated to channel estimation of each subcarrier is required. When the overhead seen in the conventional system becomes large, the problem (1) cannot be solved. Similarly, in the third conventional system, since the preambles A and B are signal sequences having a short time period for frequency synchronization, compared with the PN sequence found in the second conventional system, the symbol synchronization, The problem (4) that frame synchronization deteriorates cannot be solved.

As described above, the following problems remain in the third conventional system.
(1) Since a training signal for channel estimation is required in addition to the training signal for each synchronization, overhead is increased, and transmission efficiency is deteriorated particularly when the payload portion is short.
(4) Since the cross correlation value peak of the short preamble is less likely to occur than the PN sequence, the symbol synchronization and frame synchronization accuracy using the short preamble deteriorates compared to the synchronization accuracy using the PN sequence.

  Further, as a technique for solving the problem (2) of the second and third conventional systems and the problem (5) of the second conventional system, the UW differential encoding is used for frame synchronization and the cross correlation detection is used. A burst radio signal transmission system (hereinafter referred to as a fourth conventional system) that performs delay detection before peak detection has also been proposed (see, for example, Non-Patent Document 5).

  FIG. 41 shows a frame configuration of the fourth conventional system, and FIG. 42 is a functional block for explaining reception synchronization processing of the fourth conventional system. In the fourth conventional system, the transmission side transmits a sequence obtained by differentially encoding UW that tends to have a peak as a preamble part, and the receiving side establishes frame synchronization by detecting the cross correlation after the delay detection. To do. In the fourth conventional system, by performing delay detection on the receiving side, even if the phase is rotated in the preamble portion due to the carrier frequency offset, the phase rotation after delay detection is all the amount of phase rotation between 1 symbol, A desired cross-correlation value can be detected. Thereby, the problem (5) of the second conventional system is solved. Further, since the UW sequence is obtained after the delay detection, the problem (4) of the second and third conventional systems can be solved if the UW is changed to the PN sequence.

  However, since the fourth conventional system is based on the premise that symbol synchronization is established at the time of cross-correlation calculation, symbol synchronization cannot be established at the same time during frame synchronization (delay detection + cross-correlation calculation). In addition to the preamble for frame synchronization, a preamble for burst detection, symbol synchronization, and frequency synchronization is required. Also, as the reception synchronization processing, symbol synchronization and frequency synchronization processing are required separately from frame synchronization.

  Therefore, even in the fourth conventional system, the problem (1) of the first, second, and third conventional systems and the problem (3) of the first and second conventional systems cannot be solved. Further, since the delay detection is once performed to determine the bit, the amplitude values of the respective terms at the product-sum time in the cross-correlation calculation are all the same. Therefore, when reception level fluctuations occur in the preamble due to fading fluctuations, the magnitude of the reception level is not reflected during the cross-correlation calculation. Therefore, there is a problem that the frame synchronization accuracy is deteriorated.

As described above, the following problems still remain in the fourth conventional system.
(1) Since frame preambles and other preambles for reception synchronization are required, overhead increases, and transmission efficiency deteriorates particularly when the payload portion is large.
(2) Since each frame synchronization process and other burst detection, symbol synchronization, and frequency synchronization processes are separately required in each reception synchronization process, the circuit scale and power consumption of the receiving station are large.
(6) Since bit determination by delay detection is performed once at the time of cross-correlation, frame synchronization accuracy deteriorates when reception level fluctuations occur in the preamble due to fading fluctuations or the like.
Kiyoshi KOBAYASHI, Tetsu SAKATA, Youichi MATSUMOTO and Shuji KUBOTA, “Fully Digital Burst Modem for Satellite Multimedia Communication Systems”, IEICE Transaction on Communications, vol.E80-B, No.1, January 1997. Youichi MATSUMOTO, Shuji KUBOTA and Shuzo KATO, “A New Burst Cocherent Demodultor for Microcellular TDMA / TDD Systems”, IEICE Transaction on Communcations, vol.E77-B, No.7 July 1994. Hideaki Matsue, Masahiro Morikura, “802.11 High-Speed Wireless LAN Textbook”, Chapter 9, IDG Japan Shingo Mochizuki, Satoshi Souda, Makoto Ise, Shota Ueno, Yoichi Matsumoto, Masahiro Umehira, “Development and Performance Evaluation of OFDM-LSI for 5 GHz Band High-Speed Wireless Access”, IEICE Technical Report, DSP99-168, SAT99-123, RCS99-173 (2000-01), PP113-118; IEICE Hiroshi Ogura, Kaoru Serizawa, “Slot Synchronization Method for TDMA Digital Mobile Communications”, B-292, 1990 Autumn Meeting of IEICE

  As described above, each of the first to fourth conventional systems has any of the above-described problems, such as transmission efficiency deterioration due to overhead, circuit scale, increase in power consumption, symbol synchronization, frame synchronization. The problem of accuracy degradation cannot be solved.

  The present invention has been made in consideration of such circumstances, and an object of the present invention is to improve deterioration of transmission efficiency due to overhead in a burst radio signal transmission system, and to achieve symbol synchronization and frame synchronization accuracy with a small circuit scale. The present invention provides a burst radio signal transmission system, a radio transmission apparatus, a radio reception apparatus, and a burst radio signal transmission method.

  In order to solve the above-described problems, the present invention provides a burst radio signal transmission system including a radio transmission station and a radio reception station, wherein the radio transmission station converts an input data bit sequence into an information symbol sequence. Information symbol converting means, training signal generating means for differentially encoding a designated code sequence that is a previously designated code sequence to generate a training signal sequence, and time multiplexing the information symbol sequence and the training signal sequence Multiplexing means for forming a transmission burst signal, radio transmission means for converting the transmission burst signal into a radio signal, and a transmission antenna for radio transmission of the radio signal in space, the radio reception station, A receiving antenna for receiving a wireless signal; wireless receiving means for converting the wireless signal into a digital baseband signal; and an input signal On the other hand, a reception digital filter means for performing band limitation, a differential decoding means for performing differential decoding of the input digital signal at symbol intervals for each sample, and a signal output from the differential decoding means For a sequence, the specified code sequence and a sliding cross-correlation calculating means for performing a sliding cross-correlation calculation for each sample, a peak determination is performed on the cross-correlation value of each sample output by the sliding cross-correlation calculating means, The signal position corresponding to the cross-correlation value determined to be the peak is used as the head position of the transmission burst signal, the frame synchronization means for extracting the transmission burst portion, the signal position determined to be the peak is the symbol point, and the input signal A symbol synchronization means for extracting only the symbol points and obtaining a phase component of the cross-correlation value determined as the peak, The phase component is regarded as a phase rotation per symbol due to a carrier frequency offset, and frequency synchronization means for correcting the carrier frequency offset with respect to the input signal, and the signal corrected by the frequency synchronization means by demodulating the data bit And demodulating means for reproducing the sequence.

  The present invention is characterized in that, in the above invention, a PN code is used as the designated code sequence.

  According to the present invention, in the above invention, the radio receiving station performs processing of the frame synchronization unit, processing of the frequency synchronization unit, and processing of the reception digital filter unit with respect to an information symbol sequence output from the radio reception unit. Each processing is performed in the order of processing of the symbol synchronization means.

  According to the present invention, in the above invention, the radio reception station includes N reception antennas (N is an integer), and includes the radio reception means, the reception digital filter means, and the differential decoding means. The N reception signals received by the N reception antennas are processed by the wireless reception unit, the reception digital filter unit, and the differential decoding unit, and the sliding cross correlation calculation unit includes: The output signal sequence of each of the N differential decoding units is regarded as one signal sequence, and a sliding cross-correlation operation is performed with N of the designated code sequence for each sample. The frame synchronization unit includes the sliding unit The peak is determined using the cross-correlation value of each sample output by the cross-correlation calculation means, and the signal position corresponding to the cross-correlation value determined to be the peak is determined at the beginning of the transmission burst signal. For example, the transmission burst signal portion is extracted for N input signals, and the symbol synchronization means uses the signal position determined to be the peak as a symbol point, and the symbol point for the N input signals. And the frequency synchronization means obtains a phase component of the cross-correlation value determined to be the peak, regards the phase component as a phase rotation per symbol due to a carrier frequency offset, and outputs N phase signals. Then, the carrier frequency offset correction is performed, and the demodulating means performs diversity combining on the N received signals offset-corrected by the frequency synchronizing means to reproduce the original data bit sequence. Features.

  According to the present invention, in the above invention, the radio reception station includes N reception antennas (N is an integer), the radio reception means, the reception digital filter means, the differential decoding means, and the sliding cross-correlation. Comprising N calculation means, frame synchronization means, and symbol synchronization means, and for N received signals received by the N receiving antennas, the radio receiving means, the reception digital filter means, and the differential The decoding means, the sliding correlation calculation means, the frame synchronization means, and the symbol synchronization means are each independently processed, and the frequency synchronization means includes N peaks calculated from the N frame synchronization means. All the determined cross-correlation values are added, the phase component of the added cross-correlation value is calculated, and the calculated phase component is calculated by the carrier frequency offset. The carrier frequency offset correction is performed on N received signals, assuming phase rotation per symbol, and the demodulation means includes the received digital filter means, the frame synchronization means, the symbol synchronization means, and the frequency synchronization. Diversity combining is performed on the N received signals subjected to the processing of the means to reproduce the original data bit sequence.

  According to the present invention, in the above invention, the radio transmitting station further includes second training signal generating means for generating an alternating code that repeats +1 and −1 in two symbol periods as a second training signal sequence, and the multiplexing means The information symbol sequence, the training signal sequence, and the second training signal sequence are time-multiplexed to form a transmission burst signal, and the radio receiving station transmits the second training signal as an output signal of the frame synchronization means. The amplitude square value is subjected to discrete Fourier transform processing on the series portion, the phase component of the discrete Fourier transform value is regarded as a symbol phase error, and the tap of the reception digital filter corresponding to the symbol phase error A second symbol phase error compensation is performed on the output signal from the symbol synchronization means by adjusting the coefficient. Further characterized by comprising a symbol synchronization unit.

  According to the present invention, in the above invention, the radio transmitting station further includes second training signal generating means for generating an alternating code that repeats +1 and −1 in two symbol periods as a second training signal sequence, and the multiplexing means The information symbol sequence, the training signal sequence, and the second training signal sequence are time-multiplexed to form a transmission burst signal, and the radio receiving station applies each of the N output signals of the frame synchronization means to each other. On the other hand, the second training signal sequence portion is extracted, the same time signals of the N second training signal sequences are squared and added, and the added signal sequence is subjected to discrete Fourier transform processing. The phase component of the value obtained by the discrete Fourier transform is regarded as a symbol phase error, and the N reception digital filters correspond to the symbol phase error. By adjusting the up factor, and further comprising a N second symbol synchronization means for performing the further symbol phase error compensation on the output signal from the symbol synchronization section.

  According to the present invention, in the above invention, the radio transmitting station further includes second training signal generating means for generating an alternating code that repeats +1 and −1 in two symbol periods as a second training signal sequence, and the multiplexing means The information symbol sequence, the training signal sequence, and the second training signal sequence are time-multiplexed to form a transmission burst signal, and the radio receiving station applies each of the output signals of the N frame synchronization means to each of the output signals. On the other hand, the second training signal sequence portion is extracted, the same time signals of the N second training signal sequences are squared and added, and the added signal sequence is subjected to discrete Fourier transform processing. The phase component of the value obtained by the discrete Fourier transform is regarded as a symbol phase error, and the N reception digital filters correspond to the symbol phase error. By adjusting the up factor, and further comprising a N second symbol synchronization means for performing the further symbol phase error compensation on the output signal from the symbol synchronization section.

  According to the present invention, in the above invention, the radio receiving station may apply the training signal sequence portion or the second training signal sequence after the processing by the frame synchronization unit, the symbol synchronization unit, and the frequency synchronization unit is performed. On the other hand, by multiplying the training signal sequence or the second training signal sequence for each symbol, the phases of the symbols of the training signal sequence portion or the second training signal sequence portion are aligned, and then the phases are aligned. An autocorrelation operation with a correlation interval of 2 symbols or more is performed on the training signal sequence or the second training signal sequence, and a carrier frequency offset is estimated from the phase component of the autocorrelation operation value. For the previous stage signal, the estimated carrier frequency offset is used to capture the signal. Further characterized by comprising a second frequency synchronizing means for performing A frequency offset compensation.

  According to the present invention, in the above invention, the radio receiving station may perform the N training signal sequence portions or the second training after the processing by the frame synchronization unit, the symbol synchronization unit, and the frequency synchronization unit. After aligning the phase of each symbol of the training signal sequence portion or the second training signal sequence portion by multiplying the signal sequence by the training signal sequence or the second training signal sequence for each symbol, An autocorrelation operation with a correlation interval of 2 symbols or more is performed on the training signal sequence or the second training signal sequence having the same phase, the N autocorrelation operation values are added, and then the added self After estimating the carrier frequency offset from the phase component of the correlation calculation value, N signals before the demodulation means In contrast, further characterized by comprising a second frequency synchronization to perform carrier frequency offset compensation using the carrier frequency offset that the estimation.

  According to the present invention, in the above invention, the radio transmission station converts the data bit sequence into an information symbol sequence for synchronous detection on the assumption that the radio reception station performs synchronous detection as the information symbol conversion means. Comprising synchronous detection conversion means, wherein the radio receiving station estimates a propagation channel using the training signal sequence or the second training signal sequence after the frame synchronization, the symbol synchronization, and the frequency synchronization is established, The same number of channel estimation means as the receiving antenna, and the same number of synchronous detection means as the receiving antenna for synchronously detecting the output signal from the frequency synchronization means using the propagation channel estimated by the channel estimation means It is characterized by comprising.

  According to the present invention, in the above invention, the radio transmission station converts the data bit sequence into an information symbol sequence for synchronous detection on the assumption that the radio reception station performs synchronous detection as the information symbol conversion means. The apparatus further comprises a synchronous detection converting means and a pilot signal sequence generating means for generating a pilot signal sequence designated in advance, and the multiplexing means inserts the pilot signal sequence into the information symbol sequence at regular intervals. The radio reception station estimates a radio propagation channel from the pilot signal sequence after frame synchronization, symbol synchronization, and frequency synchronization is established, and is estimated by the same number of channel estimation means as the reception antenna and the channel estimation means Synchronously detecting an output signal from the frequency synchronization means using a radio propagation channel; and Flip characterized in that it further comprises a synchronous detection unit number.

  The present invention is characterized in that, in the above invention, the channel estimation means updates an estimated value of a propagation channel for each pilot signal.

  The present invention is characterized in that, in the above invention, the channel estimation means interpolates a radio propagation channel between the pilot signals from an estimated value of a radio channel estimated from the pilot signal.

  According to the present invention, in the above invention, the channel estimation means extrapolates a radio propagation channel between the pilot signals or after the pilot signal from an estimated value of the propagation channel estimated from the pilot signal. And

  According to the present invention, in the above invention, the wireless transmission station further includes error correction encoding means for performing error correction encoding on the data bit sequence before the information symbol conversion means, and the wireless reception station includes: As an alternative to the demodulating means, a soft decision error is made by using, as a likelihood weight, the amplitude square value of the propagation channel estimated by the channel estimating means for the output signal of the synchronous detecting means after the synchronous detecting means. It further comprises error correction decoding means for correcting and decoding.

  According to the present invention, in the above invention, the radio transmission station converts the data bit sequence into an information symbol sequence for delay detection on the assumption that the radio reception station performs delay detection as the information symbol conversion means. The radio receiving station further includes a delay detection means for performing differential decoding on an input signal of the demodulation means before the demodulation means. .

  According to the present invention, in the above invention, the wireless transmission station further includes error correction encoding means for performing error correction encoding on the data bit sequence before the information symbol conversion means, and the wireless reception station includes: As an alternative to the demodulation means, there is provided an error correction decoding means for performing soft decision error correction decoding using the amplitude value of the output signal of the delay detection means as a likelihood weight.

  In order to solve the above-described problem, the present invention provides an information symbol conversion means for converting an input data bit sequence into an information symbol sequence in a wireless transmission device that wirelessly transmits a burst signal, and a code specified in advance. Training signal generating means for differentially encoding a designated code sequence that is a sequence to generate a training signal sequence; multiplexing means for time-multiplexing the information symbol sequence and the training signal sequence to form a transmission burst signal; The wireless transmission means for converting the transmission burst signal into a wireless signal, and a transmission antenna for wirelessly transmitting the wireless signal in space are provided.

In order to solve the above-described problem, the present invention provides a radio receiving apparatus that receives a burst signal transmitted by radio, a reception antenna that receives a radio signal, and a radio that converts the radio signal into a digital baseband signal. Receiving means, receiving digital filter means for performing band limitation on the input signal, differential decoding means for differentially decoding the input digital signal at symbol intervals for each sample, and the differential With respect to the signal sequence output from the decoding means , a specified code sequence that is a pre-specified code sequence and a sliding cross-correlation calculation means for performing a sliding cross-correlation calculation for each sample, Peak determination is performed on the cross-correlation value of the sample, and the signal position corresponding to the cross-correlation value determined to be the peak is transmitted. A frame synchronization means for extracting the transmission burst portion as a leading position of a first signal, a symbol synchronization means for extracting only a symbol point from an input signal, using the signal position determined to be the peak as a symbol point, and the peak A frequency synchronization unit that obtains a phase component of the determined cross-correlation value, regards the phase component as a phase rotation per symbol due to a carrier frequency offset, and performs the carrier frequency offset correction on an input signal; and the frequency synchronization unit And demodulating means for reproducing the data bit sequence by demodulating the signal corrected by the offset.

  In order to solve the above-described problem, the present invention provides a burst wireless signal transmission method for wirelessly transmitting a burst signal between a wireless transmission station and a wireless reception station. An information symbol conversion step for converting a bit sequence into an information symbol sequence, a training signal generation step for generating a training signal sequence by differentially encoding a designated code sequence which is a pre-designated code sequence, the information symbol sequence, A multiplexing step for time-multiplexing a training signal sequence to form a transmission burst signal; a radio transmission step for converting the transmission burst signal into a radio signal; and a transmission step for radio transmission of the radio signal in space. The radio receiving station side receives a radio signal, and the radio signal is digitally A radio reception step for converting the received digital signal, a filtering step for band-limiting the input signal, and a differential decoding step for differentially decoding the input digital signal at symbol intervals for each sample. A sliding cross-correlation operation step for performing a sliding cross-correlation operation for each of the designated code sequence and the sample with respect to the differentially decoded signal sequence; and a cross-correlation value of each sample on which the sliding cross-correlation operation has been performed. The frame synchronization step for performing peak determination, extracting the transmission burst portion with the signal position corresponding to the cross-correlation value determined to be a peak as the start position of the transmission burst signal, and the signal position determined as the peak Symbol synchronization step for extracting only symbol points from the input signal Obtaining a phase component of the cross-correlation value determined to be the peak, considering the phase component as a phase rotation per symbol due to a carrier frequency offset, and performing a frequency synchronization step of performing the carrier frequency offset correction on the input signal; A demodulation step of demodulating the offset-corrected signal to reproduce a data bit sequence.

  The present invention is characterized in that, in the above invention, a PN code is used as the designated code sequence.

  According to the present invention, in the above invention, the radio receiving station side performs processing of the frame synchronization step, the frequency synchronization step, the filtering step, and the symbol synchronization step on the received radio signal, Each step is performed.

  According to the present invention, in the above invention, the radio receiving station side includes N receiving antennas, and for the N received signals received by the N receiving antennas, the receiving step, The filtering step and the differential decoding step are sequentially performed. In the sliding cross-correlation calculation step, the N decoded output signal sequences are regarded as one signal sequence, and the designation is performed for each sample. A sliding cross-correlation calculation is performed with N code sequences, and in the frame synchronization step, a peak determination is performed using a cross-correlation value of each sample on which the sliding cross-correlation calculation is performed, and a cross-correlation value determined to be a peak The transmission burst signal portion is extracted from N input signals, with the signal position corresponding to In the symbol synchronization step, the signal position determined as the peak is used as a symbol point, and only the symbol point is extracted from N input signals. In the frequency synchronization step, the phase component of the cross-correlation value determined as the peak The phase component is regarded as a phase rotation per symbol due to a carrier frequency offset, and the carrier frequency offset correction is performed for each of N input signals. In the demodulation step, the offset correction is performed. An original data bit sequence is reproduced by performing diversity combining on N received signals.

  According to the present invention, in the above invention, the radio receiving station side includes N receiving antennas, and for the N received signals received by the N receiving antennas, the receiving step, A filtering step, the differential decoding step, the sliding correlation calculation step, the frame synchronization step, and the symbol synchronization step are performed independently. In the frequency synchronization step, the N peaks calculated in the frame synchronization step and All the determined cross-correlation values are added, a phase component of the added cross-correlation value is calculated, the calculated phase component is regarded as a phase rotation per symbol due to a carrier frequency offset, and N received signals are Then, the carrier frequency offset correction is performed, and in the demodulation step, the filtering step and the filter are performed. Over arm synchronization step, the symbol synchronization step, the frequency synchronization step has been performed, to the N received signals, characterized in that it reproduces the original data bit sequence by performing diversity combining.

  In the above invention, the present invention includes a second training signal generating step for generating an alternating code that repeats +1 and −1 in two symbol periods as a second training signal sequence on the wireless transmission station side, and the multiplexing step Then, the information symbol sequence, the training signal sequence, and the second training signal sequence are time-multiplexed to form a transmission burst signal, and on the radio receiving station side, the transmission burst portion extracted in the frame synchronization step Is subjected to discrete Fourier transform processing on the second training signal sequence portion, and the phase component of the discrete Fourier transform value is regarded as a symbol phase error, corresponding to the symbol phase error. It is extracted in the symbol synchronization step by adjusting the tap coefficient of the reception digital filter. And further comprising a second symbol synchronization step of performing a further symbol phase error compensation for symbol point.

  According to the present invention, in the radio transmission station, the input data bit sequence is converted into the information symbol sequence by the information symbol conversion means, and the designated code sequence that is a predesignated code sequence is converted by the training signal generation means. A training signal sequence is generated by dynamic coding, a transmission burst signal is formed by time multiplexing the information symbol sequence and the training signal sequence by a multiplexing means, and the transmission burst signal is converted into a radio signal by a wireless transmission means. Wirelessly transmitted from the transmitting antenna to the space, and at the wireless receiving station, the wireless receiving means converts the wireless signal received by the receiving antenna into a digital baseband signal, and the receiving digital filter means limits the band, The dynamic decoding means performs differential decoding at symbol intervals for each sample and performs sliding mutual phase. After the calculation means performs a sliding cross-correlation calculation for each sample with the designated code series on the signal series output from the differential decoding means, the frame synchronization means performs a peak on the cross-correlation value of each sample. The signal position corresponding to the cross-correlation value determined to be the peak is determined as the start position of the transmission burst signal, the transmission burst portion is extracted, the signal position determined to be the peak by the symbol synchronization means is set as the symbol point, and the input signal Only the symbol point is extracted, the phase component of the cross-correlation value determined as the peak is obtained by the frequency synchronization means, the phase component is regarded as the phase rotation per symbol due to the carrier frequency offset, and the input signal The carrier frequency offset is corrected, and the demodulating means demodulates the offset-corrected signal to Play the door series.

  Therefore, frame synchronization, symbol synchronization, and frequency synchronization can be realized by using only one type of training signal sequence for the overhead portion of the transmission burst signal, the overhead portion can be reduced, and transmission efficiency can be improved. Benefits are gained.

  In addition, since frame synchronization, symbol synchronization, and frequency synchronization can be processed only from the cross-correlation values, there is an advantage that the circuit scale of the receiving station can be reduced and power consumption can be reduced. In addition, in frame synchronization, symbol synchronization, and frequency synchronization, the processing is completed by receiving the training signal sequence only once, so that the frequency initial pull-in can be realized in one burst.

  Also, by performing the cross-correlation operation after differential decoding, even if the phase is rotated within the specified code sequence due to the carrier frequency offset, all the signals after differential decoding are based on this carrier frequency offset. Since the phase rotation is aligned with the rotation of one symbol, it is possible to align the phase of each term at the time of product-sum at the time of cross-correlation calculation. Therefore, even in burst communication using a low-speed line, it is possible to improve synchronization accuracy without degrading symbol synchronization and frame synchronization.

  In addition, according to the present invention, since the PN code is used as the designated code sequence, peak determination based on the cross-correlation value after differential decoding can be performed on the PN code, so that symbol synchronization using a short preamble is performed. The accuracy of frame synchronization can be improved. Also, the differentially decoded received signal is in the form of the square value of the propagation channel, that is, the power of the propagation channel multiplied by the designated code sequence before differential encoding. Even when the received signal level fluctuates due to fluctuation or the like, the cross-correlation value reflects the magnitude of the level of the received signal, so that the frame synchronization accuracy can be improved.

  Further, according to the present invention, in the radio reception station, for the information symbol sequence output from the radio reception means, processing of the frame synchronization means, processing of the frequency synchronization means, processing of the reception digital filter means, processing of the symbol synchronization means Since each process is performed in this order, frequency synchronization can be performed before the reception digital filter process. As a result, the position of the received signal spectrum whose center frequency is shifted due to the carrier frequency offset can be returned to the normal state before the band limiting process of the reception digital filter, so that the received signal whose center frequency is shifted due to the carrier frequency offset is reduced. Power consumption due to passing through the reception digital filter can be prevented.

  Further, according to the present invention, the radio receiving station includes N (N is an integer) receiving antennas and N radio receiving means, receiving digital filter means, and differential decoding means, respectively, thereby providing frame synchronization. Diversity processing can be performed on each of the symbol synchronization and frequency synchronization processing to obtain a spatial diversity effect.

  In the present invention, the radio receiving station includes N receiving antennas (N is an integer), N radio receiving means, receiving digital filter means, and differential decoding means, respectively. Since the processing by the means is performed independently in each reception system, diversity processing can be performed on the processing of the frequency synchronization unit, and a spatial diversity effect can be obtained. Therefore, when the frame synchronization point and symbol synchronization point are different in each receiving system and only the carrier frequency offset is the same in each receiving system, for example, each receiving antenna is installed remotely, and the distance between the transmitting antenna and the receiving antenna is Although the frequency differs depending on the antenna, the reference frequency supplied to each receiving system for frequency conversion is the same, that is, when the radio receiving station realizes receiving site diversity, the reception level varies depending on the location of the receiving antenna due to fading variation etc. In such a radio wave propagation environment, the radio receiving unit is equipped with a plurality of receiving antennas and receiving systems, so that the synchronization accuracy of frequency synchronization can be improved.

  Also, in the present invention, the overhead increases by inserting the second training signal for symbol synchronization into the frame structure, but the number of oversampling can always be maintained at 4 regardless of any required value of symbol synchronization. Compared with other parts, the amount of computation is large, and the amount of computation is proportional to the number of oversampling. The increase in the amount of computation of the receiving digital filter means and the sliding cross-correlation computing means can be prevented. Electric power can be reduced.

  In the present invention, the radio receiving station is provided with N (N is an integer) receiving antennas, N radio receiving means, reception digital filter means, and differential decoding means, respectively, and a second symbol synchronization second is provided. Since the training signal is inserted into the frame configuration, diversity processing can be performed for each processing of the frame synchronization means, symbol synchronization means, and frequency synchronization means to obtain a spatial diversity effect and increase the amount of calculation. Therefore, it is possible to reduce the circuit scale and power consumption of the radio receiving station.

  In the present invention, the radio receiving station includes N receiving antennas (N is an integer), N radio receiving means, receiving digital filter means, and differential decoding means, respectively. The processing of the means is performed independently in each receiving system, and the second training signal for symbol synchronization is inserted into the frame configuration, so the processing of the frequency synchronization means performs diversity processing and obtains a spatial diversity effect. In addition, it is possible to prevent an increase in the amount of calculation and reduce the circuit scale and power consumption of the radio receiving station.

  Also, in the present invention, the second frequency synchronization for performing the autocorrelation calculation with the autocorrelation interval extended by 2 symbols or more with respect to the training signal sequence is performed, so that the synchronization characteristic of the frequency synchronization is improved without increasing the overhead. Can be made.

  In the present invention, since the propagation channel estimation is performed using the training signal, frequency detection is enabled.

  Further, in the present invention, by inserting a pilot signal sequence for propagation channel estimation into an information symbol signal sequence, synchronous detection can be performed with high accuracy even when the propagation channel varies. Note that the estimated value of the propagation channel may be updated for each pilot signal. The radio propagation channel between pilot signals may be interpolated from the estimated value of the radio channel estimated from the pilot signal. Further, the radio propagation channel between pilot signals or after the pilot signal may be extrapolated from the estimated value of the propagation channel estimated from the pilot signal.

  In the present invention, since soft decision error correction codes are used, transmission quality can be improved.

  In the present invention, since delay detection is performed, synchronous detection can be performed by a simpler process in the radio receiving station.

  Hereinafter, a burst radio signal transmission system according to an embodiment of the present invention will be described with reference to the drawings.

<First Embodiment>
First, a first embodiment of the present invention will be described.
FIG. 1 is a conceptual diagram showing a frame configuration of the burst radio signal transmission system according to the first embodiment. In the first embodiment, a transmission burst signal is configured with a training signal sequence described later at the head and an information symbol signal sequence described later at the subsequent stage. The overhead in the first embodiment is only one type of the training signal sequence.

Next, the configuration of the wireless transmission station according to the first embodiment will be described.
FIG. 2A is a block diagram showing a configuration of a radio transmission station of the burst radio signal transmission system according to the first embodiment. In the figure, a radio transmission station includes an information symbol conversion unit 1-1, a training signal generation unit 1-2, a multiplexing unit 1-3, a transmission digital filter unit 1-4, a radio transmission unit 1-5, and a transmission antenna 1-. It is comprised from 6. FIG. 2B is a block diagram illustrating a configuration of the training signal generation unit 1-2. In the figure, the training signal generation unit 1-2 includes a designated code generation unit 1-2-1 and a differential encoding unit 1-2-2.

Next, a specific operation of each functional unit described above will be described along a series of sequences from modulation of a transmission burst signal to radio transmission. First, the radio transmission station converts a data bit sequence input to the radio transmission station into an information symbol signal sequence by the information symbol conversion unit 1-1. In addition, after a designated code sequence that is known in advance is generated between the radio transmission station and the radio reception station by the designation code generation unit 1-2-1 in the training signal generation unit 1-2, the differential encoding unit 1- The designated code sequence is differentially encoded in step 2-2. Specifically, the specified code _{sequence, a m ∈ {a 1,} a 2, a 3, ..., a Na-1, a Na} (a m = + 1or-1), the differential decoded signal The sequence is set as A _m ε {A ₁ , A ₂ , A ₃ ,..., A _Na−1 , A _Na , A _{Na + 1} }, and the following equation (1) is calculated.

Note that N _a is a sequence length of the designated code sequence, and A is an arbitrary complex number of size 1. In addition, for the sake of convenience, the designated code sequence A _m ε {A ₁ , A ₂ , A ₃ ,..., A _Na−1 , A _Na , A _{Na + 1} } that has been differentially encoded is hereinafter referred to as a training signal sequence. . One specific example of the designated code sequence is a PN code.

  Next, multiplexing section 1-3 time-multiplexes the information symbol signal sequence and the differentially encoded designated code sequence to form a transmission burst signal. Next, the transmission digital filter section 1-4 limits the band of the formed transmission burst signal before digital / analog conversion (D / A conversion). Thereafter, the transmission burst signal is D / A converted by the wireless transmission unit 1-5, converted to a radio frequency, and transmitted to the space by the transmission antenna 1-6.

Next, the configuration of the radio receiving station according to the first embodiment will be described.
FIG. 3 is a block diagram showing the configuration of the radio receiving station according to the first embodiment. In the figure, the radio reception station includes a reception antenna 2-1, a radio reception unit 2-2, a reception digital filter unit 2-3, a differential decoding unit 2-4, a sliding cross-correlation calculation unit 2-5, and a frame synchronization unit. 2-6, a symbol synchronizer 2-7, a frequency synchronizer 2-8, and a demodulator 2-9.

  FIG. 4 is a block diagram showing configurations of the frame synchronization unit 2-6, the symbol synchronization unit 2-7, and the frequency synchronization unit 2-8 in the radio reception station. In the figure, the frame synchronization unit 2-6 includes a peak determination unit 2-6-1 and a burst extraction unit 2-6-2, and the symbol synchronization unit 2-7 includes a symbol point extraction unit 2-7-. Further, the frequency synchronization unit 2-8 includes a phase calculation unit 2-8-1 and a frequency correction unit 2-8-2.

Next, a specific operation of each functional unit described above will be described along a series of sequences from reception of a transmission burst signal to demodulation. First, the radio reception station receives the transmission burst signal transmitted from the radio transmission station described above by the reception antenna 2-1, and then performs frequency conversion, analog / digital conversion (A) on the reception signal by the radio reception unit 2-2. / D conversion) to convert into a digital baseband signal. Next, the digital baseband signal is band-limited by the reception digital filter unit 2-3. Here, the output signal of the radio reception unit 2-2 at the time kT / N is expressed as r _{B. B} (kT / N), and the output signal of the reception digital filter unit 2-3 is r (kT / N). T is a symbol period, N is the number of oversampling, and k is an integer.

  Next, the differential decoding unit 2-4 differentially decodes r (kT / N) between adjacent symbols. That is, the following equation (2) is obtained.

Note that r _diff (kT / N) is a signal after differential decoding at time kT / N, and (A) ^* is a complex conjugate of complex number A.

Next, with respect to the differentially decoded signal r _diff (kT / N) by the sliding cross-correlation calculation unit 2-5, r _diff (kT / N) is set to the following equation (3) at an interval of 1 symbol. A cross-correlation operation is performed for each sample for the signal sequence having the sequence length N _a and the designated code sequence a _m (1 ≦ m ≦ N _a ).

  That is, the calculation shown in the following equation (4) is performed for each sample.

Note that R _cross (kT / N) is a cross-correlation value at time kT / N. Further, the cross-correlation value portion includes a differential decoded sequence r _diff (kT / N + (m−1) T) (1 ≦ m ≦ N _a ), the specified coded sequence a _m (1 ≦ m ≦ N). it may be normalized at each norm of _a).

Next, the peak determination unit 2-6-1 in the frame synchronization unit 2-6 accumulates the maximum value of the _cross- correlation value R _cross (kT / N) and the time kT / N, and for each determination, By comparing the maximum value with the cross-correlation value at the time of determination and updating the maximum value and its time as needed, the time at which the cross-correlation value becomes maximum during the determined determination period is the peak time. Is determined. In this determination method, a threshold value of the cross-correlation value is set, and a time when the cross-correlation value exceeds the threshold value is determined as a peak time. If the correlation value continues to decrease, there is a method of determining the time when the threshold value is exceeded as the peak time.

  Next, when the peak time is determined, the burst extraction unit 2-6-2 determines the peak time as the head position of the transmission burst signal, and extracts the transmission burst signal portion from the peak time. In this extraction method, if the burst extraction unit 2-6-2 requires a transmission burst signal part required for each part of the subsequent stage, for example, if each part of the subsequent stage does not require the training signal series, the training signal series A method of transferring only the information symbol sequence excluding the signal to each part of the subsequent stage, or the burst extracting unit 2-6-2 informs each part of the subsequent stage of the transmission burst signal part, and a necessary signal part from each part of the subsequent stage There is a method that allows only the processing to be performed.

  Next, the symbol point extraction unit 2-7-1 in the symbol synchronization unit 2-7 regards the peak time as a symbol point, and extracts only the symbol point from the output signal of the burst extraction unit 2-6-2. To do.

  Specifically, a symbol point represented by the following equation (5) is extracted from the signal r (kT / N) and transferred to the subsequent stage.

Note that k _data T / N is the head time of the signal portion necessary for the subsequent processing, and d is the total number of symbols of the transmission burst signal necessary for the subsequent processing.

Further, the cross-correlation value R _cross (k _peak T / N) of the peak time k _peak T / N calculated by the frame synchronization unit 2-6 by the phase calculation unit 2-8-1 in the frequency synchronization unit 2-8. ) Is calculated according to the following equation (6).

If the propagation channel does not change in the cross-correlation section, the phase component is the phase component of the cross-correlation value after differential decoding between adjacent signals of one symbol, that is, per symbol of phase rotation due to carrier frequency offset. Is equivalent to the phase rotation. Therefore, the frequency correction unit 2-8-2, using the phase component theta _1, with respect to the output signal from the symbol synchronization section 2-7 performs frequency correction according to the following equation (7).

Finally, the demodulator 2-9 demodulates the frequency-corrected signal r _AFC1 (k _data T / N + 1T) to reproduce the data bit sequence.

  Note that the processing order of the frequency synchronization unit 2-8 and the symbol synchronization unit 2-7 may be reversed, and the symbol r is extracted after performing the frequency correction on the signal r (kT / N). It doesn't matter. In this case, frequency correction is performed for each sample of the signal r (kT / N) as shown in the following equation (8).

Further, symbol point extraction is performed by using the frequency-corrected signal r _AFC1 (k _data T / N + (h−1) T / N), starting from k _data T / N, as shown in the following equation (9). In addition, it is extracted every symbol interval.

  In the first embodiment, since there is only one type of the training signal sequence as the overhead part of the transmission burst signal, overhead is required as compared with the first to fourth conventional systems described above, which requires a plurality of types of overhead. The part can be reduced. That is, the problem (1) of the conventional system can be solved.

Further, frame synchronization, symbol synchronization, and frequency synchronization are performed only from the peak time k _peak T / N of the _cross correlation value R _cross (kT / N) and its cross correlation value R _cross (k _peak T / N), that is, Since it is possible to process only from the cross-correlation calculation, each synchronization process requires a plurality of processes such as cross-correlation, auto-correlation, DFT, etc., and the circuit scale and power consumption of the receiving station increase. The system problem (2) can be solved.

  In addition, in frame synchronization, symbol synchronization, and frequency synchronization, the processing is completed by receiving the training signal sequence only once, so the problem (3) of the conventional system can be solved.

  Also, by performing the cross-correlation operation after differential decoding, even if the phase is rotated within the specified code sequence due to the carrier frequency offset, all the signals after differential decoding are the carrier frequency offset. Therefore, the phase of each term can be aligned at the time of product-sum at the time of cross-correlation calculation. Therefore, the problem (5) of the conventional system that deteriorates due to the carrier frequency offset at the time of the peak determination of the cross-correlation value can also be solved.

  In addition, a PN code can be used as the designated code sequence before differential encoding. Therefore, since the peak determination based on the cross-correlation value after differential decoding can be performed for the PN code, the problem (4) of the conventional system can be solved.

Also, the differentially decoded received signal is in the form of the square value of the propagation channel, that is, the propagation channel power multiplied by the designated code sequence before differential coding. The cross-correlation value is the product sum of the propagation channel power of each received signal. That is, a transmission burst signal sequence at time k _peak T / N + mT (m is an integer) is s (k _peak T / N + mT), a propagation channel is h (k _peak T / N + mT), and a propagation delay wave component and a noise signal are Considering an ignorable environment, the cross-correlation value is expressed by the following equation (10).

  Considering this differential decoding, the following equation (11) is obtained.

Propagation channel fluctuation hardly changes with one symbol. That is, the following equation (12), and also, the time _{k peak T / N + (m} -1) when T is considered a designated coding sequence _{a m} the same time, the following equation (13).

  Therefore, the cross-correlation value is expressed by the following equation (14).

Incidentally, the square value of the specified coding sequence a _m is 1 regardless of m.

From this, the contribution from each term of the _cross- correlation value R _cross (k _peak T / N) is proportional to the propagation channel power. Therefore, when the received signal level fluctuates due to fading fluctuation or the like in the training signal sequence, the cross-correlation value reflects the magnitude of the received signal level, thereby solving the problem (6) of the conventional system. be able to.

  As described above, the burst radio signal transmission system according to the first embodiment can solve all the problems (1) to (6) which are problems in the conventional burst radio signal transmission system.

<Second Embodiment>
Next, a second embodiment of the present invention will be described. The second embodiment is characterized in that the processing of the frequency synchronization unit 2-8 is performed before the digital filter unit 2-2, compared to the first embodiment described above.

  FIG. 5 is a block diagram showing a configuration of a radio receiving station of the burst radio signal transmission system according to the second embodiment. Note that the frame configuration of the burst radio communication transmission system and the radio transmission station are exactly the same as those in the first embodiment described above, so only the radio reception station will be described.

  The reception antenna 3-1, the wireless reception unit 3-2, the primary reception digital filter unit 3-3, the differential encoding unit 3-4, the sliding cross-correlation calculation unit 3-5, the frame synchronization unit 3-6, The frequency synchronization unit 3-7, the secondary reception digital filter unit 3-8, the symbol synchronization unit 3-9, and the demodulation unit 3-10 are the reception antenna 2-1 and the radio reception unit 2-2 of the first embodiment described above. The differential decoding unit 2-4, the sliding cross correlation calculation unit 2-5, the frame synchronization unit 2-6, the symbol synchronization unit 2-7, and the demodulation unit 2-9 have the same functions.

  The primary reception digital filter unit 3-3 and the secondary reception digital filter 3-8 have exactly the same function as the reception digital filter unit 2-3 of the first embodiment. Here, for the sake of convenience, the names of the primary reception digital filter unit and the secondary reception digital filter unit are given here for convenience. FIG. 6 is a block diagram illustrating configurations of the frame synchronization unit 3-6, the frequency synchronization unit 3-7, and the symbol synchronization unit 3-9 in the radio reception station according to the second embodiment.

Next, a specific operation of each functional unit will be described along a series of sequences from reception of the transmission burst signal described above to demodulation. First, the _cross- correlation value R _cross (kT) is obtained by the wireless reception unit 3-2 to the frame synchronization unit 3-6 by the same function as the wireless reception unit 2-2 to the frame synchronization unit 2-6 of the first embodiment described above. The _cross- correlation value R _cross (k _peak T / N) of the peak time k _peak T / N of / N) is calculated. In addition, the output signal r _{B. A} part related to the transmission burst signal part is extracted from _B (kT / N). Next, the signal r _B. Frequency correction is performed on _B (kT / N). That is, the frequency corrected signal is expressed as r _{B.B. If B_AFC1} (kT / N), the processing represented by the following equation (15) is performed.

Θ ₁ is a phase rotation per symbol by a carrier frequency offset calculated in the same manner as the frequency synchronization unit 3-7, and n is an integer. Further, k ′ _data T / N is the start time of the signal portion necessary for the subsequent processing, and d ′ is the number of symbols of the signal portion necessary for the subsequent processing.

Thereafter, the frequency-corrected signal rB _. After band limiting _{B_AFC1} (kT / N) by the secondary reception digital filter unit 3-8, the symbol synchronization unit 3-9 performs exactly the same processing as the symbol synchronization unit 2-6. Finally, the demodulator 3-10 performs exactly the same processing as the demodulator 2-8 on the output signal from the symbol synchronizer 3-9.

According to the second embodiment described above, frequency correction can be performed before the band limiting process of the reception digital filter unit. Therefore, since the position of the received signal spectrum whose center frequency is shifted due to the carrier frequency offset can be returned to the normal state before the band limiting process of the reception digital filter unit, the received signal whose center frequency is shifted due to the carrier frequency offset is reduced. Power consumption due to passing through the reception digital filter unit can be prevented.
Here, power reduction means that, when passing through a digital filter, if the center frequency of the signal deviates from the band of the digital filter, a reduction in power occurs in the output signal that has passed through the digital filter. .

<Third Embodiment>
Next, a third embodiment of the present invention will be described. The third embodiment is different from the second embodiment described above in that the radio receiving station is equipped with a plurality (N) of receiving systems, and has a space for processing of the frame synchronization unit, symbol synchronization unit, and frequency synchronization unit. It is characterized by performing diversity processing. Note that the frame configuration and radio transmission station of the burst radio signal transmission system are exactly the same as those in the first embodiment. Therefore, only a specific operation example of the radio receiving station will be described as the third embodiment.

  FIG. 7 is a block diagram showing a configuration of a radio receiving station of the burst radio signal transmission system according to the third embodiment. FIG. 7 shows a case where there are two reception systems (FIG. 3) of the first embodiment. Hereinafter, the third embodiment will be described by taking an example in which there are two reception systems (FIG. 3) of the first embodiment. Note that the operation of the third embodiment described below is exactly the same regardless of whether the receiving system is the above-described receiving system (FIG. 5) or more than two.

  The radio reception station of the third embodiment includes reception antennas 4-1-1, 4-1-2, radio reception units 4-2-1, 4-2-2, a reception digital filter unit 4-3-1, 4-3-2, differential decoding units 4-4-1, 4-4-2, sliding cross-correlation calculation unit 4-5, frame synchronization unit 4-6, symbol synchronization unit 4-7, frequency synchronization unit 4 -8 and demodulator 4-9.

  FIG. 8 is a block diagram showing configurations of the frame synchronization unit 4-6, the symbol synchronization unit 4-7, and the frequency synchronization unit 4-8 in the wireless reception station of the third embodiment. The frame synchronization unit 4-6 includes a peak determination unit 4-6-1, burst extraction units 4-6-2-1 and 4-6-2-2, and the symbol synchronization unit 4-7 includes a symbol It is composed of point extraction units 4-7-1-1 and 4-7-1-2, and a frequency synchronization unit 4-8 includes a phase calculation unit 4-8-1 and a frequency correction unit 4-8-2-. 1, 4-8-2-2.

  Note that the reception antennas 4-1-1, 4-1-2, the radio reception units 4-2-1, 4-2-2, the reception digital filter units 4-3-1, 4-3-2, and differential decoding. 4-4-1, 4-4-2, and demodulator 4-9 are the reception antenna 2-1, wireless reception unit 2-2, reception digital filter unit 2-3, and difference of the first embodiment described above. The functions of the moving decoding unit 2-4 and the demodulating unit 2-9 are exactly the same. Also, a peak determination unit 4-6-1, a burst extraction unit 4-6-2-1, 4-6-2-2, a symbol point extraction unit 4-7-1-1, 4-7-1-2, The phase calculation unit 4-8-1, the frequency correction units 4-8-2-1 and 4-8-2-2 are the peak determination unit 2-6-1 and the burst extraction unit 2- of the first embodiment described above. 6-2, the symbol point extraction unit 2-7-1, the phase calculation unit 2-8-1, and the frequency correction unit 2-8-2 have exactly the same functions.

Next, specific operations of the functional units of the third embodiment will be described along a series of sequences from reception of a transmission burst signal to demodulation. The reception signals received by the respective radio antennas 4-1-1 and 4-1-2 are respectively converted into reception digital baseband signals by the radio reception units 4-2-1 and 4-2-2. The output signals of the radio reception units 4-2-1 and 4-2-2 at the time kT / N are respectively represented by r ¹ _{B. B} (kT / N), r ² _B.I. Let _B (kT / N). Next, each signal r ¹ _{B. ^{B (kT / N), r}} 2 B. _B (kT / N) is band-limited by the reception digital filter units 4-3-1 and 4-3-2, respectively. The output signals of the reception digital filter units 4-3-1 and 4-3-2 at time kT / N are r ¹ (kT / N) and r ² (kT / N). Further, each signal r ¹ (kT / N), r ² (kT / N) is converted into a symbol interval by differential decoding units 4-4-1 and 4-4-2 as shown in the following equation (16). To perform differential decoding.

^{_{Incidentally, r 1 diff (kT / N}} ), r 2 diff (kT / N) is the output signal of each of the differential decoding unit 4-4-1,4-4-2 at time kT / N.

Next, with respect to each output signal of the differential decoding units 4-4-1 and 4-4-2, r ¹ _diff (kT / N) and r ² _diff (kT / N) are headed at intervals of one symbol. The signal sequence represented by the following equation (17) having the sequence length N _a is regarded as one signal sequence, and the signal sequence and the corrected code sequence a _m (1) are calculated by the sliding cross correlation calculation unit 4-4. ≦ m ≦ N _a ) is performed for each sample.

  That is, the calculation represented by the following equation (18) is performed.

Incidentally, the cross-correlation value obtained from the above equation (18) R _'cross the (kT / N) at time point kT / N. Further, the cross-correlation value is obtained by differentially decoding the signal sequence r ¹ _diff (kT / N + (m−1) T) (1 ≦ m ≦ N _a ), r ² _diff (kT / N + (m−1) ) T) (1 ≦ m ≦ N _a ), normalization with each norm of the specified coded sequence a _m (1 ≦ m ≦ N _a ). The differentially decoded signals r ¹ _diff (kT / N) and r ² _diff (kT / N) of each receiving system are the square weights of the signals of each receiving system, that is, the amplitude square value of the propagation channel. Since it is weighted, R ′ _cross (kT / N) corresponds to a value obtained by combining the maximum correlation of the cross-correlation values of the respective receiving systems. Therefore, R ′ _cross (kT / N) has a space diversity effect.

The cross-correlation value R _'cross (kT / N) and later was calculated operation, R' is regarded cross-correlation value _R cross and (kT / N) in the first embodiment the _cross (kT / N), first As in the embodiment, frame synchronization, symbol synchronization, and frequency synchronization are performed. In other words, it detects the peak time _{k peak} T / N where R _'cross (kT / N) is the peak by the peak determining unit 4-5-1 in the frame synchronization unit 4-5, the peak time _{k peak} T / N is regarded as the head portion of the transmission burst signal for the output signals r ¹ (kT / N) and r ² (kT / N) of the reception digital filter units 4-3-1 and 4-3-2, and the output signal r ^{For l} (kT / N) and r ² (kT / N), a burst burst signal portion is transmitted by burst extraction units 4-6-2-1 and 4-6-2-2 in the frame synchronization unit 4-6. Are extracted by the symbol point extraction units 4-7-1-1, 4-7-1-2 in the symbol synchronization unit 4-7. Further, the phase component of the _cross- correlation value R ′ _cross (kT / N) of the peak time k _peak T / N is used as the output signal r ^l (kT) of the reception digital filter units 4-3-1 and 4-3-2. / N) and r ² (kT / N), frequency correction is performed by regarding the amount of phase rotation per symbol due to the carrier frequency offset.

In other words, the phase calculation section 4-8-1 in the frequency synchronization unit 4-8 calculates the so theta _'1 as shown in the following equation (19), the frequency correction unit in the frequency synchronization unit 4-8 The frequency correction represented by the following equation (20) is performed on each of the signals r ¹ (kT / N) and r ² (kT / N) by 4-8-2-1 and 4-8-2-2. .

When there are two reception systems (FIG. 3) according to the second embodiment, 1 is obtained from the _cross- correlation value R ′ _cross (k _peak T / N) according to the burst start portion, symbol point, and carrier frequency offset as before. calculates the amount of phase rotation per symbol, the output signal ^r _{1 B.} radio receiving section 4-2-1,4-2-2 _B (kT / N), r ² _B.I. Against _B (kT / N), in the same order as in the second embodiment, the frame synchronization unit 4-6, a frequency synchronization unit 4-8, a receiving digital filter 4-3-1,4-3-2, The symbol synchronization unit 4-7 is processed.

Finally, the signal r ¹ _AFC1 (kT /) processed by the reception digital filter units 4-3-1 and 4-3-2, the frame synchronization unit 4-6, the symbol synchronization unit 4-7, and the frequency synchronization unit 4-8. _^N), with respect to _{r 2 AFC1 (kT / N)} , and demodulates the spatial diversity processing by the demodulation unit 4-9, to reproduce the data bit sequence. As for the _spatial diversity processing, selection diversity that selects only ^{one of the} _signals r ¹ _AFC1 (kT / N) and r ² _AFC1 (kT / N) having the larger amplitude, and the signal r ¹ _AFC1 (kT / N). ), R ² _AFC1 (kT / N) is detected, then equal gain combining is performed as it is, and after the _signals r ¹ _AFC1 (kT / N) and r ² _AFC1 (kT / N) are detected, the amplitude weight is set. Any of the maximum ratio combining to be combined may be used.

  According to the third embodiment described above, the frame of the first or second embodiment is used to estimate the frame synchronization point, the symbol synchronization point, and the carrier frequency offset by making the best use of the reception signals of each reception system. A spatial diversity effect can be obtained for each process of the synchronization unit 4-6, the symbol synchronization unit 4-7, and the frequency synchronization unit 4-8.

  Therefore, in the case of a configuration of a radio receiving station in which the frame synchronization point, the symbol synchronization point, and the carrier frequency offset of the signal received by each reception system match, for example, each reception antenna is provided in the vicinity, and each reception system When the reference frequency supplied for frequency conversion is common, that is, when the radio reception station is configured to realize spatial reception diversity, the reception level varies depending on the location of the reception antenna due to fading variation, etc. Improving the synchronization accuracy of the frame synchronization unit 4-6, the symbol synchronization unit 4-7, and the frequency synchronization unit 4-8 by providing a plurality of reception antennas and reception systems in the radio wave propagation environment. Can do.

<Fourth embodiment>
Next, a fourth embodiment of the present invention will be described.
Compared to the third embodiment described above, the fourth embodiment operates the frame synchronization unit and the symbol synchronization unit independently for each reception system, and performs spatial diversity processing only on the processing of the frequency synchronization unit. It is characterized by. Note that the frame configuration and radio transmission station of the burst radio signal transmission system are exactly the same as those in the first embodiment. Therefore, only the specific operation of the radio receiving station will be described as the fourth embodiment.

  FIG. 9 is a block diagram showing a configuration of a radio receiving station of the burst radio signal transmission system according to the fourth embodiment. FIG. 9 shows a case where there are two reception systems (FIG. 3) of the first embodiment described above. Hereinafter, the fourth embodiment will be described by taking an example in which there are two reception systems (FIG. 3) of the first embodiment. Note that the operation of the present invention described below is exactly the same regardless of whether the reception system is the above-described reception system (FIG. 5) of the second embodiment or three or more.

  The wireless reception station of the fourth embodiment includes reception antennas 5-1-1 and 5-1-2, wireless reception units 5-2-1 and 5-2-2, a reception digital filter unit 5-3-1, 5-3-2, differential decoding units 5-4-1, 5-4-2, sliding cross-correlation calculation units 5-5-1, 5-5-2, frame synchronization units 5-6-1, 5 -6-2, symbol synchronization units 5-7-1 and 5-7-2, frequency synchronization unit 5-8, and demodulation unit 5-9.

  Next, FIG. 10 is a block diagram showing a configuration of the frequency synchronization unit 5-8. The frequency synchronization unit 5-8 includes an addition unit 5-8-1, a phase calculation unit 5-8-2, and frequency correction units 5-8-3-1 and 5-8-3-2.

  Note that the receiving antennas 5-1-1, 5-1-2, radio receiving units 5-2-1, 5-2-2, receiving digital filter units 5-3-1, 5-3-3, differential decoding. Conversion units 5-4-1, 5-4-2, sliding correlation calculation units 5-5-1, 5-5-2, frame synchronization units 5-6-1, 5-6-2, symbol synchronization unit 5- 7-1, 5-7-2, and demodulator 5-9 are the reception antenna 2-1, wireless receiver 2-2, reception digital filter 2-3, differential decoder of the first embodiment described above. 2-4, the sliding correlation calculation unit 2-5, the frame synchronization unit 2-6, the symbol synchronization unit 2-7, and the demodulation unit 2-9 have the same functions.

  Next, specific operations of the respective functional units of the fourth embodiment will be described along a series of sequences from reception of a transmission burst signal to demodulation. In each reception system, the operations from the reception antenna 5-1-1 to the symbol synchronization unit 5-7-1 and from the reception antenna 5-1-2 to the symbol synchronization unit 5-7-2 are the same as those in the first embodiment described above. The operation is exactly the same. That is, each receiving system independently calculates a cross-correlation value, detects a peak time, and extracts a transmission burst signal and a symbol point.

Here, at time t, radio reception units 5-2-1, 5-2-2, reception digital filter units 5-3-1, 5-3-3, differential decoding units 5-4-1, Similarly to the notation in the third embodiment, r ¹ _{B. B} (t), r ² _{B.I. B} (t), r ¹ _diff (t), r ² _diff (t). In addition, the _cross- correlation values calculated by the sliding cross-correlation computing units 5-5-1 and 5-5-2 at time t are denoted as R ¹ _cross (t) and R ² _cross (t).

When the peak times detected by the peak determination units 5-6-1-1, 5-6-2-1 of the frame synchronization units 5-6-1, 5-6-2 are t ¹ _peak and t ² _peak , t the ¹ _peak, ^t _{2 peak,} as the first part of the transmission burst signal of each signal ^{r 1 (t), r 2} (t), the burst extraction unit 5-6-1-2,5-6-2-2, symbol The symbol parts of the signal part necessary for the subsequent processing in the signals r ¹ (t) and r ² (t) by the synchronizers 5-7-1 and 5-7-2 according to the following equation (21) Extract.

Note that t ¹ _data and t ² _data are head times of signal portions necessary for subsequent processing in the signals r ¹ (t) and r ² (t). Since each process from the radio reception unit to symbol synchronization operates independently in each reception system, t ¹ _peak ≠ t ² _peak and t ¹ _peak −t ² _peak ≠ bT (b is an integer) Therefore, the notation of each peak time is shown separately from t ¹ _peak and t ² _peak .

Next, both cross-correlation values R ¹ _cross (t ¹ _peak ) and R ² _cross (t ² _peak ) by the addition unit 5-8-1 in the frequency synchronization unit 5-8 are added according to the following equation (22). To do.

Note that R ″ _cross (t ¹ _cross , t ² _peak ) is differentially decoded that is used when calculating both cross-correlation values R ¹ _cross (t ¹ _peak ) and R ² _cross (t ² _peak ). Signal sequence r ¹ _diff (t ¹ _peak + (m−1) T) (1 ≦ m ≦ N _a ), r ² _diff (t ² _peak + (m−1) T) (1 ≦ m ≦ N _a ), Normalization may be performed with each norm of the specified coded sequence a _m (1 ≦ m ≦ N _a ). R ″ _cross (t ¹ _peak , t ² _peak ) calculated here is R ′ of the third embodiment. _This is an amount corresponding to _cross (k _peak / N). Subsequent operations are the same as those in the third embodiment. That is, the phase calculation unit 5-8-2 uses R ″ _cross (t ¹ _peak , t ² _peak ) to calculate the amount of phase rotation per symbol due to the carrier frequency offset, and the frequency correction unit 5-8-3. -1, 5-8-3-2 is used to calculate the signal r ¹ _BB (t), r ² _BB (t), or r ¹ (t), r ² (t In addition, similarly to the demodulator 4-9 of the third embodiment, the data bit sequence is reproduced using the signal whose frequency is corrected by the demodulator 5-9.

According to the fourth embodiment described above, the frame synchronization point and the symbol synchronization point are estimated independently by each receiving system, and only the carrier frequency offset is estimated by making the best use of both signals. A diversity effect can be obtained in the processing of the frequency synchronization unit of the embodiment.
Therefore, when the frame synchronization point and symbol synchronization point are different in each receiving system and only the carrier frequency offset is the same in each receiving system, for example, each receiving antenna is installed remotely, and the distance between the transmitting antenna and the receiving antenna is Although it differs depending on the antenna, if the reference frequency supplied to each receiving system for frequency conversion is common, that is, if the radio receiving station is configured to achieve receiving site diversity, the location of the receiving antenna may be affected by fading fluctuations, etc. In a radio wave propagation environment in which the reception level fluctuates due to the wireless reception station equipped with a plurality of reception antennas and reception systems, the synchronization accuracy of frequency synchronization can be improved.

<Fifth Embodiment>
Next, a fifth embodiment of the present invention will be described.
In the first or second embodiment described above, symbol synchronization is an operation of the symbol synchronization unit 2-7 that performs the peak time k _peak T / N as a symbol point, so that a symbol synchronization error is generated by 1 / 2N. In order to reduce the symbol synchronization error, the oversampling number N may be increased. However, the calculation amount of the reception digital filter unit and the sliding cross-correlation calculation unit, which has a large calculation amount compared to the other units, becomes the oversampling number. Since it is proportional, there is a problem that the circuit scale and power consumption of the radio receiving station increase.

  Therefore, the fifth embodiment is characterized in that the symbol synchronization error is reduced while the oversampling number N is maintained as compared with the first or second embodiment. Specifically, a second training signal for symbol synchronization is added to the subsequent stage of the training signal, and on the radio receiving station side, symbol synchronization using the second training signal is performed as an additional process of the symbol synchronization processing by the symbol synchronization unit. A second symbol synchronization unit for processing is added. The configuration of the second training signal and the operation of the second symbol synchronization unit are the same as those of the first conventional system.

First, the frame configuration of the fifth embodiment will be described.
FIG. 11 is a conceptual diagram showing the frame configuration of the fifth embodiment. In the fifth embodiment, a later-described second training signal sequence is inserted between the training signal sequence and the information symbol in the frame configuration (FIG. 1) of the first embodiment.

Next, the configuration of the wireless transmission station according to the fifth embodiment will be described.
FIG. 12 is a block diagram illustrating a configuration of a wireless transmission station according to the fifth embodiment. The radio transmission station has an information symbol conversion unit 6-1 and a training signal generation unit 6-2 before the multiplexing unit 6-4 with respect to the block configuration (FIG. 2) of the radio transmission station of the first embodiment. A second training signal generator 6-3 is added in parallel. The other information symbol conversion unit 6-1, training signal generation unit 6-2, transmission digital filter unit 6-5, wireless transmission unit 6-6, and transmission antenna 6-7 include the information symbol conversion unit 1- 1 shown in FIG. 1, the training signal generation unit 1-2, the transmission digital filter unit 1-4, the wireless transmission unit 1-5, and the transmission antenna 1-6 are exactly the same. Therefore, here, the detailed operation of each unit will be described only for the second training signal generation unit 6-3 and the multiplexing unit 6-4, which are differences from the first embodiment.

The second training signal generator 6-3 generates an alternating code that repeats +1 and -1 for each symbol. That is, if the alternating code is B _m ε {B ₁ , B ₂ , B ₃ ,..., B _Nb −1, B _Nb }, a signal sequence represented by the following equation (23) is generated.

Incidentally, N _b is the code length of the alternating code. For the sake of convenience, the signal sequence is hereinafter referred to as a second training signal sequence. The second training signal series only needs to be alternately arranged with +1 and -1, and may start with -1. The multiplexing unit 6-4 includes the information symbol sequence output from the information symbol conversion unit 6-1, the training signal sequence output from the training signal generation unit 6-2, and the second output from the second training generation unit 6-3. The training signal sequence is time-multiplexed to form a transmission burst signal shown in FIG. Hereinafter, the formed transmission burst signal is wirelessly transmitted to the space via the transmission digital filter unit 6-5, the wireless transmission unit 6-6, and the transmission antenna 6-7.

Next, the block configuration of the radio receiving station of the present invention will be described.
FIG. 13 is a block diagram illustrating a configuration of a wireless reception station according to the fifth embodiment. In the figure, in the fifth embodiment, a second symbol synchronization section 7-8 is added to the radio receiving station (FIG. 3) of the first embodiment. Specifically, the second symbol synchronization unit 7-8 is inserted between the frame synchronization unit 7-6 and the reception digital filter unit 7-3. In FIG. 13, the transmission antenna 7-1, the radio reception unit 7-2, the differential decoding unit 7-4, the sliding cross-correlation calculation unit 7-5, the frame synchronization unit 7-6, and the symbol synchronization unit 7-7. , Frequency synchronization unit 7-9, demodulation unit 7-10, transmission antenna 2-1, wireless reception unit 2-2, differential decoding unit 2-4, sliding cross-correlation calculation unit 2-5, frame of FIG. The synchronization unit 2-6, the symbol synchronization unit 2-7, the frequency synchronization unit 2-8, and the demodulation unit 2-9 are exactly the same. Therefore, in FIG. 13, the detailed operation of each unit will be described only for the reception digital filter unit 7-3 and the second symbol synchronization unit 7-8, which are differences from the first embodiment.

  FIG. 14 is a block diagram showing a configuration of a second symbol synchronization unit 7-8 to be described later. The second symbol synchronization unit 7-8 includes a second training signal sequence extraction unit 7-8-1, an amplitude square value calculation unit 7-8-2, a DFT unit 7-8-3, and a phase calculation unit 7-8-. 4 and a filter tap coefficient adjustment unit 7-8-5.

First, the detailed operation of the second symbol synchronization unit 7-8 will be described.
First, a portion corresponding to the second training signal sequence is extracted from the output signal from the frame synchronization unit 7-6 by the second training signal sequence extraction unit 7-8-1. The extracted signal is a signal obtained by subjecting the second training signal sequence _Bm to the band limitation of the transmission digital filter and the reception digital filter, and becomes a cosine wave or a sine wave with a period of 2T.

Therefore, after the extracted signal is squared by the amplitude square value calculation unit 7-8-2 and then subjected to DFT of period T by the DFT unit 7-8-3, the phase component of the output value is converted to the DFT unit 7 This corresponds to the leading clock phase θ _{clock_error} of the input signal of _−8-3 . That is, it corresponds to the clock phase θ _{clock_error} at the peak time k _peak T / N. Since the signal is extracted every symbol interval by the peak synchronization time k _peak T / N by the symbol synchronization unit 7-7, the clock phase θ _{clock_error} , that is, all the output signals of the symbol synchronization unit 7-7, , (Θ _{clock_error} / 2π) T symbol synchronization error has occurred.

By estimating the clock phase θ _{clock_error} from the phase component of the DFT value, this symbol synchronization error (θ _{clock_error} / 2π) T can be estimated. Accordingly, the phase component of the output value is calculated by the phase calculation unit 7-8-4 for the output value of the DFT unit 7-8-3, the clock phase θ _{clock_error} is estimated, and the symbol synchronization error (θ _{clock_error} / 2π) T is estimated.

Next, in order to correct the symbol synchronization error using the estimated symbol synchronization error (θ _{clock_error} / 2π) T value, the group delay of the reception digital filter unit 7-3 is set to (−θ _{clock_error} / 2π) T. Just shift. That is, the filter tap coefficient adjustment unit 7-8-5 adjusts the filter tap coefficient of the reception digital filter unit 7-3 so that the group delay of the reception digital filter unit 7-3 is shifted by (−θ _{clock_error} / 2π) T. . Then, the reception digital filter unit 7-3 changes the filter tap coefficient to the value adjusted by the filter tap coefficient adjustment unit 7-8-5. This change in filter tap coefficients, it is possible to correct the symbol synchronization error _(θ clock_error / 2π) T of the output signal of the symbol synchronization section 7-7.

  Note that the second symbol synchronization unit according to the fifth embodiment can also be applied to the second embodiment described above. FIG. 15 is a block diagram showing a configuration in which the second symbol synchronization unit of the fifth embodiment is added to the above-described radio reception station (FIG. 5) of the second embodiment. In the figure, a second symbol synchronization unit 8-8 is added between the frame synchronization unit 8-6 and the secondary reception digital filter unit 8-9. As in FIG. 14, the transmission antenna 8-1, the radio reception unit 8-2, the primary reception digital filter unit 8-3, the differential decoding unit 8-4, and the sliding cross-correlation calculation unit 8-in FIG. 5, frame synchronization unit 8-6, frequency synchronization unit 8-7, symbol synchronization unit 8-10, demodulation unit 8-11, transmission antenna 3-1, wireless reception unit 3-2, reception digital filter unit in FIG. 5 3-3, differential decoding unit 3-4, sliding cross-correlation calculation unit 3-5, frame synchronization unit 3-6, frequency synchronization unit 3-7, symbol synchronization unit 3-9, and demodulation unit 3-10 Are the same.

  The second symbol synchronization unit 8-8 is the same as the second symbol synchronization unit 7-8 in FIG. 13 and the secondary reception digital filter unit 8-9 is exactly the same as the reception digital filter unit 7-3 in FIG. is there. In other words, the filter tap coefficient adjusted by the second symbol synchronization unit 8-8 is reflected in the filter tap coefficient of the secondary reception digital filter unit 8-9, whereby the symbol synchronization of the output signal of the symbol synchronization unit 8-10 is performed. It becomes possible to correct the error. The amplitude square value calculation unit 7-8-2 squares the amplitude of the input signal in order to eliminate the influence of the phase rotation due to the carrier frequency offset of the input signal of the DFT unit 7-3. However, if the input signal is used as the output signal of the frequency synchronization unit 8-7, the phase rotation is deleted in advance, and thus the amplitude square value calculation unit 7-8-2 is not necessary.

If the number of points of the DFT unit 7-8-3 is 4 or more and a power of 2, the phase component of the output value matches the clock phase θ _{clock_error} . Therefore, in the first embodiment and the second embodiment, the oversampling number has to be increased in order to reduce the symbol synchronization error, whereas in the fifth embodiment, the oversampling number is increased. The symbol synchronization error can be reduced while maintaining 4 at 4.

  Therefore, according to the fifth embodiment, the insertion of the second training signal sequence increases the overhead as compared with the first and second embodiments, but the oversampling number is an arbitrary required value of the symbol synchronization error. Regardless of this, since it can always be maintained at 4, it is possible to prevent an increase in the amount of calculation of the reception digital filter unit and the sliding cross-correlation calculation unit, which has a large calculation amount compared to the other units and the calculation amount is proportional to the oversampling number. In addition, the circuit scale and power consumption of the radio receiving station can be reduced as compared with the first embodiment and the second embodiment.

<Sixth Embodiment>
Next, a sixth embodiment of the present invention will be described.
In the sixth embodiment, in contrast to the fifth embodiment described above, a plurality of reception systems are provided, and a frame synchronization unit 7-6, a symbol synchronization unit 7-7, a second symbol synchronization unit 7-8, a frequency It is characterized in that space diversity processing is performed on the processing of the synchronization unit 7-9. That is, in the sixth embodiment, diversity processing in the second symbol synchronization unit 7-8 is added to the third embodiment. The frame configuration and radio transmission station of the present invention are exactly the same as those in the fifth embodiment described above. Therefore, in the sixth embodiment, only a specific operation of the radio receiving station will be described.

  FIG. 16 is a block diagram illustrating a configuration of a wireless reception station according to the sixth embodiment. Note that the configuration of FIG. 16 shows a case where there are two reception systems configured as shown in FIG. Hereinafter, also in the sixth embodiment, a case where there are two reception systems in FIG. 13 will be described as an example. The operation of the sixth embodiment described below is exactly the same regardless of whether the number of reception systems shown in FIG. 13 is three or more and the number of reception systems shown in FIG. 15 is two or three or more. .

  In FIG. 16, transmitting antennas 9-1-1 and 9-1-2, radio receiving units 9-2-1 and 9-2-2, differential decoding units 9-4-1 and 9-4-2, The sliding cross-correlation calculation unit 9-5, the frame synchronization unit 9-6, the symbol synchronization unit 9-7, the frequency synchronization unit 9-9, and the demodulation unit 9-10 are the same as those of the wireless reception station (FIG. 7) of the third embodiment. Transmitting antenna 4-1-1, 4-1-2, radio receiving unit 4-2-1, 4-2-2, differential decoding unit 4-4-1, 4-4-2, sliding cross-correlation calculation The same as the unit 4-5, the frame synchronization unit 4-6, the symbol synchronization unit 4-7, the frequency synchronization unit 4-8, and the demodulation unit 4-10. The reception digital filter units 9-3-1 and 9-3-2 are the same as the reception digital unit 7-3 of the wireless reception station (FIG. 13) of the fifth embodiment. Therefore, in the following description, only the second symbol synchronization unit 9-8 is performed. The second symbol synchronization unit 9-8 is inserted between the frame synchronization unit 9-6 and the reception digital filter units 9-3-1 and 9-3-2.

  FIG. 17 is a block diagram showing a configuration of the second symbol synchronization section 9-8 according to the sixth embodiment. In the figure, a second symbol synchronization unit 9-8 includes second training signal sequence extraction units 9-8-1-1 and 9-8-1-2, an amplitude square value calculation unit 9-8-2-1, 9-8-2-2, an adding unit 9-8-3, a DFT unit 9-8-4, a phase calculating unit 9-8-5, and a filter tap coefficient adjusting unit 9-8-6. The second training signal sequence extraction units 9-8-1-1, 9-8-1-2, the amplitude square value calculation units 9-8-2-1 and 9-8-2-2, the DFT unit 9 -8-4, phase calculation unit 9-8-5, filter tap coefficient adjustment unit 9-8-6 are the second training signal sequence extraction unit 7-8- of the radio receiving station (FIG. 13) of the fifth embodiment. 1, the amplitude square value calculation unit 7-8-2, the DFT unit 7-8-3, the phase calculation unit 7-8-4, and the filter tap coefficient adjustment unit 7-8-5 are exactly the same.

  From the received signals of the respective receiving systems from the frame synchronization unit 9-6, the second training signal of each received signal is obtained by the second training signal sequence extracting units 9-8-1-1, 9-8-1-2. To extract. Next, an amplitude square value is calculated for each received signal from which the second training signal is extracted by the amplitude square value calculation units 9-8-2-1 and 9-8-2-2, and then the addition unit These are added by 9-8-3 and input to the DFT section 9-8-4.

  Thereafter, the phase calculation unit 9-8-5 calculates the phase of the output value of the DFT unit 9-8-4 by the same operation as that of the fifth embodiment, and the filter tap coefficient adjustment unit based on the phase value. 9-8-6 adjusts the filter tap coefficients of the reception digital filter units 9-3-1 and 9-3-2, and the reception digital filter units 9-3-1 and 9-3-2 adjust the filter tap coefficients. Change to filter tap coefficient. In addition, after performing DFT with respect to each output signal of amplitude square value calculating part 9-8-2-1 and 9-8-2-2, after adding with respect to the output value of each DFT, a phase component is obtained. You may ask for it. That is, the DFT unit 9-8-4 is connected to the subsequent stage of the amplitude square value calculating units 9-8-2-1 and 9-8-2-2, and the adding unit 9-8- is connected to the subsequent stage of both DFT units. The phase calculation unit 9-8-5 may be arranged after connecting the three.

  As described above, not only the spatial diversity processing for the processing of the frame synchronization unit, symbol synchronization unit, and frequency synchronization unit of the third embodiment, but also the processing of the second symbol synchronization unit of the fifth embodiment is performed. Can be performed. Therefore, as in the third embodiment, when the radio reception station has a spatial diversity configuration in which symbol synchronization is the same for the reception signals of each reception system, reception is performed depending on the location of the reception antenna due to fading fluctuations or the like. In a radio wave propagation environment where the level fluctuates, the radio receiving station is equipped with a plurality of receiving antennas and receiving systems, so that the accuracy of estimation of the symbol synchronization error in the processing of the second symbol synchronization unit is improved. It is possible to improve the synchronization accuracy.

<Seventh embodiment>
Next, a seventh embodiment of the present invention will be described.
In the seventh embodiment, the second symbol synchronization section 7-8 of the fifth embodiment is added to the fourth embodiment, and a plurality of reception systems are provided to the fifth embodiment, and the frame The synchronization units 5-6-1 and 5-6-2, the symbol synchronization units 5-7-1 and 5-7-2, and the second symbol synchronization unit 7-8 are independently connected to each other in frequency synchronization. It is characterized by performing spatial diversity processing on the processing. Note that the frame configuration and radio transmission station of the seventh embodiment are exactly the same as those of the fifth embodiment. Therefore, hereinafter, only the specific operation of the radio receiving station will be described as the seventh embodiment.

  FIG. 18 is a block diagram illustrating a configuration of a wireless reception station according to the seventh embodiment. Note that the configuration of FIG. 18 shows a case where there are two reception systems shown in FIG. Hereinafter, also in the seventh embodiment, a case where there are two reception systems illustrated in FIG. 13 will be described as an example. The operation of the seventh embodiment described below is exactly the same regardless of whether the number of reception systems shown in FIG. 13 is three or more and the number of reception systems shown in FIG. 15 is two or three or more. .

  In FIG. 18, transmitting antennas 10-1-1 and 10-1-2, radio receiving units 10-2-1 and 10-2-2, differential decoding units 10-4-1 and 10-4-2, Sliding cross-correlation calculation units 10-5-1, 10-5-2, frame synchronization units 10-6-1, 10-6-2, symbol synchronization units 10-7-1, 10-7-2, frequency synchronization units 10-9, the demodulator 10-10 are the transmission antennas 5-1-1, 5-1-2, the radio receivers 5-2-1, 5-2-2 of the fourth embodiment described above, and differential decoding. Conversion units 5-4-1, 5-4-2, sliding cross-correlation calculation units 5-5-1, 5-5-2, frame synchronization units 5-6-1, 5-6-2, symbol synchronization unit 5 −7-1, 5-7-2, frequency synchronization unit 5-8, demodulation unit 5-9 are exactly the same. The reception digital filter units 10-3-1 and 10-3-2 and the second symbol synchronization units 10-8-1 and 10-8-2 are the reception digital filter unit 7 of the fifth embodiment (FIG. 13). −3, the same as the second symbol synchronization unit 7-8.

  FIG. 19 is a block diagram illustrating a configuration of the frequency synchronization unit 10-9 according to the seventh embodiment. In the figure, the frequency synchronization unit 10-9 includes an addition unit 10-9-1, a phase calculation unit 10-9-2, and frequency correction units 10-9-3-1 and 10-9-3-2. Yes. These addition unit 10-9-1, phase calculation unit 10-9-2, frequency correction unit 10-9-3-1 and 10-9-3-2 are the frequency synchronization unit 5 of the fourth embodiment described above. −8 is the same as the adding unit 5-8-1, the phase calculating unit 5-8-2, and the frequency correcting units 5-8-3-1 and 5-8-3-2.

  According to the configuration described above, in the radio receiving station of the seventh embodiment, the frame synchronization units 10-6-1 and 10-6-2, the symbol synchronization units 10-7-1 and 10-7-2, The processing of the 2-symbol synchronization units 10-8-1 and 10-8-2 can be performed independently in each reception system, and the spatial diversity processing can be performed only for the processing of the frequency synchronization unit 10-9.

  According to the seventh embodiment described above, when the frame synchronization point and the symbol synchronization point are different in each reception system and only the carrier frequency offset is the same in each reception system, for example, each reception antenna is installed remotely, and the transmission antenna If the receiving antenna distance is different for each receiving antenna, but the reference frequency supplied to each receiving system for frequency conversion is the same, that is, if the radio receiving station is configured to achieve receiving site diversity, fading In a radio wave propagation environment in which the reception level varies depending on the location of the reception antenna due to fluctuations, etc., the radio reception station is equipped with a plurality of reception antennas and reception systems, so that the second symbol synchronization unit 7-of the fifth embodiment The synchronization accuracy of the frequency synchronization can be improved while maintaining the synchronization accuracy of the symbol synchronization by 8.

<Eighth Embodiment>
Next, an eighth embodiment of the present invention will be described.
The eighth embodiment is characterized in that the synchronization characteristics of frequency synchronization are further improved in the first embodiment, the second embodiment, and the fifth embodiment described above. Note that the frame configuration and radio transmission station of the eighth embodiment are exactly the same as those of the first embodiment, the second embodiment, or the fifth embodiment. In addition, the configuration of the radio receiving station is such that a second frequency synchronization unit is added before the demodulation unit with respect to the radio receiving station of the third embodiment, the fourth embodiment, the sixth embodiment, or the seventh embodiment. It becomes composition.

  FIG. 20 is a block diagram showing the second frequency synchronization unit added to the radio reception station in the eighth embodiment. In the eighth embodiment, the second frequency synchronization unit 11-1 is added after the frequency synchronization unit 2-8 to the radio reception station (FIG. 3) of the first embodiment. . The same parts as those of the wireless receiving station (FIG. 3) of the first embodiment are omitted.

  In the configuration of the radio receiving station of the eighth embodiment, the difference from the configuration of the radio station of the first embodiment is only the second frequency synchronization unit 11-1. Only the operation of the second frequency synchronization unit 11-1 will be described as a typical operation. Note that the same operation is performed when the second frequency synchronization unit 11-1 is added to the radio reception station of the second embodiment and the radio reception station of the fifth embodiment.

  FIG. 21 is a block diagram illustrating a configuration of the second frequency synchronization unit 11-1 of the eighth embodiment. In the figure, the second frequency synchronization unit 11-1 includes a training signal sequence extraction unit 11-1-1, a frequency correction unit 11-1-2 (for the training signal sequence), a training signal sequence generation unit 11-1-3, It comprises a division unit 11-1-4, an autocorrelation calculation unit 11-1-5, a phase calculation unit 11-1-6, and a frequency correction unit 11-1-7 (for an input signal to the demodulation unit).

The training signal sequence extraction unit 11-1-1 is represented by the following equation (24) based on the peak time k _peak T / N from the frame synchronization unit from the output signal r (kT / N) of the reception digital filter unit. The symbol points of the training signal sequence part are extracted.

The frequency correction unit 11-1-2 is, with respect to the extracted signal based on the carrier frequency offset theta ₁ which is estimated from the frequency synchronization unit (2-8 in Figure 3), the following equation (25) The carrier frequency offset represented by is corrected.

Next, the division unit 11-1-4 converts the output signal r _AFC1 (k _peak T / N + (m−1) T) of the frequency correction unit 11-1-2 into the training signal sequence generation unit 11-1-3. After being divided by the training signal sequence A _m (1 ≦ m ≦ N _a +1) generated by the autocorrelation operation unit 11-1-5, the signal obtained by the multiplication is expressed by the following equation (26). The autocorrelation calculation of the autocorrelation interval N _auto is performed.

_{Incidentally, N auto} is may be any integer that satisfies 2 ≦ _{N auto} ≦ _{N b.}

Next, the phase calculation unit 11-1-6 calculates a phase according to the following equation (27) with respect to the autocorrelation value R _auto .

If the propagation channel does not fluctuate in the autocorrelation section, the phase θ ₂ calculated here is the residual carrier frequency offset that could not be removed by the frequency correction based on the phase θ ₁ performed by the frequency correction unit 11-1-2. _Is the phase rotation for N _auto symbols. Thus, from the phase theta _2, the phase rotation of one symbol by the phase rotation due to the residual carrier frequency offset becomes θ ₂ / _{N auto.} Finally, the frequency correction unit 11-1-7 makes θ ₂ / N _{auto with} respect to the signal r _AFC1 (k _data T / N + (l−1) T) (1 ≦ l ≦ d) before the demodulation unit. Is used to correct the frequency according to the following equation (28).

  When the second frequency synchronization unit 11-1 is added to the second embodiment described above, it is added before the secondary reception digital filter unit 3-8, and the frequency by the second frequency synchronization unit 11-1. Correction may be performed before band limitation of the secondary reception digital filter unit 3-8, and the center position of the spectrum shifted to the carrier frequency offset may be further corrected by the second frequency synchronization unit 11-1 before band limitation. . Moreover, when adding the 2nd frequency synchronizer 11-1 with respect to 5th Embodiment mentioned above, you may utilize a 2nd training signal series for the autocorrelation calculation which the autocorrelation calculating part 11-1-5 performs.

  In general, in the estimation of the carrier frequency offset by autocorrelation, the smaller the autocorrelation interval, the wider the estimation range, but the estimation accuracy becomes worse.

According to the eighth embodiment, the phase rotation of one symbol is calculated as the carrier frequency offset, and the frequency synchronization unit calculates the differential decoding by the cross-correlation calculation, that is, the autocorrelation calculation of the autocorrelation interval of one symbol (estimated value θ ₁ ) In addition, the second frequency synchronization unit calculates (estimated value θ ₂ / N _auto ) by autocorrelation calculation of the autocorrelation interval N _auto symbol, so that the estimated value is widened by the estimated value θ ₁ and the estimated value It has improved the estimation accuracy by θ ₂ / _{N auto.} Therefore, first using only estimate theta _1, second, it is possible to improve the frequency synchronization of the synchronizing characteristics than either of the fifth embodiment.

<Ninth Embodiment>
Next, a ninth embodiment of the present invention will be described.
The ninth embodiment is characterized in that, with respect to the above-described eighth embodiment, the wireless reception station is equipped with a plurality of reception systems to perform spatial diversity processing on the second frequency synchronization processing. . Note that the frame configuration and radio transmission station configuration of the ninth embodiment are exactly the same as the frame configuration and radio transmission station of the third embodiment. In addition, the configuration of the radio receiving station of the ninth embodiment is the second before the demodulator with respect to the radio receiving station of the third embodiment, the fourth embodiment, the sixth embodiment, or the seventh embodiment. The frequency synchronization unit is added.

  FIG. 22 is a block diagram showing the second frequency synchronization unit added to the radio reception station in the ninth embodiment. In the ninth embodiment, the second frequency synchronization unit 12-1 is added to the radio reception station (FIG. 7) of the third embodiment when the number of reception systems is two. In the configuration of the radio receiving station of the ninth embodiment, the difference from the configuration of the radio receiving station of the third embodiment described above is only the second frequency synchronization unit 12-1. Only the operation of the second frequency synchronization unit 12-1 will be described as a typical operation. In addition, when there are three or more reception systems, and when the second frequency synchronization unit 12-1 is added to the radio reception station of the fourth embodiment, the sixth embodiment, or the seventh embodiment, it is completely The operation is similar.

  FIG. 23 is a block diagram showing a configuration of the second frequency periodic unit 12-1 according to the ninth embodiment. In the figure, the second frequency periodic unit 12-1 is configured to have two systems of the configuration of the second frequency synchronization unit 11 of the eighth embodiment described above. That is, the second frequency period unit 12-1 includes training signal sequence extraction units 12-1-1-1, 12-1-1-2, frequency correction units 12-1-2-1 (for the training signal sequence), 12-1-2-2, training signal sequence generation unit 12-1-3, division units 12-1-4-1, 12-1-4-2, autocorrelation calculation units 12-1-5-1, 12 -1-5-2, adder 12-1-4, phase calculator 12-1-7, frequency correctors 12-1-8-1, 12-1-8- (for the input signal to the demodulator) It consists of two.

  Training signal sequence extraction unit 12-1-1-2 to autocorrelation calculation unit 12-1-5-1, training signal sequence extraction unit 12-1-1-2 to autocorrelation calculation unit 12-1- The processing up to 5-2 performs exactly the same processing as the training signal sequence extraction unit 11-1-1-1 to the autocorrelation calculation unit 11-1-5 of the eighth embodiment described above.

  Next, the adding unit 12-1-4 adds the autocorrelation values calculated from the respective reception systems. The output values of the autocorrelation calculation units 12-1-5-1 and 12-1-5-2 are weighted by the square weight of the signal of each reception system, that is, the amplitude square value of the propagation channel of each reception system. Therefore, the autocorrelation value added by the adding unit 12-1-4 corresponds to a value obtained by combining the autocorrelation values of the respective reception systems with the maximum ratio. Therefore, the added autocorrelation value has a diversity effect. Thereafter, similarly to the eighth embodiment, the phase calculation unit 12-1-7 calculates the phase of the added autocorrelation value, and the frequency correction units 12-1-8-1 and 12-1-8-2. Thus, frequency correction is performed on the signal before the demodulator of each reception system or the signal before the primary reception digital filter.

  According to the ninth embodiment, it is possible to obtain a spatial diversity effect with respect to the second frequency synchronization. Therefore, in a radio wave propagation environment where the reception level fluctuates depending on the location of the receiving antenna due to fading fluctuations, etc., the radio receiving station is equipped with multiple receiving antennas and receiving systems, improving the synchronization accuracy of the second frequency synchronization can do.

<Tenth Embodiment>
Next, a tenth embodiment of the present invention will be described.
In the present tenth embodiment, when synchronous detection is performed as a modulation / demodulation method with respect to the first to ninth embodiments described above, propagation channel estimation for synchronous detection at a radio receiving station is performed using a training signal sequence or second training. It is characterized by using a signal sequence.

  The frame configuration and radio transmission station configuration of the tenth embodiment are exactly the same as those of the first to ninth embodiments described above, and each frame configuration and radio transmission station of the first to ninth embodiments described above. Any of these configurations may be used. The information symbol conversion unit of the wireless transmission station converts the data bit sequence into an information symbol sequence for synchronous detection.

  FIG. 24 is a block diagram showing a channel estimation unit and a synchronous detection unit added to the radio reception station in the tenth embodiment. In the figure, the radio receiving station of the tenth embodiment is different from the configurations of the radio receiving stations of the first to ninth embodiments described above in that the channel estimation unit 13-1 and the synchronous detection unit 13- 2 is added. The demodulation unit 13-3 is the same as the demodulation units 2-9, 3-10, 4-9, 5-9, 7-10, 8-11, 9-10, and 10-10. Note that the channel estimation unit 13-1 and the synchronous detection unit 13-2 of the tenth embodiment are added to the configurations of the radio receiving stations of the second to ninth embodiments other than the first embodiment. It is the same.

  Since the difference between the tenth embodiment and the first to ninth embodiments described above is only the channel estimation unit 13-1 and the synchronous detection unit 13-2, the channel estimation unit 13-1 and the synchronous detection unit 13 are included. Only a specific operation example of -2 will be described.

First, the channel estimation unit 13-1 will be described.
FIG. 25 is a block diagram showing the configuration of the channel estimation unit of the tenth embodiment. In the figure, a channel estimation unit 13-1 includes a training signal sequence extraction unit 13-1-1, a frequency correction unit 13-1-2, a training signal sequence generation unit 13-1-3, and a cross correlation calculation unit 13-1-. It is composed of four.

  The training signal sequence extraction unit 13-1-1, the frequency correction unit 13-1-2, and the training signal sequence generation unit 13-1-3 are the same as the training signal sequence extraction unit 11- described in the eighth embodiment. 1-1, the function is exactly the same as that of the frequency correction unit 11-1-2 and the training signal sequence generation unit 11-1-3. First, similarly to the eighth embodiment described above, the training signal sequence extraction unit 13-1-1 extracts the symbol points of the training signal sequence part or the second training signal sequence part, and the frequency correction unit 13-1-2. Thus, frequency correction is performed on the training signal sequence portion or the second training signal sequence portion. Thereafter, a cross-correlation calculation is performed between the frequency-corrected signal and the training signal sequence portion or the second training signal sequence portion generated by the training signal sequence generation unit 13-13.

Specifically, the following calculation is performed. As an example, the training signal sequence _Am will be described. Note that it is exactly the same when using the second training signal sequence B _m. Let the propagation channel be h. Note that the duration of the training signal series A _m, it is assumed that h does not change. The output signal of the frequency correction unit 13-1-2 at time _k peak / T + of the duration of the training signal series _{A m (m-1) T} (1 ≦ m ≦ N a +1), r training (k peak / T + (m−1) T) is expressed as the following equation (29).

However, frame synchronization, symbol synchronization, and frequency synchronization (frequency correction) were performed ideally, and noise signals were ignored. With respect to the signal _{r training (k peak / T +} (m-1) T), as shown in the following equation (30), performing a cross-correlation operation with the training signal series _{A m.}

Then, according to the following equation (31), by normalizing with the norm of the training signal series A _m, calculating the final propagation channel h.

The operation of the above, it is possible to estimate the propagation channel h by using a training signal series A _m. In the case where the propagation channel h it is assumed to vary during the said training signal sequence A _m are using only interval propagation channel h from the said training signal sequence A _m are assumed substantially constant Then, propagation channel estimation may be performed.

After estimating the propagation channel h, the synchronous detector 13-2 uses the signal r _AFC1 (k _data T / N + (l−1) T) (l = 1, 2,..., D) _{before the} demodulator 13-3. ), Synchronous detection is performed using the estimated propagation channel h ^. That is, the calculation is performed in accordance with the following equation (32), and is transferred to the demodulation unit 13-3.

  Normally, in the case of synchronous detection, a training signal for propagation channel estimation is required. However, according to the tenth embodiment, a training signal for frame synchronization, symbol synchronization, and frequency synchronization and a training signal for propagation channel estimation are combined. Can do. Therefore, in the first to ninth embodiments, synchronous detection can be realized without adding a further training signal.

<Eleventh embodiment>
Next, an eleventh embodiment of the present invention will be described.
In the eleventh embodiment, when synchronous detection is performed as a modulation / demodulation method with respect to the first to tenth embodiments described above, when the propagation channel fluctuates during the period of the transmission burst signal, the radio signal is transmitted after the training signal. By inserting a known pilot signal sequence between the transmitting station and the wireless receiving station, it is characterized by following propagation channel fluctuation at the time of propagation channel estimation.

  FIGS. 26A to 26C are conceptual diagrams showing a frame configuration of the burst radio signal transmission system according to the eleventh embodiment. FIG. 27 is a block diagram showing the configuration of the radio transmission station according to the eleventh embodiment. FIGS. 28A and 28B are the configuration of the radio reception station and channel estimation according to the eleventh embodiment. It is a block diagram which shows the structure of a part. 26 (a) to (c), FIG. 27, FIG. 28 (a), and (b), the eleventh embodiment is different from the frame configuration, the wireless transmission station, and the wireless reception station of the first embodiment described above. The case where embodiment is added is shown. The same applies to the case where the eleventh embodiment is added to the frame configurations, radio transmitting stations, and radio receiving stations of the second to tenth embodiments.

  Compared to the frame configurations of the first to tenth embodiments described above, the frame configuration is between the radio transmitting station and the radio receiving station after the training signal sequence (including the second training signal sequence), that is, after the information symbol sequence. A known pilot signal sequence is inserted at regular intervals. The pilot signal sequence may be anything as long as it is known between the radio transmission station and the radio reception station. Also, as an insertion method, as shown in FIG. 26 (a), before and after the information symbol series and at intervals in the information symbol series, or as shown in FIG. There are a method of inserting at a constant interval in the subsequent stage of the symbol sequence and the information symbol sequence, and a method of inserting at a constant interval only in the information symbol sequence as shown in FIG.

Note that in FIG 26, the pilot # 1, pilot # 2, ..., the pilot _#N p -1 in each pilot signal sequence portion of the pilot #N _p, respective pilot signal sequence in part may be in one symbol, Multiple symbols may be used. In addition, each pilot signal sequence portion may be the same signal sequence or different multiple sequences. In addition, each pilot monk sequence portion may have the same signal sequence length or a different signal sequence length.

  Next, as shown in FIG. 27, the radio transmission station has a configuration in which a pilot signal generation unit 14-3 is added to the preceding stage of the multiplexing unit 14-4 with respect to the first embodiment. Specifically, an information symbol signal sequence and a training signal sequence are generated from the information symbol conversion unit 14-1 and the training signal generation unit 14-2, and a pilot signal is generated by the pilot signal generation unit 14-3. Next, each signal sequence generated by the multiplexing unit 14-4 is time-multiplexed and then wirelessly transmitted to the space by the transmission digital filter unit 14-5, the wireless transmission unit 14-6, and the transmission antenna 14-7. .

  The information symbol conversion unit 14-1, the training signal generation unit 14-2, the transmission digital filter unit 14-5, the wireless transmission unit 14-6, and the transmission antenna 14-7 are the information described in the first embodiment. The symbol conversion unit 1-1, the training signal generation unit 1-2, the transmission digital filter unit 1-4, the wireless transmission unit 1-5, and the transmission antenna 1-6 are exactly the same.

  Next, as shown in FIG. 28A, the radio reception station adds a channel estimation unit 15-1 and a synchronous detection unit 15-2 to the radio reception station of the first embodiment before the demodulation unit. This is the configuration. Since the difference from the radio receiving station of the first embodiment described above is only the channel estimation unit 15-1 and the synchronous detection unit 15-2, as the eleventh embodiment, the channel estimation unit 16-1, Only a specific operation example of the detector 15-2 will be described.

The channel estimation unit 15-1 includes a pilot signal extraction unit 15-1-1 and a channel calculation unit 15-1-2. The pilot signal extraction unit 15-1-1 extracts a pilot signal sequence “Pilot #p (p = 1, 2,..., N _p )”. It is assumed that the pilot signal sequence has been subjected to frame synchronization, symbol synchronization, and frequency synchronization processing together with the information symbol sequence. Next, using the extracted pilot signal sequence “Pilot #p (p = 1, 2,..., N _p )”, the channel calculation unit 15-1-2 estimates the propagation channel by the following operation.

First, the propagation channel at the time of each pilot signal sequence “Pilot #p (p = 1, 2,..., N _p )” is the same as the propagation channel estimation using the training signal sequence described in the tenth embodiment. Calculated by Specifically, the propagation channel h _p ^ at each pilot #p is estimated by performing a cross-correlation operation on the extracted pilot signal sequence with the original pilot signal sequence within one pilot #p section. . Then, based on between each pilot #p, or between a training signal and a pilot #p, or the propagation channel at the time of the subsequent information symbol sequence of the pilot #N _p, propagation channel h _{p ^} a estimated in each pilot #p To estimate.

The estimation method is
(A) The propagation channel at the time of the information symbol signal sequence between the training signal and pilot # 1 is a propagation channel h ^ estimated from the training signal according to the tenth embodiment, and information between pilot #p and pilot # p + 1 the propagation channel at symbol signal sequence and propagation channel h _{p ^} and estimated by the pilot #p, also propagation channel h _Np where the propagation channel when subsequent information symbol signal sequence of pilot #N _p estimated by pilot #N _p A method of updating the propagation channel at any time is conceivable by setting ＾.

Also,
(B) A method of interpolating the propagation channel at the time of each information symbol sequence using the propagation channel h ^ estimated from the training signal and each propagation channel h _p ^ estimated from each pilot #p, for example, a training signal and the propagation channel at the time of the information symbol signal sequence between the pilot # 1, and ^ propagation channel h estimated from the training signal, and interpolating using a propagation channel h _{1 ^} estimated by pilot # 1, also, the pilot #p There is a method of interpolating the propagation channel at the time of the information symbol signal sequence between the pilot # p + 1 and the propagation channel h _p ^ estimated by the pilot #p and the propagation channel h _{p + 1} ^ estimated by the pilot # p + 1. I can think. Further, as interpolation methods, methods such as linear interpolation and spline interpolation can be considered.

Also,
(C) A method of extrapolating the propagation channel at each information symbol sequence using the propagation channel h ^ estimated from the training signal and each propagation channel h _p ^ estimated from each pilot #p, for example, pilot #p and Interpolating the propagation channel at the time of the information symbol signal sequence between pilot # p + 1 using propagation channel h _p-1 ^ estimated by pilot # p-1 and propagation channel h _p ^ estimated by pilot #p Can be considered.

  Further, as an extrapolation interpolation method, methods such as linear extrapolation interpolation and spline extrapolation interpolation are conceivable.

  Next, the synchronous detector 15-2 uses the estimated propagation channel at the time of each information symbol signal sequence, in the same manner as the synchronous detector 13-2 described in the tenth embodiment, and the demodulator 15- 3 is subjected to synchronous detection for the previous stage signal and transferred to the demodulator 15-3.

  According to the eleventh embodiment described above, when the propagation channel fluctuates within the transmission burst signal period, the propagation channel estimation can be made to follow the fluctuation. Therefore, in the first to tenth embodiments described above, overhead is increased due to the insertion of the pilot signal, but synchronous detection can be performed with high accuracy even when the propagation channel varies.

<Twelfth embodiment>
Next, a twelfth embodiment of the present invention will be described.
The present twelfth embodiment is characterized in that soft-decision error correction (Feedback Error Correct (FEC)) is performed when modulation / demodulation performs synchronous detection with respect to the tenth and eleventh embodiments described above. In particular, the main purpose is to reflect the estimated propagation channel amount to the likelihood calculation at the time of soft decision error correction. Note that the frame configuration of the twelfth embodiment is exactly the same as the frame configuration of the tenth and eleventh embodiments.

  FIG. 29 is a block diagram illustrating an error correction coding unit added to a wireless transmission station in the twelfth embodiment. FIG. 30A is a block diagram showing an error correction decoding unit added to the radio receiving station in the twelfth embodiment, and FIG. 30B is an error correction decoding added to the radio receiving station. It is a block diagram which shows the structure of a part.

  The radio transmission station of the twelfth embodiment is configured by adding an error correction encoding unit 16-1 before the information symbol conversion unit 14-1 to the radio transmission stations of the tenth and eleventh embodiments. It becomes. That is, after the error correction coding unit 16-1 performs error correction coding on the data bit sequence, the data bit sequence is delivered to the information symbol conversion unit 14-1. If necessary, an interleaving process may be performed. The subsequent processing is exactly the same as the processing of the radio transmission station of the tenth and eleventh embodiments.

  The radio receiving station according to the twelfth embodiment has a configuration in which an error correction decoding unit 17-1 is inserted in place of the demodulating unit with respect to the above-described non-reception receiving stations according to the tenth and eleventh embodiments. Therefore, only a specific operation example of the error correction decoding unit 17-1 will be described as an operation example of the radio reception station of the twelfth embodiment. The error correction decoding unit 17-1 includes a likelihood calculation unit 17-1-1 and a soft decision error correction decoding unit 17-1-2.

  The likelihood calculating unit 17-1-1 calculates the likelihood of the output signal of the synchronous detecting unit 13-2 using the propagation channel estimated by the channel estimating unit 13-1. Since the amplitude square value of the propagation channel is proportional to the SNR of the signal after synchronous detection, the likelihood can be calculated using the amplitude square value of the propagation channel. For example, in the case of BPSK modulation, a calculation method is conceivable in which the I component of the signal after synchronous detection is multiplied by the amplitude square value of the propagation channel. Next, the soft decision error correction decoding unit 17-1-2 performs soft decision error correction decoding using the calculated likelihood. When interleaving processing is performed on the wireless transmission station side, deinterleaving processing is performed before soft decision error correction decoding. As specific soft decision error correction decoding, soft decision Viterbi decoding, turbo decoding, LDPC decoding, and the like can be considered.

  According to the twelfth embodiment described above, soft decision error correction (FEC) can be performed when modulation / demodulation performs synchronous detection as compared with the tenth and eleventh embodiments, thereby improving transmission quality. It becomes possible.

<13th Embodiment>
Next, a third embodiment of the present invention will be described.
The thirteenth embodiment is characterized in that delayed detection is used as a modulation / demodulation method in contrast to the first to ninth embodiments described above. Note that the frame configuration of the thirteenth embodiment is exactly the same as the frame configuration of the first to ninth embodiments described above.

  FIG. 31 is a block diagram showing the differential encoding unit added to the wireless transmission station in the thirteenth embodiment. FIG. 32 is a block diagram showing the differential decoding unit added to the radio receiving station in the thirteenth embodiment. FIG. 31 and FIG. 32 both show the case where the thirteenth embodiment is added to the radio transmission station and radio reception station of the first embodiment. The same applies to the case where the thirteenth embodiment is added to the frame configuration, the wireless transmission station, and the wireless reception station of the second to ninth embodiments described above.

  The radio transmission station of the thirteenth embodiment inserts a differential encoding unit 18-1 between the information symbol conversion unit 1-1 and the multiplexing unit 1-3 with respect to the radio transmission station of the first embodiment. It becomes the composition which did. Specifically, the output signal from the information symbol conversion unit 1-1 is differentially encoded by the differential encoding unit 18-1, that is, the adjacent ones of the output signals are multiplied with each other, and then the multiplexing unit 1- Pass to 3. In addition, when the magnitude | size of the output signal from the information symbol conversion part 1-1 is not constant, you may take a complex conjugate only on one side instead of multiplication, and you may divide this. The subsequent operation is exactly the same as the operation of the wireless transmission station of the first embodiment.

  On the other hand, the radio receiving station of the thirteenth embodiment has a configuration in which a differential decoding unit 19-1 is inserted before the demodulating unit 2-9 with respect to the radio receiving station of the first embodiment. Since the difference from the radio receiving station of the first embodiment is only the differential decoding unit 19-1, only the differential decoding unit 19-1 is used as a specific operation example of the radio receiving station of the thirteenth embodiment. Only an operation example will be described. The differential decoding unit 19-1 performs differential decoding on the signal immediately before the demodulation unit 2-9. Specifically, the inverse calculation of the differential encoding unit 18-1 is performed. After differential decoding, the data is transferred to the demodulator 2-9. Even if the carrier frequency offset remains in the output signal of differential decoding section 19-1, the signal point error is at most about the phase rotation per symbol due to the carrier frequency offset, so the phase rotation is sufficiently small. When expected, the processing of the frequency synchronization unit 2-8 may not be performed.

  According to the thirteenth embodiment described above, it is possible to perform delayed detection with respect to the first to ninth embodiments described above. Therefore, the radio receiving station can demodulate with simpler processing than the synchronous detection that requires channel estimation.

<Fourteenth embodiment>
Next, a fourteenth embodiment of the present invention is described.
The fourteenth embodiment is characterized in that soft decision error correction is performed with respect to the thirteenth embodiment described above. Note that the frame configuration of the fourteenth embodiment is exactly the same as the frame configuration of the thirteenth embodiment.

  FIG. 33 is a block diagram showing a partial configuration of the wireless transmission station according to the fourteenth embodiment. FIG. 34 (a) is a block diagram showing the configuration of a part of the radio receiving station according to the fourteenth embodiment, and FIG. 34 (b) shows the configuration of the error correction decoding unit of the radio receiving station. It is a block diagram. FIG. 33, FIG. 34 (a), and FIG. 34 (b) both show the case where the fourteenth embodiment is added to the wireless transmission station and the wireless reception station of the first embodiment and the thirteenth embodiment. The same applies to the case where the fourteenth embodiment is added to the frame configuration, the wireless transmission station, and the wireless reception station of the second to ninth embodiments and the thirteenth embodiment.

  The radio transmission station according to the fourteenth embodiment has a configuration in which an error correction encoding unit 20-1 is inserted before the information symbol conversion unit 1-1 with respect to the radio transmission stations according to the first embodiment and the thirteenth embodiment. It becomes. Specifically, the error correction coding unit 20-1 performs error correction coding on the data bit sequence, and then transfers the data bit sequence to the information symbol conversion unit 1-1. The subsequent operation is exactly the same as the operation after the information symbol conversion unit of the thirteenth embodiment.

  On the other hand, the radio receiving station of the fourteenth embodiment has a configuration in which an error correction decoding unit 21-2 is inserted in place of the demodulating unit with respect to the radio receiving stations of the first embodiment and the thirteenth embodiment. . Therefore, only a specific operation example of the error correction decoding unit 21-2 will be described as an operation example of the radio reception station according to the fourteenth embodiment. The error correction decoding unit 21-2 includes a likelihood calculation unit 21-2-1 and a soft decision error correction decoding unit 21-2-2.

  The likelihood calculating unit 21-2-1 calculates the likelihood for the output signal of the differential decoding unit 2-4. Since the amplitude value of the output signal of the differential decoder 2-4 is proportional to the square of the amplitude of the propagation channel and proportional to the SNR of the output signal, the likelihood can be calculated using the output signal itself. It becomes possible. For example, in the case of BPSK modulation, a calculation method is conceivable in which the I component of the output signal of the differential decoding unit 2-4 is used as the likelihood. The subsequent operation is exactly the same as the operation of the soft decision error correction decoding unit 16-1 of the twelfth embodiment.

  According to the fourteenth embodiment described above, soft decision error correction can be performed with respect to the thirteenth embodiment described above, and transmission quality can be improved.

It is a conceptual diagram which shows the frame structure of the burst radio | wireless signal transmission system by this 1st Embodiment. It is a block diagram which shows the structure of the wireless transmission station by this 1st Embodiment. It is a block diagram which shows the structure of the radio | wireless receiving station by this 1st Embodiment. It is a block diagram which shows the structure of the frame synchronizing part in a radio | wireless receiving station, a symbol synchronizing part, and a frequency synchronizing part. It is a block diagram which shows the structure of the radio | wireless receiving station by this 2nd Embodiment. It is a block diagram which shows the structure of the frame synchronizing part in the radio | wireless receiving station of this 2nd Embodiment, a frequency synchronizing part, and a symbol synchronizing part. It is a block diagram which shows the structure of the radio | wireless receiving station of this 3rd Embodiment. It is a block diagram which shows the structure of the frame synchronizing part, symbol synchronizing part, and frequency synchronizing part in the radio | wireless receiving station of this 3rd Embodiment. It is a block diagram which shows the structure of the radio | wireless receiving station by this 4th Embodiment. It is a block diagram which shows the structure of the frequency synchronization part by this 4th Embodiment. It is a conceptual diagram which shows the frame structure of this 5th Embodiment. It is a block diagram which shows the structure of the wireless transmission station of this 5th Embodiment. It is a block diagram which shows the structure of the radio | wireless receiving station of this 5th Embodiment. It is a block diagram which shows the structure of the 2nd symbol synchronization part by this 5th Embodiment. It is a block diagram which shows the structure which added the 2nd symbol synchronization part of the 5th embodiment. It is a block diagram which shows the structure of the radio | wireless receiving station of this 6th Embodiment. It is a block diagram which shows the structure of the 2nd symbol synchronization part of this 6th Embodiment. It is a block diagram which shows the structure of the radio | wireless receiving station of this 7th Embodiment. It is a block diagram which shows the structure of the frequency synchronizer of this 7th Embodiment. It is a block diagram which shows the 2nd frequency synchronization part added to the radio | wireless receiving station in this 8th Embodiment. It is a block which shows the structure of the 2nd frequency synchronization part of this 8th Embodiment. It is a block diagram which shows the 2nd frequency synchronization part added to the radio | wireless receiving station in this 9th Embodiment. It is a block diagram which shows the structure of the 2nd frequency period part by this 9th Embodiment. It is a block diagram which shows the channel estimation part added to the radio | wireless receiving station in this 10th Embodiment, and a synchronous detection part. It is a block diagram which shows the structure of the channel estimation part of this 10th Embodiment. It is a conceptual diagram which shows the frame structure of this 11th Embodiment. It is a block diagram which shows the structure of the wireless transmission station by this 11th Embodiment. It is a block diagram which shows the structure of the radio | wireless receiving station by this 11th Embodiment, and the structure of a channel estimation part. It is a block diagram which shows the error correction encoding part added to the wireless transmission station in this 12th Embodiment. It is a block diagram which shows the structure of the error correction decoding part added to the radio | wireless receiving station in this 12th Embodiment, and an error correction decoding part. It is a block diagram which shows the differential encoding part added to the wireless transmission station in this 13th Embodiment. It is a block diagram which shows the differential decoding part added to the radio | wireless receiving station in this 13th Embodiment. It is a block diagram which shows the structure of a part of radio transmission station by this 14th Embodiment. It is a block diagram which shows the structure of a part of radio | wireless receiving station by this 14th Embodiment, and the structure of the error correction decoding part of a radio | wireless receiving station. It is a conceptual diagram which shows the frame structure of a 1st conventional system. It is a block diagram which shows the function structure of the reception synchronous process of a 1st conventional system. It is a conceptual diagram which shows the frame structure of a 2nd conventional system. It is a functional block diagram for demonstrating the reception synchronous process of a 2nd conventional system. It is a conceptual diagram which shows the frame structure of a 3rd conventional system. It is a functional block diagram for demonstrating the reception synchronous process of a 3rd conventional system. It is a conceptual diagram which shows the frame structure of a 4th conventional system. It is a functional block for demonstrating the reception synchronous process of a 4th conventional system.

Explanation of symbols

1-1, 6-1 and 20-2 Information symbol conversion unit (information symbol conversion means, synchronous detection conversion means)
1-2, 6-2 Training signal generator (training signal generator)
1-3, 6-4 Multiplexer (multiplexer)
1-5 Wireless transmission unit (wireless transmission means)
1-6 Transmitting antenna 2-1, 3-1 Receiving antenna 2-2, 3-2 Radio receiving unit (radio receiving means)
2-3, 3-3, 3-8 Reception digital filter section (reception digital filter means)
2-4, 3-4 differential encoding unit (differential decoding means)
2-5, 3-5 Sliding cross-correlation calculation unit (sliding cross-correlation calculation means)
2-6, 3-6 Frame synchronization unit (frame synchronization means)
2-7, 3-9 Symbol synchronization section (symbol synchronization means)
2-8, 3-7 Frequency synchronization unit (frequency synchronization means)
2-9, 3-10 Demodulator (demodulator)
4-1-1, 4-1-2 Receiving antenna 4-2-1, 4-2-2 Wireless receiving unit (wireless receiving means)
4-3-1, 4-3-2 Reception digital filter section (reception digital filter means)
4-4-1, 4-4-2 Differential encoding unit (differential decoding means)
4-5 Sliding cross-correlation calculation unit (sliding cross-correlation calculation means)
4-6 Frame synchronization unit (frame synchronization means)
4-7 Symbol synchronization unit (symbol synchronization means)
4-8 Frequency synchronization unit (frequency synchronization means)
4-9 Demodulator (Demodulator)
5-1-1, 5-1-2 Receiving antenna 5-2-1, 5-2-2 Wireless receiving unit (wireless receiving means)
5-3-1, 5-3-2 Reception digital filter section (Reception digital filter means)
5-4-1, 5-4-2 Differential encoding unit (differential decoding means)
5-5-1, 5-5-2 Sliding cross-correlation calculation unit (sliding cross-correlation calculation means)
5-6-1, 5-6-2 Frame synchronization unit (frame synchronization means)
5-7-1, 5-7-2 Symbol synchronization section (symbol synchronization means)
5-8 Frequency synchronization unit (frequency synchronization means)
5-9 Demodulator (Demodulator)
6-3 Second training signal generator (second training signal generator)
7-1, 8-1 Receiving antenna 7-2 Wireless receiver (wireless receiving means)
7-3, 8-3, 8-9 Reception digital filter section (reception digital filter means)
7-4, 8-4 Differential encoding unit (differential decoding means)
7-5, 8-5 Sliding cross-correlation calculation unit (sliding cross-correlation calculation means)
7-6, 8-6 Frame synchronization unit (frame synchronization means)
7-7, 8-10 Symbol synchronization section (symbol synchronization means)
7-8, 8-8, 9-8 Second symbol synchronization section (second symbol synchronization means)
7-9, 8-7 Frequency synchronization unit (frequency synchronization means)
7-10, 8-11 Demodulator (demodulator)
9-1-1, 9-1-2 Receiving antenna 9-2-1, 9-2-2 Wireless receiving unit (wireless receiving means)
9-3-1, 9-3-2 Reception digital filter section (reception digital filter means)
9-4-1, 9-4-2 Differential encoding unit (differential decoding means)
9-5 Sliding cross-correlation calculation unit (sliding cross-correlation calculation means)
9-6 Frame synchronization unit (frame synchronization means)
9-7 Symbol synchronization unit (symbol synchronization means)
9-8 Second symbol synchronization section (second symbol synchronization means)
9-9 Frequency synchronization unit (frequency synchronization means)
9-10 Demodulator (demodulator)
10-1-1, 10-1-2 receiving antenna 10-2-1, 10-2-2 wireless receiving unit (wireless receiving means)
10-3-1, 10-3-2 Reception digital filter section (Reception digital filter means)
10-4-1, 10-4-2 Differential encoding unit (differential decoding means)
10-5-1, 10-5-1 Sliding cross-correlation calculation unit (sliding cross-correlation calculation means)
10-6-1, 10-6-2 Frame synchronization unit (frame synchronization means)
10-7-1, 10-7-2 Symbol synchronization section (symbol synchronization means)
10-8-1, 10-8-2 Second symbol synchronization section (second symbol synchronization means)
10-9 Frequency synchronization unit (frequency synchronization means)
10-10 Demodulator (demodulator)
11-1, 12-1 Second frequency synchronization section (frequency synchronization means)
13-1, 15-1 Channel estimation unit (channel estimation means)
13-2, 15-2 Synchronous detection unit (synchronous detection means)
14-3 Pilot signal generation unit (pilot signal sequence generation means)
16-1, 20-1 Error correction coding unit (error correction coding means)
17-1, 21-2 Error correction decoding unit (error correction decoding means)
18-1, 20-3 Differential encoding unit (conversion means for delay detection)
19-1, 21-1 Differential decoding unit (delay detection means)

Claims (26)

In a burst radio signal transmission system composed of a radio transmitter station and a radio receiver station,
The radio transmitting station is
Information symbol conversion means for converting an input data bit sequence into an information symbol sequence;
Training signal generation means for differentially encoding a designated code sequence that is a pre-designated code sequence to generate a training signal sequence;
Multiplexing means for time-multiplexing the information symbol sequence and the training signal sequence to form a transmission burst signal;
Wireless transmission means for converting the transmission burst signal into a wireless signal;
A transmitting antenna for wirelessly transmitting the wireless signal in space,
The radio receiving station is
A receiving antenna for receiving radio signals;
Wireless receiving means for converting the wireless signal into a digital baseband signal;
A receiving digital filter means for performing band limitation on the input signal;
Differential decoding means for differentially decoding the input digital signal for each sample at symbol intervals;
A sliding cross-correlation calculating unit that performs a sliding cross-correlation calculation for each of the designated code sequence and the sample with respect to the signal sequence output by the differential decoding unit;
A peak determination is performed on the cross-correlation value of each sample output by the sliding cross-correlation calculating means, and the transmission burst portion is determined with a signal position corresponding to the cross-correlation value determined to be a peak as the head position of the transmission burst signal Frame synchronization means for extracting
Symbol synchronization means for extracting the symbol point from the input signal, using the signal position determined as the peak as a symbol point;
A frequency synchronization unit that obtains a phase component of the cross-correlation value determined to be the peak, regards the phase component as a phase rotation per symbol by a carrier frequency offset, and performs the carrier frequency offset correction on the input signal;
Demodulating means for demodulating the signal whose offset has been corrected by the frequency synchronization means to reproduce a data bit sequence.   The burst radio signal transmission system according to claim 1, wherein a PN code is used as the designated code sequence. The radio receiving station is
The information symbol sequence output from the wireless reception means is processed in the order of the frame synchronization means processing, the frequency synchronization means processing, the reception digital filter means processing, and the symbol synchronization means processing. The burst radio signal transmission system according to claim 1. The radio receiving station is
N receiving antennas (N is an integer),
N radio reception means, reception digital filter means, and differential decoding means,
The N reception signals received by the N reception antennas are processed by the wireless reception means, the reception digital filter means, and the differential decoding means,
The sliding cross-correlation calculating means regards the output signal sequence of each of the N differential decoding means as one signal sequence, performs a sliding cross-correlation calculation with N of the designated code sequences for each sample,
The frame synchronization means performs peak determination using the cross-correlation value of each sample output by the sliding cross-correlation calculation means, and a signal position corresponding to the cross-correlation value determined to be a peak is used as a head position of the transmission burst signal. Extracting the transmission burst signal portion for N input signals,
The symbol synchronization means uses the signal position determined to be the peak as a symbol point, and extracts only the symbol point from N input signals,
The frequency synchronization means obtains a phase component of the cross-correlation value determined to be the peak, regards the phase component as a phase rotation per symbol by a carrier frequency offset, and each of the N input signals Perform carrier frequency offset correction,
4. The demodulating unit reproduces an original data bit sequence by performing diversity combining on the N received signals offset-corrected by the frequency synchronizing unit. The burst radio signal transmission system described. The radio receiving station is
N receiving antennas (N is an integer),
The radio reception means, the reception digital filter means, the differential decoding means, the sliding cross correlation calculation means, the frame synchronization means, and N symbol synchronization means,
For N received signals received by the N receiving antennas, the radio receiving means, the received digital filter means, the differential decoding means, the sliding correlation calculating means, the frame synchronization means, the symbol synchronization Process each means independently,
The frequency synchronization means adds all the cross-correlation values determined as N peaks calculated from each of the N frame synchronization means, calculates a phase component of the added cross-correlation value, and calculates the calculated phase component Is regarded as a phase rotation per symbol by a carrier frequency offset, and the carrier frequency offset correction is performed on N received signals,
The demodulating means performs diversity combining on the N received signals subjected to the processing of the reception digital filter means, the frame synchronization means, the symbol synchronization means, and the frequency synchronization means to perform original synthesis. 4. The burst radio signal transmission system according to claim 1, wherein a bit sequence is reproduced. The radio transmitting station is
Comprising second training signal generating means for generating an alternating code that repeats +1 and −1 in two symbol periods as a second training signal sequence;
The multiplexing means time-multiplexes the information symbol sequence, the training signal sequence, and the second training signal sequence to form a transmission burst signal,
The radio receiving station is
With respect to the second training signal sequence portion of the output signal of the frame synchronization means, the square value of the amplitude is subjected to a discrete Fourier transform process, the phase component of the discrete Fourier transform value is regarded as a symbol phase error, and the symbol Adjusting the tap coefficient of the reception digital filter corresponding to the phase error to further comprise second symbol synchronization means for performing further symbol phase error compensation on the output signal from the symbol synchronization means; The burst radio signal transmission system according to any one of claims 1 to 3. The radio transmitting station is
Comprising second training signal generating means for generating an alternating code that repeats +1 and −1 in two symbol periods as a second training signal sequence;
The multiplexing means time-multiplexes the information symbol sequence, the training signal sequence, and the second training signal sequence to form a transmission burst signal,
The radio receiving station is
The second training signal sequence portion is extracted for each of the N output signals of the frame synchronization means, and the same time signal of the N second training signal sequences is squared in amplitude and added. The added signal series is subjected to discrete Fourier transform processing, the phase component of the discrete Fourier transform value is regarded as a symbol phase error, and tap coefficients of the N reception digital filters corresponding to the symbol phase error 5. The burst radio signal according to claim 4, further comprising second symbol synchronization means for performing further symbol phase error compensation on N output signals from said symbol synchronization means by adjusting Transmission system. The radio transmitting station is
Comprising second training signal generating means for generating an alternating code that repeats +1 and −1 in two symbol periods as a second training signal sequence;
The multiplexing means time-multiplexes the information symbol sequence, the training signal sequence, and the second training signal sequence to form a transmission burst signal,
The radio receiving station is
The second training signal sequence portion is extracted for each of the output signals of the N frame synchronization means, and the same time signals of the N second training signal sequences are squared in amplitude and added. The added signal series is subjected to discrete Fourier transform processing, the phase component of the discrete Fourier transform value is regarded as a symbol phase error, and tap coefficients of the N reception digital filters corresponding to the symbol phase error 6. The burst radio signal according to claim 5, further comprising second symbol synchronization means for performing further symbol phase error compensation on N output signals from said symbol synchronization means by adjusting Transmission system. The radio receiving station is
The training signal sequence or the second training for each symbol with respect to the training signal sequence portion or the second training signal sequence after the processing by the frame synchronization unit, the symbol synchronization unit, and the frequency synchronization unit is performed. After aligning the phase of each symbol of the training signal sequence portion or the second training signal sequence portion by multiplying the signal sequence, the training signal sequence or the second training signal sequence having the aligned phases, After performing autocorrelation calculation with a correlation interval of 2 symbols or more and estimating the carrier frequency offset from the phase component of the autocorrelation calculation value, the estimated carrier frequency offset is used for the signal preceding the demodulation means Second frequency synchronization means for performing carrier frequency offset compensation Burst radio signal transmission system according to any one of claims 1 to 3 or 6, characterized in that it further comprises. The radio receiving station is
For each of the N training signal sequence portions or the second training signal sequence after the processing by the frame synchronization means, the symbol synchronization means, and the frequency synchronization means, the training signal sequence or The training signal sequence or the second training signal sequence in which the phases are aligned after the phases of the symbols of the training signal sequence portion or the second training signal sequence portion are aligned by multiplying the second training signal sequence. Then, after performing an autocorrelation calculation with a correlation interval of 2 symbols or more, adding the N autocorrelation calculation values, estimating a carrier frequency offset from the phase component of the added autocorrelation calculation values, The estimated carrier frequency offset is used for N signals before the demodulation means. Burst radio signal transmission system according to any of claims 4, 5, 7 or 8, characterized by comprising further a second frequency synchronization to perform carrier frequency offset compensation Te. The radio transmitting station is
As the information symbol conversion means, comprising: synchronous detection conversion means for converting the data bit sequence into an information symbol sequence for synchronous detection on the assumption that synchronous detection is performed at the radio receiving station,
The radio receiving station is
Channel estimation means for estimating the number of propagation channels using the training signal sequence or the second training signal sequence after the frame synchronization, the symbol synchronization, and the frequency synchronization have been established;
2. The apparatus according to claim 1, further comprising the same number of synchronous detection means as that of the receiving antenna, wherein the output signal from the frequency synchronization means is synchronously detected using the propagation channel estimated by the channel estimation means. The burst radio signal transmission system according to any one of 10. The radio transmitting station is
As the information symbol conversion means, synchronous detection conversion means for converting the data bit sequence into an information symbol sequence for synchronous detection on the assumption that synchronous detection is performed at the radio receiving station;
Pilot signal sequence generating means for generating a pilot signal sequence designated in advance,
The multiplexing means inserts the pilot signal sequence at regular intervals into the information symbol sequence,
The radio receiving station is
Channel estimation means that estimates the radio propagation channel from the pilot signal sequence after frame synchronization, symbol synchronization, and frequency synchronization is established;
2. The apparatus according to claim 1, further comprising: the same number of synchronous detection means as that of the receiving antenna, wherein the output signal from the frequency synchronization means is synchronously detected using the radio propagation channel estimated by the channel estimation means. The burst radio signal transmission system according to any one of 1 to 10. The channel estimation means includes
13. The burst radio signal transmission system according to claim 12, wherein an estimated value of a propagation channel is updated for each pilot signal. The channel estimation means includes
13. The burst radio signal transmission system according to claim 12, wherein a radio propagation channel between the pilot signals is interpolated from an estimated value of the radio channel estimated from the pilot signal. The channel estimation means includes
14. The burst radio multiple transmission system according to claim 13, wherein a radio propagation channel between the pilot signals or after the pilot signal is extrapolated from an estimated value of the propagation channel estimated from the pilot signal. The radio transmitting station is
An error correction encoding means for error correcting encoding the data bit sequence before the information symbol conversion means;
The radio receiving station is
As an alternative to the demodulating means, a soft decision error is made by using, as a likelihood weight, the amplitude square value of the propagation channel estimated by the channel estimating means for the output signal of the synchronous detecting means after the synchronous detecting means. 16. The burst radio signal transmission system according to claim 11, further comprising error correction decoding means for performing correction decoding. The radio transmitting station is
The information symbol conversion means further comprises delay detection conversion means for converting the data bit sequence into an information symbol sequence for delay detection on the assumption that the radio reception station performs delay detection,
The radio receiving station is
11. The burst radio signal transmission system according to claim 1, further comprising delay detection means for performing differential decoding on an input signal of the demodulation means before the demodulation means. . The radio transmitting station is
An error correction encoding means for error correcting encoding the data bit sequence before the information symbol conversion means;
The radio receiving station is
The error correction decoding means for performing soft decision error correction decoding on the output signal of the delay detection means using the amplitude value as a likelihood weight as an alternative to the demodulation means. 18. The burst radio signal transmission system according to 17. In a wireless transmission device that wirelessly transmits a burst signal,
Information symbol conversion means for converting an input data bit sequence into an information symbol sequence;
Training signal generation means for differentially encoding a designated code sequence that is a pre-designated code sequence to generate a training signal sequence;
Multiplexing means for time-multiplexing the information symbol sequence and the training signal sequence to form a transmission burst signal;
Wireless transmission means for converting the transmission burst signal into a wireless signal;
A wireless transmission apparatus comprising: a transmission antenna that wirelessly transmits the wireless signal in space. In a wireless receiver that receives a burst signal transmitted wirelessly,
A receiving antenna for receiving radio signals;
Wireless receiving means for converting the wireless signal into a digital baseband signal;
A receiving digital filter means for performing band limitation on the input signal;
Differential decoding means for differentially decoding the input digital signal for each sample at symbol intervals;
A sliding cross-correlation calculating means for performing a sliding cross-correlation calculation for each sample and a designated code sequence that is a pre-specified code sequence for the signal sequence output by the differential decoding means;
A peak determination is performed on the cross-correlation value of each sample output by the sliding cross-correlation calculating means, and the transmission burst portion is determined with a signal position corresponding to the cross-correlation value determined to be a peak as the head position of the transmission burst signal Frame synchronization means for extracting
Symbol synchronization means for extracting the symbol point from the input signal, using the signal position determined as the peak as a symbol point;
A frequency synchronization unit that obtains a phase component of the cross-correlation value determined to be the peak, regards the phase component as a phase rotation per symbol by a carrier frequency offset, and performs the carrier frequency offset correction on the input signal;
A radio receiving apparatus comprising: demodulating means for demodulating the signal whose offset has been corrected by the frequency synchronizing means to reproduce a data bit sequence. In a burst wireless signal transmission method for wirelessly transmitting a burst signal between a wireless transmission station and a wireless reception station,
On the wireless transmission station side,
An information symbol conversion step for converting the input data bit sequence into an information symbol sequence;
A training signal generation step of generating a training signal sequence by differentially encoding a specified code sequence which is a pre-specified code sequence;
A multiplexing step of time-multiplexing the information symbol sequence and the training signal sequence to form a transmission burst signal;
A radio transmission step of converting the transmission burst signal into a radio signal;
Transmitting the wireless signal wirelessly in space, and
On the wireless receiving station side,
A receiving step of receiving a radio signal;
A radio reception step of converting the radio signal into a digital baseband signal;
A filtering step for band limiting the input signal;
A differential decoding step for performing differential decoding of the input digital signal for each sample at symbol intervals; and
A sliding cross-correlation operation step for performing a sliding cross-correlation operation for each of the designated code sequence and the sample with respect to the differentially decoded signal sequence;
A peak determination is performed on the cross-correlation value of each sample on which the sliding cross-correlation calculation has been performed, and a signal position corresponding to the cross-correlation value determined to be a peak is used as a head position of the transmission burst signal, and the transmission burst portion Frame synchronization step to extract,
A symbol synchronization step for extracting the symbol point from the input signal, using the signal position determined as the peak as a symbol point;
A frequency synchronization step of obtaining a phase component of the cross-correlation value determined to be the peak, regarding the phase component as a phase rotation per symbol by a carrier frequency offset, and performing the carrier frequency offset correction on the input signal;
A demodulation step of demodulating the offset-corrected signal to reproduce a data bit sequence.   The burst radio signal transmission method according to claim 21, wherein a PN code is used as the designated code sequence. On the wireless receiving station side,
The burst radio according to claim 21, wherein each step is performed on the received radio signal in the order of processing of the frame synchronization step, the frequency synchronization step, the filtering step, and the symbol synchronization step. Signal transmission method. On the wireless receiving station side,
N reception antennas are provided, and the reception step, the filtering step, and the differential decoding step are sequentially performed on the N reception signals received by the N reception antennas,
In the sliding cross-correlation calculation step, the differentially decoded N output signal sequences are regarded as one signal sequence, and a sliding cross-correlation calculation is performed with N of the designated code sequences for each sample;
In the frame synchronization step, peak determination is performed using the cross-correlation value of each sample on which the sliding cross-correlation calculation has been performed, and the signal position corresponding to the cross-correlation value determined to be the peak is used as the head position of the transmission burst signal. Extracting the transmission burst signal portion for N input signals,
In the symbol synchronization step, the signal position determined as the peak is set as a symbol point, and only the symbol point is extracted from N input signals,
In the frequency synchronization step, a phase component of the cross-correlation value determined to be the peak is obtained, the phase component is regarded as a phase rotation per symbol by a carrier frequency offset, and each of N input signals Perform carrier frequency offset correction,
The burst radio according to any one of claims 21 to 23, wherein in the demodulation step, the original data bit sequence is reproduced by performing diversity combining on the N received signals subjected to offset correction. Signal transmission method. On the wireless receiving station side,
N reception antennas, and for the N reception signals received by the N reception antennas, the reception step, the filtering step, the differential decoding step, the sliding correlation calculation step, The frame synchronization step and the symbol synchronization step are performed independently,
In the frequency synchronization step, all cross-correlation values determined as N peaks calculated in the frame synchronization step are added, a phase component of the added cross-correlation value is calculated, and the calculated phase component is used as a carrier frequency offset. And the carrier frequency offset correction is performed on N received signals.
In the demodulation step, the original data bit sequence is reproduced by performing diversity combining on the N received signals subjected to the filtering step, the frame synchronization step, the symbol synchronization step, and the frequency synchronization step. The burst radio signal transmission method according to any one of claims 21 to 23, wherein: On the wireless transmission station side,
A second training signal generating step of generating an alternating code that repeats +1 and −1 in two symbol periods as a second training signal sequence;
In the multiplexing step, the information symbol sequence, the training signal sequence, and the second training signal sequence are time-multiplexed to form a transmission burst signal,
On the wireless receiving station side,
The amplitude square value is subjected to discrete Fourier transform processing on the second training signal sequence portion of the transmission burst portion extracted in the frame synchronization step, and the phase component of the discrete Fourier transform value is defined as a symbol phase error. In view of this, the second symbol synchronization for performing further symbol phase error compensation on the symbol points extracted in the symbol synchronization step by adjusting the tap coefficient of the reception digital filter corresponding to the symbol phase error. The burst radio signal transmission method according to claim 21, further comprising a step.

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