JP4523294B2 - Communication device - Google Patents

Description

  The present invention relates to a communication apparatus constituting a digital wireless communication system or a digital broadcasting system, and in particular, an OFDM (Orthogonal Frequency Division Multiplexing) signal or a multicarrier CDMA (Code Division Multiplexing) signal used in the system. (Method: Code Division Multiple Access) The present invention relates to a communication apparatus that performs transmission path estimation based on a signal.

  Hereinafter, a conventional communication apparatus will be described. For example, in the OFDM modulation scheme, on the transmission side, an input signal is modulated by a scheme such as BPSK (Binary Phase Shift Keying), QPSK (Quadrature PSK), or QAM (Quadrature Amplitude Modulation) for each subcarrier, and thereby obtained. An OFDM modulated signal is generated by performing an inverse fast Fourier transform (IFFT) on the signal.

  On the other hand, the receiving side performs fast Fourier transform (FFT) on the received OFDM modulated signal, and demodulates the signal after the FFT processing.

  In the OFDM modulation method, a repetitive signal of a data portion called a guard interval (GI) is usually added and transmitted, thereby improving multipath tolerance. However, even when a delayed wave having a delay time exceeding the GI length is received or when the delay time of the delayed wave is within the GI, the FFT window timing is changed from the ideal timing due to the influence of noise and the delayed wave. If they are shifted, the preceding and succeeding OFDM symbols leak into the FFT window, causing intersymbol interference (ISI) and intercarrier interference (ICI). Deteriorates greatly. However, the ideal timing when the delay time of the delay wave is within the GI is the timing of the leading wave.

  For example, in a general “two-wave model of preceding wave and delayed wave”, when the FFT timing is shifted backward from the ideal symbol timing, the subsequent OFDM symbol component leaks into the preceding wave, and ISI and ICI occur. Specifically, the OFDM symbol (desired signal) to be demodulated and the subsequent OFDM symbol (interference symbol: ISI) leaking into the FFT window do not satisfy the number of samples of the FFT window width. Equivalent to multiplying a rectangular window by the number of samples above. Therefore, when such a signal is converted into a frequency axis signal by FFT processing, a function sin (Nx) / sin (x) (N: rectangular window width) determined by the rectangular window width is convolved with each symbol component. It will be.

  In this case, the OFDM symbol component to be demodulated in the FFT window is represented by the sum of the desired signal component and the ICI component leaking from other carriers at each carrier position. Similarly, the interference symbol component is represented by the sum of the ISI component and the ICI component leaking from other carriers at each carrier position.

The 24th Symposium on Information Theory and Its Applications (SITA2001) Kobe, Hyogo, Japan, Dec.4-7,2001 "OFDM demodulation method with variable effective symbol length using multistage inter-carrier interference canceller" IEICE technical report NS2002-75 RCS2002-103 (2002-07) "Adaptive equalization reception for multipath delay exceeding guard interval in scattered pilot OFDM signal"

  However, in the conventional communication apparatus, when the interference symbol leaks into the desired signal component, the function sin (Nx) / sin (x) determined by the rectangular window width is convoluted with the desired signal as described above. There are ICI from other carriers, ISI due to leaked symbols, and ICI due to the ISI component, and the SNR (Signal to Noise Ratio) is greatly reduced during transmission path estimation. Therefore, there has been a problem that the transmission path estimation accuracy is degraded and the reception characteristics are greatly degraded.

  The present invention has been made in view of the above, and an object of the present invention is to obtain a communication device that can realize highly accurate transmission path estimation even in a communication environment where intersymbol interference and intercarrier interference exist. To do.

  In order to solve the above-described problems and achieve the object, the communication apparatus according to the present invention employs a multi-carrier modulation / demodulation method and demodulates a reception signal in which a known signal is repeatedly inserted at a constant period. A frequency characteristic calculating means for calculating a frequency characteristic based on the known signal extracted from a received signal after Fourier transform, and generating a delay profile by performing an inverse Fourier transform on the frequency characteristic. A delay profile generating means for performing interference component removal for removing an interference signal component (including noise) below a predetermined threshold based on the delay profile, and a time axis based on the delay profile after removing the interference component By performing Fourier transform on the signal, the transmission path estimation value generator generates the frequency characteristics (transmission path estimation value) after removing the interference components. Characterized in that it comprises a means.

  According to the present invention, the frequency characteristic is calculated using a known signal (pilot signal) extracted from the frequency axis signal after the Fourier transform process, the frequency characteristic is converted from the frequency signal to the time axis signal, and then the Fourier transform is performed. By setting appropriate thresholds, intersymbol interference, intercarrier interference, noise, etc. that occur when the timing of the signal is shifted or when a delayed wave with a long delay time exceeding the guard interval length is received Then, the time axis signal is converted into the frequency axis signal again.

  According to the present invention, a frequency characteristic is calculated using a known pilot signal extracted from a frequency axis signal after FFT processing, the frequency characteristic is converted from a frequency signal to a time axis signal, and then the FFT timing is shifted. If a delayed wave with a long delay time exceeding the guard interval length is received, the intersymbol interference, intercarrier interference, and noise generated by setting an appropriate threshold value are removed, and the time axis is again displayed. The signal is converted into a frequency axis signal. Thereby, even in a communication environment where intersymbol interference and intercarrier interference exist, there is an effect that transmission path estimation with high accuracy can be realized.

  Embodiments of a communication apparatus according to the present invention will be described below in detail with reference to the drawings. Note that the present invention is not limited to the embodiments.

Embodiment 1 FIG.
FIG. 1 is a diagram showing a configuration of a first embodiment of a communication apparatus according to the present invention. This communication apparatus includes an FFT (Fast Fourier Transform) unit 1 that converts a received signal that is a time axis signal into a frequency axis signal, a symbol timing detection unit 2 that detects a symbol timing using the received signal, and the frequency axis signal. Is composed of a transmission path estimation section 3 that performs transmission path estimation using a signal and a demodulation section 4 that demodulates a frequency axis signal based on the transmission path estimation result.

  Here, the transmission path estimation process in the communication apparatus will be described. First, the received signal is input to the FFT timing unit 1 and simultaneously to the symbol timing detection unit 2. The symbol timing detection unit 2 establishes symbol synchronization using the received signal, and the FFT unit 1 extracts a preamble from the received signal based on the synchronization timing established by the symbol timing detection unit 2 and performs FFT processing. The received signal, which is a time axis signal, is converted into a frequency axis signal. Then, the transmission path estimation unit 3 performs transmission path estimation using the frequency axis signal, and the demodulation unit 4 demodulates the reception frequency axis signal based on the estimated value.

  FIG. 2 is a diagram illustrating the configuration of the transmission path estimation unit 3 according to the first embodiment. The pilot signal generation unit 5, the frequency characteristic calculation unit 6, the IFFT (Inverse FFT) unit 7, and the interference cancellation. A unit 8, a threshold setting unit 9, and an FFT unit 10 are included.

In the frequency characteristic calculation unit 6 that has received the frequency axis signal after the FFT processing, a known pilot signal (a signal in which a known symbol is repeatedly inserted at a constant period on the time axis) notified from the pilot signal generation unit 5 To obtain the frequency characteristics. When the frequency component of the received signal is Y (z) and the frequency component of the transmission signal (pilot signal) is X (z), the transfer function H (z) is obtained by the following (1).
H (z) = Y (z) / X (z) (1)

  The IFFT unit 7 converts the frequency characteristic from a frequency axis signal to a time axis signal to obtain a delay profile. In the obtained delay profile, the desired signal component appears as an impulse train, and interference components of ISI and ICI are scattered on the delay profile and exist in a state where the average signal level is low (Electronic Information and Communication Society Society Conference 2002, B-5-11 "Refer to threshold setting in OFDM symbol timing detection method").

  FIG. 3 is a diagram showing specific processing of the IFFT unit 7. Here, it is assumed that the desired signal component, the ISI component, and the ICI component exist at substantially the same level as the frequency axis signal. Under the condition that the pilot signal is scrambled, the interference signal component after the pilot demodulation is a random series, and the frequency characteristic calculation unit 6 can correctly obtain the frequency characteristic of only the desired signal component. Then, the IFFT unit 7 converts the received frequency characteristic from a frequency axis signal to a time axis signal by IFFT processing. As a result, the desired signal component appears in the impulse train, while the interference signal component is scattered on the delay profile, so that the average signal level is reduced. At this time, the level difference between the desired signal and the interference signal is determined by the FFT size.

  The interference removal unit 8 sets an appropriate threshold value sent from the threshold value setting unit 9, and forcibly sets the value of the sample point below the threshold value in the delay profile to zero, thereby reducing the interference. Remove signal components. Only the impulse train of the desired signal component is obtained. Note that noise (thermal noise) can be similarly removed by the interference removal unit 8 because signal components are scattered on the delay profile according to the Gaussian distribution. The threshold value set above is determined by the FFT size, propagation environment, noise level at the time of reception, etc., for example, may be calculated by a receiver, or stored in advance in a ROM table or the like. May be.

  Finally, the FFT unit 10 converts the time axis signal into the frequency axis signal again based on the delay profile after removing the interference signal component, and obtains a highly accurate transmission path estimation value.

  Thus, in the present embodiment, the frequency characteristic is calculated using a known pilot signal extracted from the frequency axis signal after the FFT processing, the frequency characteristic is converted from the frequency signal to the time axis signal, and then Inter-symbol interference, inter-carrier interference, and noise generated when the FFT timing is shifted or when a delayed wave with a long delay time exceeding the guard interval length is received can be eliminated by setting appropriate thresholds. Again, the time axis signal is converted to the frequency axis signal. Thereby, even in a communication environment where intersymbol interference and intercarrier interference exist, highly accurate transmission path estimation can be realized.

Embodiment 2. FIG.
FIG. 4 is a diagram showing a configuration of the transmission path estimation unit 3 according to the second embodiment, and includes a pilot extraction unit 11 in addition to the configuration of the first embodiment. In addition, about the structure similar to FIG. 1 and FIG. 2 of Embodiment 1 demonstrated previously, the same code | symbol is attached | subjected and the description is abbreviate | omitted. Here, only operations different from those of the first embodiment will be described.

  In this embodiment, for example, a transmission path estimation method using a scattered pilot (SP) of digital terrestrial broadcasting will be described. The scattered pilot is a signal in which a pilot signal having a known amplitude / phase characteristic is previously inserted in each subcarrier repeatedly at a constant period on the frequency axis or the time axis. FIG. 5 is a diagram illustrating an example of a scattered pilot.

  The pilot extraction unit 11 extracts a pilot signal arranged according to a predetermined rule from the frequency axis signal after FFT processing in the FFT unit 1. The pilot signal extracted by the pilot extraction unit 11 is thereafter processed in the same manner as in the first embodiment, and the transmission path estimation unit 3 finally outputs a transmission path estimation value.

  As described above, in the present embodiment, “a known pilot signal arranged according to a predetermined rule (repeatedly inserted at a constant cycle on the frequency axis or the time axis) extracted from the frequency axis signal after FFT processing. Frequency characteristic is calculated using the pilot signal), and the frequency characteristic is converted from a frequency signal to a time axis signal. After that, when the FFT timing is deviated or has a long delay time exceeding the guard interval length Intersymbol interference, intercarrier interference, and noise that occur when a delayed wave is received are removed by setting appropriate threshold values, and the time axis signal is converted to the frequency axis signal again. Thereby, even in a communication environment where intersymbol interference and intercarrier interference exist, highly accurate transmission path estimation can be realized.

  The processing of the present embodiment is not limited to the SP method for terrestrial digital broadcasting. For example, any transmission method in which pilot signals are repeatedly inserted at a constant period on the frequency axis or the time axis. Applicable.

Embodiment 3 FIG.
FIG. 6 is a diagram illustrating a configuration of the transmission path estimation unit 3 according to the third embodiment. In addition to the configuration according to the second embodiment, the frequency characteristic interpolation unit 12 is further included. In addition, about the structure similar to FIG. 1 and FIG. 4 demonstrated previously, the same code | symbol is attached | subjected and the description is abbreviate | omitted. Here, only operations different from those of the second embodiment will be described.

  In the present embodiment, for example, a pilot signal (for example, a scattered pilot signal) inserted repeatedly at a constant period on the time axis or the frequency axis as shown in the second embodiment described above is used. Assuming that the frequency characteristic is used, the frequency characteristic interpolating unit 12 further interpolates the frequency characteristic output from the frequency characteristic calculating unit 6 to obtain the frequency characteristic of the entire signal band.

  In the interpolation processing in the frequency characteristic interpolation unit 12, in addition to primary interpolation and secondary interpolation, interpolation by convolution of the ideal sinc function on the frequency axis is used.

  As described above, in the present embodiment, since the frequency characteristic obtained by the frequency characteristic calculation unit is subjected to the interpolation process to obtain the frequency characteristic of the entire signal band, compared with the first and second embodiments. Further, it is possible to realize transmission path estimation with higher accuracy.

Embodiment 4 FIG.
FIG. 7 is a diagram illustrating the configuration of the transmission path estimation unit 3 according to the fourth embodiment. In addition to the configuration of the third embodiment, the transmission path estimation unit 3 further includes an averaging unit 13. In addition, about the structure similar to FIG. 1 and FIG. 6 demonstrated previously, the same code | symbol is attached | subjected and the description is abbreviate | omitted. Further, in the present embodiment, for convenience of explanation, the averaging unit 13 is applied to the configuration of the third embodiment. However, the present invention is not limited to this, and the averaging unit 13 is the same as that of the first or second embodiment. The present invention can also be applied to the configuration (arranged after the frequency characteristic calculation unit 6). Here, only operations different from those of the third embodiment will be described.

  In the transmission path estimation method according to the present embodiment, the frequency characteristic of the frequency characteristic interpolation unit 12 output (or the frequency characteristic calculation unit 6 output) is averaged by the averaging unit 13 and then the time axis response delay profile is obtained. It was set as the structure which asks. As an averaging method, for example, a moving average using an FIR filter may be used, or averaging using an IIR filter may be used. Thereby, since the influence of noise can be reduced, more accurate transmission path estimation can be realized.

Embodiment 5 FIG.
FIG. 8 is a diagram illustrating a configuration of the transmission path estimation unit 3 according to the fifth embodiment, and includes a waveform shaping unit 14 instead of the IFFT unit 7 according to the fourth embodiment. In addition, about the structure similar to FIG. 1 and FIG. 7 which were demonstrated previously, the same code | symbol is attached | subjected and the description is abbreviate | omitted. In the present embodiment, the waveform shaping unit 14 is applied to the configuration of the fourth embodiment for convenience of explanation. However, the present invention is not limited to this, and the waveform shaping unit 14 is not limited to the first, second, or second embodiment. The present invention can also be applied to the third configuration. Here, only operations different from those of the fourth embodiment will be described.

  In the transmission path estimation method of the present embodiment, transmission path estimation is performed by removing waveform distortion that occurs when the FFT size and the number of carriers are different.

  Here, waveform distortion that occurs when the FFT size and the number of carriers are different will be described. FIG. 9 is a diagram illustrating an example when a two-wave model is assumed, FIG. 10 is a diagram illustrating frequency characteristics when the FFT size and the number of carriers are different, and FIG. 11 is a diagram illustrating frequency axis signals by IFFT processing. It is a figure which shows the time-axis response obtained when converting into a time-axis signal.

  In FIG. 10, since the number of carriers and the FFT size are different, this is equivalent to multiplication of a rectangular window on the frequency axis. In the time axis response of FIG. 11, a function sin (Nx) corresponding to the multiplied rectangular window. A waveform in which / sin (x) (N represents a rectangular window width) is convoluted is output. In this way, originally, only the impulse waveform exists, but the waveform of the function sin (Nx) / sin (x) is output to other sample points, resulting in waveform distortion. In the present embodiment, when the waveform distortion occurs, a good time axis response is obtained by performing waveform shaping.

  Next, processing of the waveform shaping unit 14 according to the present embodiment will be described. FIG. 12 is a diagram illustrating a configuration of the waveform shaping unit 14 according to the fifth embodiment. The waveform shaping unit 14 includes a frequency characteristic memory unit 31 that stores a frequency characteristic output from the averaging unit 13 and a frequency characteristic memory unit. IFFT unit 32 that converts the output signal 31 from a frequency axis signal to a time axis signal, a distortion removal unit 33 that shapes the time axis signal, a determination unit 34 that determines a signal after distortion removal, and after distortion removal Frequency characteristic update for calculating an updated frequency characteristic using the FFT unit 35 for converting the time signal from the time axis signal to the frequency axis signal, the output signal of the frequency characteristic memory unit 31 and the frequency axis signal output from the FFT unit 35 The unit 36 is configured.

  One output signal of the determination unit 34 is a switch a that selects a signal to be input to the IFFT unit 32 (selects an output signal of the frequency characteristic memory unit 31 or an output signal of the frequency characteristic update unit 36), and distortion. This is a signal (control signal shown) for controlling the switch b that selects the transmission destination of the output signal of the removal unit 33 (selects whether to output to the FFT unit 35 or output as a delay profile).

  First, the frequency characteristic obtained by the averaging unit 13 is stored in the frequency characteristic memory unit 31 and simultaneously sent to the IFFT unit 32. The IFFT unit 32 converts the received signal from the frequency axis signal to the time axis signal. Convert.

  Next, the distortion removing unit 33 shapes the waveform of the time axis signal output from the IFFT unit 32 according to a predetermined procedure.

  Here, the processing of the distortion removing unit 33 will be described in detail. FIG. 13 is a diagram illustrating a configuration of the distortion removing unit 33 according to the fifth embodiment. The distortion removing unit 33 includes a level calculating unit 41, a maximum value detecting unit 42, a threshold value defining unit 43, and a comparing unit. 44, a gate circuit 45, a waveform adding unit 46, and a memory unit 47. Here, the output signal from the IFFT unit 32 is m (t, i), where t represents time and i represents the i-th time waveform at time t.

  The m (t, i) is sent to the gate circuit 45 and simultaneously sent to the level calculation unit 41, and the level calculation unit 41 calculates the level of the time axis response in each sample. Next, the maximum value detection unit 42 detects the maximum value in the observation interval from the level of the time axis response, and the threshold value definition unit 43 uses the first threshold value for removing distortion from the maximum value. Th (1) is determined, and the first threshold th (1) is notified to the determination unit 34. The level of the time axis response may be an absolute value or a power value, and the threshold value defining unit 43 determines a threshold value according to each level.

  Next, the comparison unit 44 compares the level of the time axis response with the first threshold th (1), and the gate circuit 45 determines, for example, the sample based on the comparison result by the comparison unit 44. When the absolute value of the time response is larger than the first threshold th (1), the time axis signal (time axis response signal) is output as it is, and when it is small, it is forcibly set to 0. Output as.

Next, the waveform adding unit 46 adds the received time axis signal and the output signal of the memory unit 47 and outputs the result as a signal to the delay profile or the FFT unit 35. At the same time, this signal is stored in the memory unit 47. The output signal of the waveform adder 46 can be expressed as the following equation (2).
m (t, i) = m (t, i) + m (t, i−1) (2)
Note that when the IFFT unit 32 is output for the first time, the memory unit 47 is reset and the initial value is 0, so that the output signal of the gate circuit 45 is output as the addition result of the initial waveform addition unit 46. become.

  Next, in the determination unit 34 that has received the first threshold th (1) by the processing of the threshold value defining unit 43, the first threshold th (1) has been set in advance. It is determined whether it is larger or smaller than the threshold value TH. For example, when it is larger than the predetermined threshold value TH, the control signal is output so that the switch a is connected to the frequency characteristic updating unit 36 and the switch b is connected to the FFT unit 35. On the other hand, when it is smaller than the predetermined threshold TH, the control signal is connected so that the switch a is connected to the frequency characteristic memory unit 31 and the output signal of the distortion removing unit 33 is output as a delay profile. Is output. Further, the threshold value TH set above is determined by the FFT size, the number of carriers, the propagation environment, the noise level at the time of reception, etc., for example, may be calculated by the receiver, It may be described.

  Next, for example, when the output signal of the distortion removing unit 33 is input to the FFT unit 35 by the processing of the determination unit 34, the FFT unit 35 converts the received time axis signal into a frequency axis signal.

  Next, the frequency characteristic updating unit 36 uses the output signal of the frequency characteristic memory unit 31 and the frequency axis signal output from the FFT unit 35 to update the frequency characteristic in a predetermined procedure.

Here, the processing of the frequency characteristic updating unit 36 will be described in detail. FIG. 14 is a diagram illustrating a configuration of the frequency characteristic update unit 36 according to the fifth embodiment, and the frequency characteristic update unit 36 includes a frequency characteristic difference calculation unit 51. The frequency characteristic difference calculation unit 51 obtains a signal by taking a difference in the signal band between the frequency characteristic (frequency axis signal) received from the FFT unit 35 and the frequency characteristic (frequency axis signal) received from the frequency characteristic memory unit 31. The frequency characteristic in the band is updated, and the update result is sent to the IFFT unit 32 again. The processing of the frequency characteristic update unit 36 can be expressed by the following equation (3).
H ′ (z) = H (z) −H ′ (z) (3)
Note that H ′ (z) on the right side of the above expression represents the frequency characteristic after distortion is removed, H (z) represents the frequency characteristic input to the frequency characteristic memory unit 31, and H ′ ( z) represents the updated frequency characteristics.

  Thereafter, the determination unit 34 determines that the i-th threshold th (i) (i is the number of repetitions) output from the distortion removal unit 33 is smaller than the predetermined threshold value TH. The above process is repeated until the output signal is output as a delay profile. Then, when the output signal of the distortion removing unit 33 is output as a delay profile, the value of the memory unit 47 in the distortion removing unit 33 is reset. Thereafter, when a new frequency characteristic is output from the averaging unit 13, the waveform shaping operation by the waveform shaping unit 14 is performed again.

  Next, the operation of the waveform shaping unit 14 will be described in detail with reference to the drawings. Here again, the two-wave model shown in FIG. 9 is assumed. Further, the frequency characteristic input to the frequency characteristic memory unit 31 is expressed as shown in FIG. 10, and further, the time response converted from the frequency axis signal to the time axis signal by the IFFT unit 32 and input to the distortion removing unit 33. Is expressed as shown in FIG.

  In this time axis response observation range, the level calculating unit 41 obtains the amplitude value at each sample point, the maximum value detecting unit 42 detects the maximum value, and the threshold value defining unit 43 is the optimum first threshold value. Th (1) is determined. Then, the comparison unit 44 and the gate circuit 45 forcibly set the time response equal to or less than the first threshold th (1) to zero. FIG. 15 is a diagram illustrating processing results of the comparison unit 44 and the gate circuit 45. Here, only the first wave is output.

  The first threshold th (1) has a known FFT size and the number of carriers, and the shape of the function sin (Nx) / sin (x) to be convolved is also known. Set. In the above function, for example, the maximum side lobe is about 13 dB smaller than the main lobe. Therefore, the first threshold th (1) takes into account the error due to noise or the like, and takes a maximum value and a level 13 dB down from that value. It is desirable to set between.

  When the determination unit 34 determines that the first threshold th (1) is greater than TH, the output signal after waveform shaping is stored in the memory unit 47 and at the same time the FFT unit 35 The time axis signal is converted into the frequency axis signal, and the result is output to the frequency characteristic update unit 36. FIG. 16 is a diagram illustrating a processing result of the FFT unit 35. Here, by removing the influence of the convolution of the function sin (Nx) / sin (x), the frequency characteristic of only one wave is obtained, and the frequency characteristic outside the signal band is also extrapolated.

  The frequency characteristic updating unit 36 that has received the frequency axis signal shown in FIG. 16 takes the difference in the signal band between the frequency axis signal and the output signal of the frequency characteristic memory unit 31 and updates the frequency characteristic in the signal band. Thereby, for example, the frequency characteristic of the second wave deleted by the first waveform shaping is obtained. FIG. 17 is a diagram illustrating a processing result of the frequency characteristic update unit 36.

  The IFFT unit 32 that has received the frequency characteristics shown in FIG. 17 converts the received signal from the frequency axis signal to the time axis signal again and inputs it to the distortion removing unit 33. The time response at this time appears only in the state in which the function sin (Nx) / sin (x) is convoluted only for the signal component that was discarded in the previous waveform shaping. FIG. 18 is a diagram illustrating a processing result of the IFFT unit 32.

  The distortion removing unit 33 that has received the time response shown in FIG. 18 performs the same process as described above again to determine the second threshold th (2), and the second threshold th (2) or less. The time response is forcibly set to zero. FIG. 19 is a diagram illustrating a processing result of the gate circuit 45.

  Further, the waveform adder 46 that receives the time axis signal shown in FIG. 19 adds the time axis signal to the signal stored in the memory unit 47. FIG. 20 is a diagram illustrating a processing result of the waveform adding unit 46. With this process, ideal waveforms of the first wave and the second wave are obtained. Thereafter, the waveform shaping unit 14 repeats the process shown in the above drawing until the i-th threshold value th (i) is smaller than a preset threshold value TH.

  Thus, in the present embodiment, time response waveform shaping (first time) is performed using the first threshold value determined to be an optimal value, and the frequency response of the time response after the waveform shaping is obtained. Furthermore, the frequency characteristic is updated using the difference between the frequency characteristic and the original frequency characteristic. Then, the time response waveform shaping (second time) is performed again using the newly determined second threshold value, and the time response is combined with the time response shaped in the first processing. Thereafter, the waveform shaping (i-th time) and the synthesis process are repeated until the i-th threshold value becomes smaller than a predetermined threshold value set in advance. Thereby, even when the FFT size and the number of carriers are different, high-accuracy transmission path estimation can be realized.

  Note that the processing of the present embodiment is applicable to all transmission schemes in which transmission path estimation is performed by FFT processing or IFFT processing, and the FFT size and the number of carriers are different. It is not limited by a transmission method such as single carrier transmission.

Embodiment 6 FIG.
FIG. 21 is a diagram illustrating the configuration of the transmission path estimation unit 3 according to the sixth embodiment, in which the averaging unit 13 subsequent to the frequency characteristic interpolation unit 12 illustrated in the fifth embodiment is deleted, and the subsequent stage of the interference cancellation unit 8 is deleted. The averaging unit 15 is inserted in the configuration. In addition, about the structure similar to FIG. 1 and FIG. 8 demonstrated previously, the same code | symbol is attached | subjected and the description is abbreviate | omitted. Further, in the present embodiment, for convenience of explanation, the averaging unit 15 is applied to the configuration of the fifth embodiment. However, the present invention is not limited to this, and the averaging unit 15 is the same as that of the first to fourth embodiments. It can also be applied to configurations. Here, only operations different from those of the fifth embodiment will be described.

  In the transmission path estimation method of the present embodiment, the time response output from the interference removal unit 8 is averaged by the averaging unit 15 and the transmission path estimation value is obtained from the averaged time response. As an averaging method, for example, a moving average using an FIR filter or an averaging using an IIR filter may be used as in the averaging unit 13. Thereby, the influence of noise can be further reduced, and transmission path estimation can be performed with higher accuracy.

Embodiment 7 FIG.
FIG. 22 is a diagram showing the configuration of the communication apparatus according to the seventh embodiment of the present invention. This communication device includes a canceller unit 21 in addition to the configuration of the communication device shown in the first embodiment, and can be applied to the configurations of the second to sixth embodiments, for example. In addition, about the structure similar to Embodiment 1-6 demonstrated previously, the same code | symbol is attached | subjected and the description is abbreviate | omitted. Here, only operations different from the first to sixth embodiments will be described.

  In the present embodiment, the received signal on the time axis is converted into a frequency axis signal by FFT processing (processing of the FFT unit 1), and the canceller unit 21 transmits the frequency axis signal and the transmission output from the transmission path estimation unit 3. The ISI and ICI are removed using the path estimation value, and the result is output to the demodulator 4.

  Here, the processing of the canceller unit 21, which is a feature of the present embodiment, will be described. FIG. 23 is a diagram illustrating a configuration of the canceller unit 21 according to the seventh embodiment. The canceller unit 21 includes a remodulation unit 61, a subtraction unit 62, a frequency signal interpolation unit 63, and a subtraction unit 64. The

  The frequency axis signal output from the FFT unit 1 is input to the subtraction units 62 and 64 and simultaneously to the transmission path estimation unit 3, and the transmission path estimation unit 3 uses the frequency axis signal according to the second embodiment. The transmission path estimation value is obtained by the processes of ˜6.

  The remodulation unit 61 that has received the transmission channel estimation value output from the transmission channel estimation unit 3 remodulates (re-scrambles) the transmission channel estimation value, and regenerates the desired signal component (re-synchronization) at the pilot carrier position. Modulation signal).

  The subtractor 62 subtracts the remodulated signal output from the remodulator 61 from the frequency axis signal output from the FFT unit 1. This difference becomes an interference component at the pilot carrier position. FIG. 24 is a diagram illustrating an interference signal component at the pilot carrier position. In FIG. 24, for convenience of explanation, a function sin (Nx) / sin (x) is convolved with one carrier and ICI is generated by ISI, and pilots are generated at a rate of one for every two carriers in the frequency axis direction. A system in which a signal is inserted is assumed. In this case, only the interference signal component from which the desired signal component is removed appears at the pilot carrier position.

  The frequency signal interpolation unit 63 that has received the interference signal component at the pilot carrier position obtains the interference signal component at the data carrier position other than the pilot carrier position by interpolation processing. By this interpolation processing, ISI and ICI components of the entire signal band are obtained. In addition to the primary interpolation and the secondary interpolation, for example, interpolation by convolution on the frequency axis of the ideal sinc function may be used as the interference signal component interpolation processing.

  The subtracting unit 64 that has received the signal after interpolation processing (post-interpolation signal) subtracts the interpolated signal from the frequency axis signal output from the FFT unit 1, and outputs the desired signal component after removing the interference signal component. The demodulator 4 performs a predetermined demodulation process using the desired signal component.

  Thus, in the present embodiment, the remodulation signal of the transmission path estimation value output from transmission path estimation section 3 is subtracted from the frequency axis signal output from FFT section 1 to generate an interference signal component at the pilot carrier position. . Thereafter, the interference signal component at the data carrier position is obtained by performing interpolation processing on the interference signal component at the pilot carrier position. Then, after removing the interference signal component from the frequency axis signal output from the FFT unit 1, demodulation processing is performed using only the desired signal component. Thereby, since ISI and ICI can be removed with high accuracy, a good demodulation result can be obtained.

Embodiment 8 FIG.
FIG. 25 is a diagram showing the configuration of the eighth embodiment of the communication apparatus according to the present invention. In this communication apparatus, the transmission path estimation unit 3 generates a signal for controlling the FFT window timing of the FFT unit 1 based on the obtained delay profile in addition to the processing of the second to sixth embodiments described above. The signal is output to the FFT unit 1. In addition, about the structure similar to Embodiment 1-7 demonstrated previously, the same code | symbol is attached | subjected and the description is abbreviate | omitted. Here, only operations different from those of the seventh embodiment will be described.

  For the interpolation processing performed in the present embodiment, the interpolation possible range is determined by the pilot signal insertion interval. For example, when all the signal bands are pilot signals, the interpolation possible range is a range of a total FFT size of −FFT size / 2 to + FFT size / 2. When every other pilot signal is inserted in the entire signal band, the interpolation possible range is a range of the total FFT size / 2 of −FFT size / 4 to + FFT size / 4, and the pilot insertion interval is Np. Then, the interpolation possible range is a range of a total FFT size / Np of −FFT size / 2Np to + FFT size / 2Np.

  For example, when the number of leaking samples exceeding the interpolation range occurs, ISI and ICI interpolation accuracy deteriorates and characteristics deteriorate. This will be described with reference to FIG. FIG. 26 is a diagram illustrating an example of the timing of the preceding wave and the delayed wave. Here, the FFT window is set in accordance with the timing of the delayed wave (1). In this case, as shown in the figure, subsequent OFDM symbols leak into the FFT window, so that ISI and ICI occur. Since the number of leaked samples (rectangular window width) exceeds the interpolable range determined by the pilot signal insertion interval, the interpolation accuracy of the ISI and ICI components deteriorates, and the interference signal remains. As a result, the demodulation characteristics deteriorate.

  Therefore, in the present embodiment, the FFT window timing in the FFT unit 1 is shifted forward by the control by the transmission path estimation unit 3 so that the number of leaked samples of the subsequent OFDM symbol with respect to the preceding wave falls within the above interpolation range. FIG. 27 is a diagram illustrating control of the transmission path estimation unit 3 according to the present embodiment. Thereby, the interpolation accuracy is improved as compared with the case of FIG. 26, and ISI and ICI due to leakage of the subsequent OFDM symbol with respect to the preceding wave can be canceled with high accuracy.

  In the case of FIG. 27, since the FFT window timing is shifted forward, the OFDM symbol component of the delayed wave (2) leaks into the FFT window. The number of leaked samples of this OFDM symbol is also equal to the pilot signal. If it is within the interpolation range, ISI and ICI can be canceled with high accuracy.

  In the present embodiment, when a subsequent OFDM symbol leaks, the FFT window timing is shifted forward so that the number of leaked samples falls within the interpolable range of the pilot signal. The present invention is not limited to this, and when leakage of the previous OFDM symbol occurs, it is possible to shift the FFT window timing backward to perform processing for keeping the number of leaked samples within the interpolation range of the pilot signal.

  In addition, if the number of leaked samples exceeding the interpolation range occurs regardless of how it is shifted, for example, it is considered that the interference signal component of the path with large power is large and the interference component of the small path is small. Therefore, the FFT window timing is controlled so that the number of leaked samples in a path with a large power is preferentially reduced based on the power of each path.

  Then, after a new FFT window timing is determined by the processing of the transmission path estimation unit 3, the FFT unit 1 performs an FFT process on the received signal again at that timing. At this time, the FFT unit 1 holds the received signal until the FFT window timing is notified from the transmission path estimation unit 3.

  As described above, in the present embodiment, the transmission path estimation unit 3 uses the FFT window timing so that the number of leaked samples is within the interpolation range determined by the pilot signal insertion interval based on the delay profile. To control. Thereby, the interpolation accuracy is improved and ISI and ICI can be canceled with high accuracy.

Embodiment 9 FIG.
FIG. 29 is a diagram showing the configuration of the ninth embodiment of the communication apparatus according to the present invention. In this communication apparatus, the transmission path estimation result obtained by the transmission path estimation unit 3 is output to the interference amount calculation unit 23, and the interference amount in the FFT window is calculated based on the delay amount and transmission path fluctuation value of each path obtained from the delay profile. Is calculated, a signal for controlling the FFT window timing of the FFT unit 1 is generated based on the information, and the signal is output to the FFT unit 1. In addition, about the structure similar to Embodiment 1-8 demonstrated previously, the same code | symbol is attached | subjected and the description is abbreviate | omitted. Here, only operations different from those of the eighth embodiment will be described.

  In the present embodiment, the amount of interference in the FFT window when the FFT window timing is set over a preset time range is calculated, and the FFT window timing at which the amount of interference is minimized is detected.

  Specifically, the interference amount calculation unit 23 uses the delay time of each path obtained by the transmission path estimation unit 3 and the transmission path fluctuation value, and the FFT window at each FFT window timing within a preset time range. The total amount of interference based on the ISI and ICI generated within is calculated, and the FFT window timing at which the amount of interference is minimized is detected. Then, the information is output to the FFT unit 1. The FFT unit 1 performs FFT processing on the received signal anew at that timing. At this time, the FFT unit 1 holds the received signal until the FFT window timing is notified from the interference amount calculation unit 23.

  As described above, in the present embodiment, the interference amount calculation unit 23 minimizes the interference amount in the FFT window based on the delay time and the transmission path fluctuation value of each path obtained by the transmission path estimation unit 3. Thus, the FFT window timing is controlled (first method using the configuration of FIG. 29). Thereby, ISI and ICI can be canceled with high accuracy.

  In the present embodiment, not limited to the above, for example, the residual interference amount in the FFT window in the signal after passing through the canceller unit 21 when the FFT window timing is set within a preset time range is calculated. Then, the FFT window timing that minimizes the residual interference amount may be detected, and the FFT window timing in the FFT unit 1 may be controlled (second method using the configuration of FIG. 29).

  Here, the second method will be specifically described. First, the process in the canceller unit 21 is a technique of subtracting a replica generated by the interference signal interpolation process from the received signal, and the process is constituted by a linear process. In addition, since ISI generated in the FFT window is caused by leakage of preceding and subsequent symbols into the FFT window, and ICI is generated due to mismatch between the FFT window width and the OFDM symbol length included therein, The amount of interference can be obtained from the path delay time and the path transmission line fluctuation value.

  Therefore, the residual interference amount after passing through the canceller unit 21 when the FFT window timing is set within a preset time range is calculated from the delay amount of each path obtained by the transmission line estimation unit 3 and the transmission line fluctuation value. Then, the FFT window timing that minimizes the amount of interference is detected. The subsequent processing is the same as in the first method. Thereby, the effect similar to the above can be acquired. In the present embodiment, the interference amount in the FFT window before and after passing through the canceller unit 21 has been described. However, instead of the interference amount, processing is performed using SIR in the FFT window. You can also. For example, a more optimal FFT window timing can be detected when there is a temporal variation in the transmission path within a preset time range in which the FFT window timing at which the amount of interference is minimized is detected. On the other hand, when the temporal variation of the transmission line can be ignored within the set time range, the amount of calculation can be reduced by using the interference amount.

Embodiment 10 FIG.
FIG. 28 is a diagram showing the configuration of the communication apparatus according to the tenth embodiment of the present invention. In this communication apparatus, the function of the interference removal unit 8 of the transmission path estimation unit 3, the function of the canceller unit 21, and the control signal generation function of the FFT window timing in the transmission path estimation unit 3 can be stopped. Note that the stop function of the present embodiment is also applicable to the configuration of the ninth embodiment described above. In this case, the function of the interference amount calculation unit 23 is stopped instead of the control signal generation function of the FFT window timing in the transmission path estimation unit 3. Moreover, about the structure similar to Embodiment 1-9 demonstrated previously, the same code | symbol is attached | subjected and the description is abbreviate | omitted.

  For example, the reception determination unit 22 determines the current reception status based on the received signal and information generated by the transmission path estimation unit 3. For example, when the interference level and the noise level are low, the reception determination unit 22 Control to stop the function of the interference removal unit 8, the function of the canceller unit 21, and the FFT window timing control signal generation function in the transmission path estimation unit 3 is performed (switch control is described as an example in FIG. 28). . As a result, it is possible to realize high-speed demodulation processing according to the reception status. In the present embodiment, for convenience of explanation, the control of the reception determination unit 22 is applied to the communication apparatus of FIG. 25 described above. However, the present embodiment is not limited to this, and FIGS. The present invention can also be applied to the above configuration.

  In the first to tenth embodiments, the case where the OFDM scheme is used as the communication scheme has been described. However, the present invention is not limited to this, and can be applied to, for example, a communication apparatus employing a multicarrier CDMA scheme. .

Embodiment 11 FIG.
FIG. 30 is a diagram showing the configuration of the eleventh embodiment of the communication apparatus according to the present invention. This communication apparatus includes receiving antennas 101 and 111, FFT units 102 and 112, multiplication units 103 and 113, a symbol timing detection unit 121, a transmission path estimation / weight coefficient calculation unit 122, a synthesis unit 123, and a demodulation unit. It consists of part 4. Here, the weighting coefficient calculation method regarding the diversity combining reception method will be described.

  First, in the branch (1), the FFT unit 102 performs FFT processing on the signal received by the receiving antenna 101 based on the FFT window timing information transmitted from the symbol timing detection unit 121, and generates a time axis signal. The received signal is converted into a frequency axis signal. Then, the multiplication unit 103 multiplies the frequency axis signal by the weighting factor obtained by the transmission path estimation / weighting factor calculation unit 122. On the other hand, in the branch (2), the FFT unit 112 performs FFT processing on the signal received by the receiving antenna 111 based on the FFT window timing information transmitted from the symbol timing detection unit 121, and generates a time axis signal. The received signal is converted into a frequency axis signal. Then, the multiplication unit 113 multiplies the frequency axis signal by the weighting factor obtained by the transmission path estimation / weighting factor calculation unit 122. Next, the synthesizer 123 synthesizes the outputs of the multipliers of the branch (1) and the branch (2), and then the demodulator 4 performs a predetermined demodulation process.

  The symbol timing detection unit 121 is provided with the functions of the symbol timing detection unit 2 described above for the number of branches. The symbol timing detection unit 121 is provided for the FFT unit of each branch so as to achieve symbol synchronization between the branches. FFT window timing information is output.

  FIG. 31 is a diagram showing a detailed configuration of transmission path estimation / weighting coefficient calculation section 122 in the eleventh embodiment. Transmission path estimation sections 131 and 141, remodulation sections 132 and 142, subtraction sections 133, and 143, SIR calculation units 134 and 144, and a weight coefficient calculation unit 151. The operation of the transmission path estimation / weighting coefficient calculation unit 122 will be described below.

  First, the output signal of FFT section 102 is sent to multiplication section 103 and also to transmission path estimation section 131. Transmission path estimation section 131 uses transmission path estimation section 3 described above (Embodiment 1). (Corresponding to .about.6), and the output is notified to the weighting factor calculation unit 151 and the remodulation unit 132. Note that the output of the transmission path estimation unit 131 is also notified to the demodulation unit 4 and the symbol timing detection unit 121.

  In addition, subtraction unit 133 obtains an interference signal component at the pilot carrier position by the same processing as that of the seventh embodiment described above (see FIG. 23). Then, SIR calculation section 134 calculates an SIR (signal power to interference signal power ratio) at each pilot carrier position using the interference signal component at this pilot carrier position (e.g., equivalent to a scattered pilot signal position). . Furthermore, the SIR calculation unit 134 obtains the SIR value over the entire signal band (all subcarriers) by obtaining the SIR value at the data carrier position by interpolation processing using the SIR value at the pilot carrier position. As the interpolation processing used here, there are various methods such as step interpolation, primary interpolation, and secondary interpolation that directly give the SIR value of the pilot carrier position with a small carrier number (low frequency).

  On the other hand, the same processing is performed for the branch (2), and the SIR calculation unit 144 obtains the SIR value over the entire signal band.

  Next, the weighting factor calculation unit 151 determines weighting factors in all signal bands based on the SIR values over the entire signal bands output from the SIR calculation units 134 and 144 and the outputs of the transmission path estimation units 131 and 141. .

Specifically, the output of the transmission path estimation unit 131 is H1 (f), the output of the transmission path estimation unit 141 is H2 (f), the output of the SIR calculation unit 134 is SIR1 (f), and the output of the SIR calculation unit 144 is SIR2. Assuming that (f), the weight coefficient W1 (f) of the branch (1) and the weight coefficient W2 (f) of the branch (2) are obtained as follows.
W1 (f) = SIR1 (f) / {H1 (f) × (SIR1 (f) + SIR2 (f))}
... (4)
W2 (f) = SIR2 (f) / {H2 (f) × (SIR1 (f) + SIR2 (f))}
... (5)

  Then, the weighting factors W1 (f) and W2 (f) obtained as described above are sent to the multipliers 103 and 113 and to the demodulator 4 respectively.

  Next, the multipliers 103 and 113 multiply the frequency coefficient signal of each branch by the weight coefficient, the synthesizer 123 synthesizes the multiplication results, and finally the demodulator 4 determines the transmission path estimation value. Then, demodulation processing is performed using the weighting coefficient.

  Thus, in the present embodiment, the diversity combining is performed using the SIR value in each branch as the weighting factor. As a result, it is possible to greatly improve reception characteristics in a reception environment in which interference components and noise components are dominant. In the present embodiment, the case where the number of branches is 2 has been described as an example. However, the present invention is not limited to this, and the present invention can be similarly applied even when the number of branches is larger.

Embodiment 12 FIG.
FIG. 32 is a diagram illustrating a detailed configuration of the transmission path estimation / weighting coefficient calculation unit 122 according to the twelfth embodiment. The transmission path estimation units 131 and 141, the SIR calculation units 135 and 145, and the weighting coefficient calculation unit 151 are illustrated in FIG. I have. Here, a weighting coefficient calculation method different from that in Embodiment 11 will be described. In addition, about the structure similar to Embodiment 11 demonstrated previously, the same code | symbol is attached | subjected and the description is abbreviate | omitted. Here, only operations different from those of the eleventh embodiment will be described.

  First, the output signal of FFT section 102 is sent to multiplication section 103 and also to transmission path estimation section 131. Transmission path estimation section 131 uses transmission path estimation section 3 described above (Embodiment 1). (Corresponding to .about.6) and the output is notified to the weighting factor calculation unit 151 and the SIR calculation unit 135.

  Next, the SIR calculation unit 135 calculates the SIR in the currently set FFT window based on the delay time and the transmission channel fluctuation value obtained by the transmission channel estimation unit 131.

  On the other hand, the same processing is performed for the branch (2), and the SIR calculation unit 145 obtains the SIR value in the FFT window of the branch (2). The SIR values of branch (1) and branch (2) are the same in all signal bands (all subcarriers).

  Next, the weighting factor calculation unit 151 determines weighting factors in the entire signal band based on the SIR values over the entire signal band output from the SIR calculation units 135 and 145 and the outputs of the transmission path estimation units 131 and 141. .

Specifically, when the output of the transmission path estimation unit 131 is H1 (f), the output of the transmission path estimation unit 141 is H2 (f), the output of the SIR calculation unit 135 is SIR1, and the output of the SIR calculation unit 145 is SIR2. The weighting factor W1 (f) of the branch (1) and the weighting factor W2 (f) of the branch (2) are obtained as follows.
W1 (f) = SIR1 / {H1 (f) × (SIR1 + SIR2)}
... (6)
W2 (f) = SIR2 / {H2 (f) × (SIR1 + SIR2)}
... (7)

  Thus, in the present embodiment, the diversity combining is performed using the SIR value in the FFT window of each branch as the weighting factor. As a result, it is possible to greatly improve reception characteristics in a reception environment in which interference components and noise components are dominant. In the present embodiment, the case where the number of branches is 2 has been described as an example. However, the present invention is not limited to this, and the present invention can be similarly applied even when the number of branches is larger.

Embodiment 13 FIG.
FIG. 33 is a diagram showing the configuration of the communication device according to the thirteenth embodiment of the present invention. This communication apparatus includes receiving antennas 101 and 111, canceller function units 104 and 114, multiplication units 103 and 113, a weight coefficient calculation unit 124, a synthesis unit 123, and a demodulation unit 4. In the present embodiment, the weighting coefficient of each branch is calculated after the interference cancellation function is operated in each branch, and then diversity combining is performed on the signal of each branch after being multiplied by the weighting coefficient. In addition, about the structure similar to Embodiment 11 demonstrated previously, the same code | symbol is attached | subjected and the description is abbreviate | omitted. Here, only operations different from those of the eleventh embodiment will be described.

  In the thirteenth embodiment, in each branch, the interference cancellation function described in the first to tenth embodiments (corresponding to the processing of the transmission path estimation unit 3, the canceller unit 21, and the interference amount calculation unit 23) is used to cause interference from the received signal. Remove ingredients. FIG. 34 is a diagram illustrating a configuration for the canceller function units 104 and 114 to realize the canceller function illustrated in FIG. 25 as an example.

  Here, the signal after interference cancellation is sent to multiplication sections 103 and 113 and also sent to weighting coefficient calculation section 124 (see FIG. 33). FIG. 35 is a diagram showing the configuration of the weighting factor calculation unit 124. The weighting factor calculation unit 124 is the transmission channel estimation / weighting factor calculation unit of FIG. 31 except that the transmission channel estimation value is given from the outside. The same processing as 122 is performed.

  Thus, in this embodiment, after the interference cancellation function according to Embodiments 1 to 10 is operated for each branch, the weighting factor of each branch is based on the SIR value obtained by the method of Embodiment 11. It was set as the structure which determines. Accordingly, the interference cancellation function and the diversity function operate effectively, and the reception characteristics can be further improved. In the present embodiment, the case where the number of branches is 2 has been described as an example. However, the present invention is not limited to this, and the present invention can be similarly applied even when the number of branches is larger.

Embodiment 14 FIG.
FIG. 36 is a diagram showing a detailed configuration of the weight coefficient calculation unit 124 according to the fourteenth embodiment, and includes SIR calculation units 135 and 145. In addition, about the structure similar to Embodiment 13 demonstrated previously, the same code | symbol is attached | subjected and the description is abbreviate | omitted. Here, only operations different from those of the thirteenth embodiment will be described.

  The weighting factor calculation unit 124 of the present embodiment performs the same processing as the transmission channel estimation / weighting factor calculation unit 122 in FIG. 32 except that the transmission channel estimation value is given from the outside.

  As described above, in this embodiment, after the interference cancellation function according to Embodiments 1 to 10 is operated for each branch, the weight coefficient of each branch is based on the SIR value obtained by the method of Embodiment 12. It was set as the structure which determines. Accordingly, the interference cancellation function and the diversity function operate effectively, and the reception characteristics can be further improved. In the present embodiment, the case where the number of branches is 2 has been described as an example. However, the present invention is not limited to this, and the present invention can be similarly applied even when the number of branches is larger.

Embodiment 15 FIG.
FIG. 37 is a diagram showing the configuration of the fifteenth embodiment of the communication apparatus according to the present invention. This communication apparatus includes receiving antennas 101 and 111, FFT units 102 and 112, multiplication units 103 and 113, a symbol timing detection unit 121, a transmission path estimation / weight coefficient calculation unit 122, a synthesis unit 123, and a canceller. The function unit 126 and the demodulation unit 4 are included. In this embodiment, the diversity reception system shown in the above-described eleventh and twelfth embodiments is provided with a canceller function unit. In addition, about the structure similar to Embodiment 11 and 12 demonstrated previously, the same code | symbol is attached | subjected and the description is abbreviate | omitted. Here, a canceller function unit 126 different from the eleventh or twelfth embodiment will be described.

  FIG. 38 is a diagram illustrating a configuration of the canceller function unit 126. Here, using the transmission path estimation values and weighting coefficients of branch (1) and branch (2) obtained by transmission path estimation / weighting coefficient calculation section 122, transmission path estimation value correction section 127 The channel estimation value is calculated, and the canceller unit 21 removes interference components from the combined signal. Specifically, the corrected transmission path estimation value is obtained by multiplying the transmission path estimation value of each branch by the weighting coefficient and combining the same as the processing performed by the multiplication sections 103 and 113 and the combining section 123. . Note that the transmission path estimation / weighting factor calculation unit 122 may have the function of the interference amount calculation unit 23, thereby further obtaining the effects shown in the ninth embodiment.

  As described above, in this embodiment, the interference cancellation function is operated after weighted synthesis of the received signals of the respective branches. Accordingly, the interference cancellation function and the diversity function operate effectively, and the reception characteristics can be further improved. In the present embodiment, the case where the number of branches is 2 has been described as an example. However, the present invention is not limited to this, and the present invention can be similarly applied even when the number of branches is larger.

Embodiment 16 FIG.
FIG. 39 is a diagram showing the configuration of the communication device according to the sixteenth embodiment of the present invention. This communication apparatus includes receiving antennas 101 and 111, FFT units 102 and 112, symbol timing detection unit 2, transmission path estimation unit 3, interference amount calculation unit 31, combining unit 123, canceller unit 21, It comprises a demodulator 4. In the present embodiment, the circuit is simplified and the operation speed is increased by combining without performing weighted combining. In addition, about the structure similar to embodiment demonstrated previously, the same code | symbol is attached | subjected and the description is abbreviate | omitted.

  According to the present embodiment, for example, when the reception propagation environment of each branch is independent, the reception signals of each branch are combined without weighting. As a result, simplification of the circuit and high-speed operation can be realized, and furthermore, the influence of the path generated by the ISI or ICI in the received signal can be suppressed relatively small, thereby improving the characteristics. The configuration in FIG. 39 is equivalent to the configuration in FIG. 40 or FIG.

  As described above, the communication apparatus according to the present invention is useful for a digital wireless communication system and a digital broadcasting system, and in particular, a communication apparatus that performs transmission path estimation based on an OFDM signal or a multicarrier CDMA signal used in the system. Suitable as

It is a figure which shows the structure of Embodiment 1 of the communication apparatus concerning this invention. It is a figure which shows the structure of Embodiment 1 of a transmission-line estimation part. It is a figure which shows the specific process of an IFFT part. It is a figure which shows the structure of Embodiment 2 of a transmission-line estimation part. It is a figure which shows an example of a scattered pilot. It is a figure which shows the structure of Embodiment 3 of a transmission-line estimation part. It is a figure which shows the structure of Embodiment 4 of a transmission-line estimation part. It is a figure which shows the structure of Embodiment 5 of a transmission-line estimation part. It is a figure which shows an example at the time of assuming a 2 wave model. It is a figure which shows the frequency characteristic when FFT size and the number of carriers differ. It is a figure which shows the time-axis response obtained when converting from a frequency-axis signal to a time-axis signal by IFFT processing. FIG. 10 is a diagram illustrating a configuration of a waveform shaping unit according to a fifth embodiment. FIG. 10 is a diagram illustrating a configuration of a distortion removing unit according to a fifth embodiment. FIG. 10 is a diagram illustrating a configuration of a frequency characteristic update unit according to a fifth embodiment. It is a figure which shows the processing result of a comparison part and a gate circuit. It is a figure which shows the processing result of an FFT part. It is a figure which shows the processing result of a frequency characteristic update part. It is a figure which shows the processing result of an IFFT part. It is a figure which shows the processing result of a gate circuit. It is a figure which shows the processing result of a waveform addition part. It is a figure which shows the structure of Embodiment 6 of a transmission-line estimation part. It is a figure which shows the structure of Embodiment 7 of the communication apparatus concerning this invention. FIG. 10 is a diagram illustrating a configuration of a canceller unit according to a seventh embodiment. It is a figure which shows the interference signal component in a pilot carrier position. It is a figure which shows the structure of Embodiment 8 of the communication apparatus concerning this invention. It is a figure which shows an example of the timing of a preceding wave and a delay wave. FIG. 20 is a diagram illustrating control of a transmission path estimation unit according to an eighth embodiment. It is a figure which shows the structure of Embodiment 10 of the communication apparatus concerning this invention. It is a figure which shows the structure of Embodiment 9 of the communication apparatus concerning this invention. It is a figure which shows the structure of Embodiment 11 of the communication apparatus concerning this invention. 218 is a diagram illustrating a detailed configuration of a transmission path estimation / weighting coefficient calculation unit in Embodiment 11. [FIG. 218 is a diagram illustrating a detailed configuration of a transmission path estimation / weighting coefficient calculation unit in Embodiment 12. [FIG. It is a figure which shows the structure of Embodiment 13 of the communication apparatus concerning this invention. FIG. 26 is a diagram illustrating a configuration example for a canceller function unit to realize a canceller function illustrated in FIG. 25. 218 is a diagram illustrating a detailed configuration of a weight coefficient calculation unit in Embodiment 13. [FIG. FIG. 38 is a diagram illustrating a detailed configuration of a weighting factor calculation unit according to the fourteenth embodiment. It is a figure which shows the structure of Embodiment 15 of the communication apparatus concerning this invention. 3 is a diagram illustrating a configuration of a canceller function unit 126. FIG. It is a figure which shows one structural example of Embodiment 16 of the communication apparatus concerning this invention. It is a figure which shows one structural example of Embodiment 16 of the communication apparatus concerning this invention. It is a figure which shows one structural example of Embodiment 16 of the communication apparatus concerning this invention.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 FFT part 2 Symbol timing detection part 3 Transmission path estimation part 4 Demodulation part 5 Pilot signal generation part 6 Frequency characteristic calculation part 7 IFFT part 8 Interference removal part 9 Threshold setting part 10 FFT part 11 Pilot extraction part 12 Frequency characteristic interpolation Unit 13, 15 averaging unit 14 waveform shaping unit 21 canceller unit 22 reception determination unit 23 interference amount calculation unit 31 frequency characteristic memory unit 32 IFFT unit 33 distortion removal unit 34 determination unit 35 FFT unit 36 frequency characteristic update unit 41 level calculation unit 42 Maximum Value Detection Unit 43 Threshold Value Definition Unit 44 Comparison Unit 45 Gate Circuit 46 Waveform Addition Unit 47 Memory Unit 51 Frequency Characteristic Difference Calculation Unit 61 Remodulation Unit 62 Subtraction Unit 63 Frequency Signal Interpolation Unit 64 Subtraction Unit 101, 111 Receiving Antenna 102, 112 FFT unit 103, 113 Multiply unit 104, 11 4, 125, 126 Canceller function section 121 Symbol timing detection section 122 Transmission path estimation / weighting coefficient calculation section 123 Combining section 124 Weighting coefficient calculation section 127 Transmission path estimation value correction section 131, 141 Transmission path estimation section 132, 142 Remodulation section 133, 143 Subtraction unit 134, 144 SIR calculation unit 135, 145 SIR calculation unit 151 Weight coefficient calculation unit

Claims (30)

In a communication device on the receiving side that employs a multi-carrier modulation / demodulation method and demodulates a received signal in which a known signal is repeatedly inserted at a fixed period,
A frequency characteristic calculating means for calculating a frequency characteristic based on the known signal extracted from the received signal after Fourier transform;
Delay profile generation means for generating a delay profile by performing inverse Fourier transform on the frequency characteristics;
Interference component removing means for removing an interference signal component (including noise) whose signal level is a certain value or less based on the delay profile;
A transmission path estimation value generating means for generating a frequency characteristic (transmission path estimation value) after removing the interference component by performing Fourier transform on the time axis signal based on the delay profile after the interference component removal;
Equipped with a,
The known signal is a scattered pilot signal inserted repeatedly on a time axis and a frequency axis at a constant period,
Further, remodulation means for performing remodulation of the channel estimation value and generating a desired signal at a pilot signal position;
First interference component generating means for generating an interference component at a pilot signal position by subtracting the desired signal from the received signal;
Second interference component generation means for generating an interference component at a data position by interpolation processing on the interference component at the pilot signal position;
Desired signal generating means for generating a desired signal by subtracting all interference components from the received signal;
Canceller means, including
Communication apparatus according to claim Rukoto equipped with. Furthermore, the timing of Fourier transform of the received signal is controlled so that the number of leaked samples is within the interpolation range determined by the pilot signal insertion interval based on the delay profile. The communication apparatus according to 1 . Further, the timing of the Fourier transform of the received signal is controlled based on the delay profile so that the amount of interference or SIR (signal power to interference signal power ratio) within the Fourier transform window is minimized. The communication apparatus according to claim 1 . Furthermore, the timing of Fourier transform of the received signal is controlled based on the delay profile so that the interference signal power value or SIR (signal power to interference signal power ratio) of the canceller means output is minimized. The communication device according to claim 1 . 5. The communication apparatus according to claim 2 , 3 or 4 , wherein processing of the interference component removing unit, processing of the canceller unit, and timing control of the Fourier transform are stopped according to a current reception state. The communication apparatus according to claim 1 , wherein the processing of the interference component removing unit and the processing of the canceller unit are stopped according to a current reception state. The delay profile generation means includes
Using the first threshold value determined to be an optimum value, the distortion of the time axis signal after the inverse Fourier transform is removed (first time), and the reference value defined in advance by the first threshold value is used. Outputs the time axis signal after distortion removal as a delay profile when it is smaller than the threshold,
When the first threshold value is larger than the reference threshold value, a Fourier transform is performed on the time axis signal after the distortion removal to obtain a frequency characteristic again, and the frequency characteristic and the inverse of the frequency characteristic are obtained. Update the frequency characteristics using the difference from the frequency characteristics before the Fourier transform,
Then, the inverse Fourier transform is performed again on the updated frequency characteristic, and the distortion of the time axis signal after the inverse Fourier transform is removed using the second threshold value newly determined as the optimum value (2 And the time axis signal after the distortion removal and the time axis signal after the first distortion removal are combined, and the time axis after the combination when the second threshold value is smaller than the reference threshold value. Output the signal as a delay profile,
When the second threshold value is larger than the reference threshold value, the frequency characteristic update process, the distortion removal process, and the synthesis process are repeatedly executed using the synthesized time-axis signal, and each time is determined. at the stage where the threshold is becomes smaller than the reference threshold, the communication device according to any one of claims 1-6, characterized in that outputs a time domain signal at that time as a delay profile . In a communication device on the receiving side that employs a multi-carrier modulation / demodulation method and demodulates a received signal in which a known signal is repeatedly inserted at a constant period,
  A frequency characteristic calculating means for calculating a frequency characteristic based on the known signal extracted from the received signal after Fourier transform;
  Delay profile generation means for generating a delay profile by performing inverse Fourier transform on the frequency characteristics;
  Interference component removing means for removing an interference signal component (including noise) having a signal level equal to or lower than a predetermined value based on the delay profile;
  A transmission path estimation value generating means for generating a frequency characteristic (transmission path estimation value) after removing the interference component by performing Fourier transform on the time axis signal based on the delay profile after the interference component removal;
  With
  The delay profile generation means includes
  Using the first threshold value determined to be the optimum value, the distortion of the time axis signal after the inverse Fourier transform is removed (first time), and the first threshold value is a standard defined in advance. Outputs the time axis signal after distortion removal as a delay profile when it is smaller than the threshold,
  When the first threshold value is larger than the reference threshold value, a Fourier transform is performed on the time axis signal after the distortion removal to obtain a frequency characteristic again, and the frequency characteristic and the inverse of the frequency characteristic are obtained. Update the frequency characteristics using the difference from the frequency characteristics before the Fourier transform,
  Then, the inverse Fourier transform is performed again on the updated frequency characteristics, and the distortion of the time axis signal after the inverse Fourier transform is removed using the second threshold value newly determined as the optimum value (2 And the time axis signal after the distortion removal and the time axis signal after the first distortion removal are combined, and the time axis after the combination when the second threshold value is smaller than the reference threshold value. Output the signal as a delay profile,
  When the second threshold value is larger than the reference threshold value, the frequency characteristic update process, the distortion removal process, and the synthesis process are repeatedly executed using the time axis signal after the synthesis, and determined each time. When the threshold value to be set is smaller than the reference threshold value, the time axis signal at that time is output as a delay profile. Furthermore, the frequency characteristic calculation means performs interpolation processing on the frequency characteristics, according to any one of claims 1-8, characterized in that determines and outputs the frequency characteristic of the entire signal band Communication device. Furthermore, the frequency characteristic calculation means, the communication device according to any one of claims 1-9, characterized by outputting the frequency characteristic by temporally averaged. Furthermore, the said interference component removal means averages the time-axis signal after interference removal, and outputs it, The communication apparatus as described in any one of Claims 1-10 characterized by the above-mentioned. The communication apparatus according to any one of claims 1 to 11, characterized by employing the OFDM (Orthogonal Frequency Division Multiplexing) scheme as the multi-carrier modulation and demodulation system. The communication apparatus according to any one of claims 1 to 11, characterized by employing a multi-carrier CDMA (Code Division Multiple Access) scheme as the multi-carrier modulation and demodulation system. In a communication apparatus that employs a multi-carrier modulation / demodulation method and demodulates a received signal in which a known signal is repeatedly inserted at a constant period using a diversity combining reception function,
For each receiving system (branch)
A frequency characteristic calculating means for calculating a frequency characteristic based on the known signal extracted from a received signal (received signal for each branch) after Fourier transform;
Delay profile generation means for generating a delay profile by performing inverse Fourier transform on the frequency characteristics;
Interference component removing means for removing an interference signal component (including noise) whose signal level is a certain value or less based on the delay profile;
A transmission path estimation value generating means for generating a frequency characteristic (transmission path estimation value) after removing the interference component by performing Fourier transform on the time axis signal based on the delay profile after the interference component removal;
SIR calculation means for calculating SIR (signal power to interference signal power ratio) over the entire signal band based on the transmission path estimation value;
With
Further, a weighting factor of each branch is calculated based on the SIR of the entire signal band output from the SIR calculation means of each branch and the transmission path estimation value output from the transmission path estimation value generation means of each branch. Weighting factor calculation means;
With
Multiplying the corresponding weight signal of the branch by the corresponding reception signal for each branch, further combining the multiplication results, demodulating the combination result ,
Also,
The SIR calculation means is
Remodulating means in the SIR calculation means for performing remodulation of the transmission path estimated value and generating a desired signal at a known signal position;
An interference component generating means in SIR computing means for generating an interference component at a known signal position by subtracting the desired signal from the received signal for each branch;
The SIR at the known signal position is calculated using the interference component at the known signal position, and the SIR of the data signal position is obtained by interpolation using the SIR at the known signal position. SIR calculating means for calculating
A communication apparatus comprising: 15. The communication apparatus according to claim 14 , wherein the frequency characteristic calculation unit performs an interpolation process on the frequency characteristic to obtain and output the frequency characteristic of the entire signal band. The communication device according to claim 14 or 15 , wherein the frequency characteristic calculation means averages the frequency characteristic over time and outputs the averaged frequency characteristic. The delay profile generation means includes
Using the first threshold value determined to be an optimum value, the distortion of the time axis signal after the inverse Fourier transform is removed (first time), and the reference value defined in advance by the first threshold value is used. Outputs the time axis signal after distortion removal as a delay profile when it is smaller than the threshold,
When the first threshold value is larger than the reference threshold value, a Fourier transform is performed on the time axis signal after the distortion removal to obtain a frequency characteristic again, and the frequency characteristic and the inverse of the frequency characteristic are obtained. Update the frequency characteristics using the difference from the frequency characteristics before the Fourier transform,
Then, the inverse Fourier transform is performed again on the updated frequency characteristic, and the distortion of the time axis signal after the inverse Fourier transform is removed using the second threshold value newly determined as the optimum value (2 And the time axis signal after the distortion removal and the time axis signal after the first distortion removal are combined, and the time axis after the combination when the second threshold value is smaller than the reference threshold value. Output the signal as a delay profile,
When the second threshold value is larger than the reference threshold value, the frequency characteristic update process, the distortion removal process, and the synthesis process are repeatedly executed using the synthesized time-axis signal, and each time is determined. 17. The communication device according to claim 14, wherein the time axis signal at that time is output as a delay profile when the threshold value to be set is smaller than the reference threshold value. The communication device according to any one of claims 14 to 17 , wherein the interference component removal means averages and outputs the time axis signal after interference removal. In addition, for each branch,
Remodulating means for performing remodulation of the transmission path estimated value and generating a desired signal at a known signal position;
First interference component generation means for generating an interference component at a known signal position by subtracting the desired signal from the received signal for each branch;
Second interference component generation means for generating an interference component at a data position by interpolation processing on the interference component at the known signal position;
Desired signal generating means for generating a desired signal by subtracting interference components of all frequency bands from the received signal for each branch;
Canceller means, including
With
One of claims 14 to 18, the weight coefficient of the branches, multiplies the output signal of the corresponding canceller means further combines the multiplication results, wherein the demodulating the combined result The communication apparatus as described in. In addition, for each branch,
Based on the delay profile, the as number of samples leak into the interpolation range which is determined by the insertion interval of the known signal falls, claim 19, characterized in that to control the timing of the Fourier transform of the received signal The communication apparatus as described in. In addition, for each branch,
20. The communication apparatus according to claim 19 , wherein the timing of Fourier transform on the received signal is controlled based on the delay profile so that the amount of interference in the Fourier transform window is minimized. In addition, for each branch,
20. The communication apparatus according to claim 19 , wherein the timing of Fourier transform for the received signal is controlled based on the delay profile so that an interference signal power value of the canceller means output is minimized. In addition, for each branch,
Depending on the current reception situation, the interference process component removing means, a communication device according to claim 20, 21 or 22, characterized in that to stop the process and timing control of the Fourier transform of the canceller unit. In addition, for each branch,
The communication apparatus according to claim 19 , wherein the processing of the interference component removing unit and the processing of the canceller unit are stopped according to a current reception state. In addition, for each branch,
The communication device according to any one of claims 14 to 18 , wherein the processing of the interference component removing unit is stopped according to a current reception state. further,
Remodulating means for remodulating the transmission line estimation value of each branch and generating a desired signal at a known signal position;
First interference component generation means for generating an interference component at a known signal position by subtracting the desired signal from the combined result after the weighting factor multiplication;
Second interference component generation means for generating an interference component at a data position by interpolation processing on the interference component at the known signal position;
Desired signal generating means for generating a desired signal by subtracting interference components of all frequency bands from the combined result after the weighting factor multiplication,
Canceller means, including
With
The communication apparatus according to any one of claims 14 to 18 , wherein the output of the canceller means is demodulated. Further, the timing of Fourier transform of the received signal is controlled based on the delay profile so that the number of leaked samples is within an interpolable range determined by an insertion interval of the known signal. Item 27. The communication device according to Item 26 . 27. The communication apparatus according to claim 26 , further comprising: controlling a Fourier transform timing of the received signal based on the delay profile so that an interference amount in a Fourier transform window is minimized. 27. The communication apparatus according to claim 26 , further comprising: controlling a Fourier transform timing of the received signal based on the delay profile so that an interference signal power value of the canceller means output is minimized. In a communication apparatus that employs a multi-carrier modulation / demodulation method and demodulates a received signal in which a known signal is repeatedly inserted at a constant period using a diversity combining reception function,
Combining means for synthesizing the received signal after Fourier transform for each receiving system (branch);
A frequency characteristic calculating means for calculating a frequency characteristic based on the known signal extracted from the synthesized signal;
Delay profile generation means for generating a delay profile by performing inverse Fourier transform on the frequency characteristics;
Interference component removing means for removing an interference signal component (including noise) whose signal level is a certain value or less based on the delay profile;
A transmission path estimation value generating means for generating a frequency characteristic (transmission path estimation value) after removing the interference component by performing Fourier transform on the time axis signal based on the delay profile after the interference component removal;
Equipped with a,
Furthermore, as an interference cancellation function,
Remodulating means for performing remodulation of the transmission path estimated value and generating a desired signal at a known signal position;
First interference component generation means for generating an interference component at a known signal position by subtracting the desired signal from the combined signal;
Second interference component generation means for generating an interference component at a data position by interpolation processing on the interference component at the known signal position;
Desired signal generating means for generating a desired signal by subtracting interference components of all frequency bands from the combined signal;
With
A communication apparatus for demodulating a desired signal output from the desired signal generating means.

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