JP4510569B2 - Digital converter and control method thereof - Google Patents

Description

  The present invention relates to a digital converter using a step-up chopper, and more particularly to a device that can realize precise PWM control even when a general coil is used, and can appropriately cope with load fluctuations, and a control method therefor. .

The applicant has already applied for a similar application that can realize precise PWM control (Patent Document 1). This patent application proposes a digital converter capable of precise control over a wide range of input current while having a simple configuration that can also use inexpensive general coils, and a control cycle that is repeated at a frequency of about 20 KHz. Every time, the coil input current, the AC input voltage, and the DC output voltage are measured, and the control ON time of the PWM control in the next control cycle is determined based on the measured values.
Japanese Patent Application No. 2004-268135

  However, in the above-described invention, there is room for further improvement in measures against a sudden load change or the like. That is, in the above invention, because a part of the control algorithm is configured by PI control, it is not always possible to respond quickly when the load suddenly changes, and the output DC voltage may greatly exceed the reference value. there were.

  Such overshoot will eventually be resolved with the passage of time, but since there is a possibility of destruction of the switching element or the like during that time, it is desirable to take appropriate measures. As a countermeasure, it is conceivable to increase the capacity of the smoothing capacitor provided on the output side. However, this has a greater negative effect of increasing the size and cost of the apparatus.

  Also, in a transient state immediately after the power is turned on, the PWM wave duty ratio calculated by the control algorithm becomes large because the output voltage is low. If this calculated value is applied as it is, an overshoot similar to the above will occur. Problem arises. Such an overshoot problem may occur not only when the power is turned on, but also when returning from an instantaneous power failure.

  The present invention has been made in view of the above problems, and is characterized in that precise control is possible over a wide range of input current while having a simple configuration in which an inexpensive general coil can be used. It is an object of the present invention to provide a digital converter and a control method therefor that are sufficient to take measures such as sudden load changes and power on.

  In order to achieve the above object, an invention according to claim 1 is directed to a step-up chopper provided with a coil and a switching element, a rectifier circuit for supplying an input current to the coil, and PWM control of the switching element in a predetermined control cycle. In a steady state, a digital converter having a computer circuit that determines whether the input current to the coil is a continuous mode in which the current is not interrupted during the control cycle or a discontinuous mode in which the current is interrupted in the middle of the control cycle. On the other hand, when the output DC voltage increases and exceeds the upward limit value, the PWM control is stopped and the output DC voltage is rapidly lowered when the output DC voltage increases and exceeds the upper limit value.

  In the present invention, whether or not the output DC voltage exceeds the upward limit value is monitored, and if it exceeds the upward reference value, the PWM control is stopped and the output DC voltage is forcibly dropped. The switching element or the like can be prevented from being destroyed without increasing the capacity of the smoothing capacitor provided in the circuit. An appropriate value is adopted as the upward limit value. Preferably, a value obtained by adding about twice the boost amount α in the coil to the reference value Vo (target value) of the output DC voltage (≈Vo + 2 × α). The boosting amount of the coil is preferably set to about 5 to 30V.

  The invention according to claim 2 includes a step-up chopper provided with a coil and a switching element, a rectifier circuit that supplies an input current to the coil, and a computer circuit that PWM-controls the switching element in a predetermined control cycle. It is a digital converter, and in steady state, it is determined whether the input current to the coil is a continuous mode in which the current is not interrupted during the control cycle or a discontinuous mode in which the current is interrupted in the middle of the control cycle. While the control is performed, in the transient state up to the steady state, the original control time calculated by the algorithm is corrected and the PWM control is performed.

  The transient state typically means a time period from when the computer circuit starts to operate until a predetermined time elapses. Here, “the computer circuit starts to operate” typically means a state in which the CPU is reset, and this corresponds to the time when the power is turned on or the time of recovery from an instantaneous power failure. The “state in which the computer circuit starts operating” may include a state in which a CPU is reset by starting a watchdog timer that monitors a program runaway state.

  Another typical example of the transient state is a time period from when the PWM control is stopped and the output DC voltage is lowered to fall below the downward limit value until a predetermined time elapses. In order to generate such a transient state, it is necessary to precede the operation for stopping the PWM control. However, as the preceding operation, the PWM control is performed on the condition that the output DC voltage increases and exceeds the upward limit value. It is effective to stop. Here, stopping the PWM control means maintaining the switching element in the OFF state. For simplicity, the same value as the upward limit value is selected as the downward limit value.

  In any case, in the transient state, it is preferable to employ a correction algorithm that approaches the original control time as time passes. As the correction algorithm, the maximum value of the control time (time for specifying the duty ratio of the PWM wave) is specified corresponding to the elapsed time, and the calculated original control time is set not to exceed the maximum value. Limit it.

  The present invention is also a control method for realizing the operation of each of the above inventions. In other words, the present invention is a control method for a digital converter including the technical elements according to claims 1 to 3.

  In each of the above inventions, the determination of whether the mode is continuous mode or discontinuous mode is performed by determining whether the coil charging start current Iv (n−2), the AC input voltage Vac (n−1) to the boost chopper, and the boost chopper in the measurement cycle. It is preferably determined based on each measured value of the DC output voltage Vdc (n−1), the control time of the PMW wave in the measurement cycle, and the inductance value of the coil. In this case, it is preferable that the inductance value is corrected corresponding to the measured value of the coil current in the measurement cycle.

  In the present invention, preferably, the control time Ton (n) of the PMW wave in the next and subsequent control cycles is determined using an arithmetic expression corresponding to the determination result of the continuous mode or the discontinuous mode. Further, it is preferable that the control time Ton (n) of the PMW wave in the subsequent control cycle is determined by using the predicted value Iav (n) of the coil average current in the next control cycle as an arithmetic element. Furthermore, the predicted value Iav (n) of the coil average current is determined by the integration of the predicted value Vac (n) of the AC input voltage and the integration parameter β, and the integration parameter β is the target of the DC output voltage Vdc. It is preferably determined by PI (Proportional-Integral) control based on a deviation Verr with respect to the value Vo.

  According to the present invention described above, it is possible to perform precise PWM control over a wide range of input current with a simple configuration. In addition, even if a large-capacity smoothing capacitor is not used, it is possible to appropriately cope with load fluctuations and the like, and destruction of the switching element due to overshoot is prevented.

  Hereinafter, the present invention will be described in more detail based on examples. FIG. 1 is a circuit configuration diagram showing a digital converter under software control. The digital converter of the embodiment is incorporated as a part of a motor control system. That is, in the converter 1 of this embodiment, a single-phase AC voltage (for example, 200 V) is rectified by the full-wave rectifier circuit 2 to become a pulsating current, and then boosted by PWM (Pulse Width Modulation) control by the one-chip microcomputer 3 The chopper 4 converts the voltage into a predetermined DC voltage Vdc (for example, 350 V). The three-phase motor M is driven by an inverter circuit 5 controlled by the one-chip microcomputer 3.

  The step-up chopper 4 includes a coil L, a capacitor C, a switching element Q, and a shunt resistor r for current detection. The coil L, the capacitor C, and a shunt resistor r are connected in series, and the capacitor C is short-circuited to the capacitor C. Switching elements Q to be connected are connected in parallel. The input current of the boost chopper 4 flows toward the coil L and returns through the shunt resistor r. In this embodiment, an insulated gate bipolar transistor IGBT (Insulated Gate Bipolar Transistor) is used as the switching element Q.

  As shown in the figure, the input current I, the input AC voltage Vac, and the output DC voltage Vdc are input to the one-chip microcomputer 3 through the signal input units IN1 to IN3, respectively, and the built-in A / D converters AD1 to AD4 are used. , Each has been digitally converted. The A / D converter AD1 and the A / D converter AD4 acquire the same input current I at different timings. Here, the signal input unit IN1 is composed of an OP amplifier circuit that receives the voltage across the shunt resistor r, and functions as a current detection sensor together with the shunt resistor r. Further, the signal input unit IN2 and the signal input unit IN3 are configured by a resistance voltage dividing circuit and an OP amplifier amplifier circuit.

  The data acquired through the signal input units IN1 to IN3 and the A / D converters AD1 to AD4 are processed by the one-chip microcomputer 3 to calculate the PWM signal ON time (hereinafter referred to as control on time). The generated PWM signal is supplied from the output port PT to the gate of the switching element Q through the signal output unit OUT1. The one-chip microcomputer 3 is connected to the inverter circuit 5 through the signal input unit IN4 and the signal output unit OUT2, and controls the three-phase motor M by inverter.

  Hereinafter, the operation principle of this embodiment in which the switching element Q is ON / OFF controlled by the PWM wave will be described. FIG. 2 is a time chart illustrating the relationship between the PWM wave output from the one-chip microcomputer 3 and the current flowing through the coil L. As shown in the figure, a triangular wave due to the coil charging current and the coil discharging current flows through the coil L, but the energy stored in the coil L is sufficient and the continuous current flows (see FIG. 2A). Since there is insufficient energy, there is a discontinuous mode (see FIG. 2B) in which the current is interrupted in the middle.

  In this embodiment, since different PWM control is performed depending on which operation mode is in use, first, a determination formula for determining the operation mode will be described. In the time chart of FIG. 2, it is assumed that the current time is the control cycle (n−1). Then, it is considered that the control on time Ton (n) in the next control cycle (n) is determined based on the measured value in the control cycle (n−1). Note that the frequency of the AC input voltage is 50 Hz or 60 Hz, but in order to control it sufficiently quickly, in this embodiment, the control cycle T is set to 50 μS. In addition, although the measurement cycle and the non-measurement cycle are repeated in the control cycle of the embodiment (see FIG. 13), since it is a principle explanation, hereinafter, all the control cycles are treated as measurement cycles.

The circuit equation during coil charging (switching element ON) is
Vac (n−1) = L × {Ip (n−1) −Iv (n−2)} / Ton (n−1) (Expression 1) Here, Iv (n-2) is a coil charging start current, Ip (n-1) is a coil charging peak current, and Ton (n-1) is a control ON time, and each value in the control cycle (n-1). It is. Vac (n-1) is an input voltage in the control cycle (n-1). Since the control cycle (50 μS) is sufficiently short with respect to the power supply frequency, Vac (n-1) is regarded as a constant value. Can do.

On the other hand, the circuit equation at the time of coil discharge (switching element OFF) is
Vdc (n−1) −Vac (n−1) = L × {Ip (n−1) −Iv (n−1)} / Toff (n−1) (Expression 2) Here, Vdc (n-1) is the voltage across the capacitor C, Iv (n-1) is the coil current at the end of the current control cycle (coil charging start current of the next control cycle), Toff (n-1). Are OFF times, which are values in the control cycle (n−1), respectively.

If Ip (n-1) is eliminated from (Equation 1) and (Equation 2) and Iv (n-1) is solved,
Iv (n−1) = Iv (n−2) + T / L × [{Vac (n−1) −Vdc (n−1)} + Ton (n−1) × Vdc (n−1)]... (Equation 3) The control period T is T = Ton (n−1) + Toff (n−1).

  In the above (Formula 3), if Iv (n−1)> 0, the continuous mode is selected, and if Iv (n−1) = 0, the discontinuous mode is selected. However, (Formula 3) uses the coil charging start current Iv (n-2) in the current control cycle (n-1), and the coil charging start current Iv (n-1) in the next control cycle (n). Therefore, if the charging start current Iv (n−2) is not accurate, the determination of the continuous mode or the discontinuous mode will be wrong. That is, if Iv (n−2) is determined by a prediction calculation based on the control on time (Ton (n−2)) in the immediately preceding control cycle, an error is accumulated by the calculation of (Equation 3). As a result, the control instruction value itself may diverge from the target. Therefore, in this embodiment, the input current ad1 at the start of coil charging is measured for each control cycle.

  However, since a time delay Ts inevitably occurs from the start of the control cycle to the time of acquisition of the input current (see FIG. 4B), the coil charge is started by correcting the measured value ad1 in consideration of this time delay Ts. The current is Iv (n−2). Assuming that the current measurement value at the timing Ts of the control cycle (n−1) is ad1, the slope (Δi / Δt) of the coil charging current is Δi / Δt = e / from the relationship of L × Δi / Δt = e. L. Here, e is voltage, i is current, and t is time.

Therefore, the current increase amount at the time delay Ts can be calculated as Vac (n−1) / L × Ts, and using this value, Iv (n−2) = ad1−Vac (n−1) / L × Ts (Expression 4) And if this (Equation 4) is substituted into (Equation 3),
Iv (n-1) = ad1- [Vac (n-1) * Ts + T * {Vdc (n-1) -Vac (n-1)}-Ton (n-1) * Vdc (n-1)] / L (Expression 5), and the input current value Iv (n−1) at the final timing of the current control cycle (n−1) (= start timing of the next control cycle) is expressed as the current control cycle (n -1) can be accurately determined based on the measured value ad1 of the input current at the start timing.

  Then, depending on whether or not the coil charging start current Iv (n-1) of the next control cycle obtained in this way is positive, it is possible to accurately determine whether it is a continuous mode or a discontinuous mode, and an optimal control corresponding to that mode. Is possible. That is, the measured values of the AC input voltage Vac (n-1), the DC output voltage Vdc (n-1), and the input current ad1 in the current control cycle (n-1) and the previous control cycle are determined. On the basis of the control on time Ton (n−1), it can be determined whether the control for the continuous mode or the control for the discontinuous mode should be performed. In addition, the calculation procedure of the above (Formula 1)-(Formula 5) is supplementary description to FIGS.

  By the way, the DC output voltage Vdc (n−1) does not necessarily need to be updated every control cycle. Therefore, in this embodiment, Vdc (i) whose value is updated every 1 mS is used ( (See step ST30 in FIG. 5). Therefore, the discriminant of the present embodiment is, exactly, Iv (n-1) = ad1- [Vac (n-1) * Ts + T * {Vdc (i) -Vac (n-1)}-Ton (n −1) × Vdc (i)] / L (Expression 5 ′).

  Furthermore, the average value Vdc for the past 0.5 seconds calculated by the process of step ST38 in FIG. 5 may be used as the DC output voltage. In this case, Iv (n−1) = ad1− [Vac (n−1) × Ts + T × {Vdc−Vac (n−1)} − Ton (n−1) × Vdc] / L. The discriminant of equation 5 ″) is adopted.

  Next, a method for calculating the control on time Ton (n) based on the coil average current Iav during each control cycle will be described. First, the control on time Ton (n) in the discontinuous mode is calculated (see FIG. 2B).

<Discontinuous mode>
The circuit equation at the time of charging the coil is Vac (n) = L × {Ip (n) −Iv (n−1)} / Ton (n), but Iv (n−1) = 0 because of the discontinuous mode. In the end, Vac (n) = L × Ip (n) / Ton (n) (Formula 6). Here, Vac (n) is the AC input voltage, Ip (n) is the current value at the coil charging peak, Iv (n−1) is the current value at the start of coil charging, and Ton (n) is the control on time. .

On the other hand, the circuit equation at the time of coil discharge is Vdc (n) −Vac (n) = L / Tcut (n) × {Ip (n) −Iv (n)}. Tcut (n) is a time until the current value Ip (n) in the coil charge peak state is discharged and becomes zero (see FIG. 2B). Since the circuit equation in the discontinuous mode is considered here, Iv (n) = 0, and Vdc (n) −Vac (n) = L / Tcut (n) × Ip (n) (Expression 7) ) The average value Iav (n) of the input current in this control cycle is
Iav (n) = {Ip (n) × Ton (n) + Ip (n) × Tcut (n)} / (2 × T) (Expression 8).

And solving these (Equation 6) to (Equation 8) for Ton (n),
Ton (n) × Ton (n) = {2 × T × L × Iav (n) × (Vdc (n) −Vac (n))} / {Vac (n) × Vdc (n)} Equation 9) is calculated. In addition, the calculation process of (Formula 9) was shown in FIGS.

  Subsequently, the control ON time Ton (n) in the continuous mode is calculated (see FIG. 2A).

<Continuous mode>
The circuit equation during coil charging is Vac (n) = L × {Ip (n) −Iv (n−1)} / Ton (n) (Equation 10) On the other hand, the circuit equation at the time of coil discharge is Vdc (n) −Vac (n) = L / Toff (n) × {Ip (n) −Iv (n)} (Equation 11). Here, Toff (n) = T−Ton (n), which is the time from the start of coil discharge to the start of coil charge in the next control cycle.

The average current Iav (n) in this control cycle is Iav (n) =
[{Ip (n) + Iv (n−1)} × Ton (n) + {Ip (n) + Iv (n)} × Toff (n)] / {2 × T} (Expression 12) . Here, solving (Equation 10) to (Equation 12) for Toff (n) while erasing Ip (n) and Iv (n),
Toff (n) × Toff (n) = [2 × T × L × {Iv (n−1) −Iav (n)} / Vdc (n)] + T × T × Vac (n) / Vdc (n) · Since (Equation 13) is obtained, Ton (n) is eventually calculated as Ton = T−Toff (n) (Equation 14). Note that (Equation 14) and other calculation processes are supplemented in FIGS.

  In this embodiment, the control on-time Ton (n) is calculated using (Equation 9) or (Equation 14) according to the discontinuous mode or the continuous mode. Prediction parameters for the AC input voltage Vac (n), the DC output voltage Vdc (n), and the average input current Iav (n) in (n) are required.

The AC input voltage Vac (n) is predicted based on the current AC input voltage measurement value Vac (n-1) and the previous AC input voltage measurement value Vac (n-2). The difference between the control cycle (n-2) and the control cycle (n-1) measurement value is added to the current measurement value Vac (n-1) as follows.
Vac (n) = 2 × Vac (n−1) −Vac (n−2) (Equation 15)
On the other hand, for the DC output voltage Vdc (n), an average value Vdc of past measurement values for the DC voltage is adopted. Although the calculation method of the average value Vdc is determined as appropriate, in this embodiment, the measured values for the past 0.5 seconds are averaged by the averaging process executed every 0.5 seconds to obtain the DC output voltage Vdc. (See step ST38 in FIG. 5). This DC output voltage Vdc is stored in an appropriate work area of the memory, and the value Vdc of this work area is updated every 0.5 seconds.

  Therefore, in this case, in the discontinuous mode, Ton (n) × Ton (n) = {2 × T × L × Iav (n) × (Vdc−Vac (n))} / {Vac (n) × On the other hand, in continuous mode, Toff (n) × Toff (n) = [2 × T × L × {Iv (n−1) −Iav (n)} / Vdc. ] + T × T × Vac (n) / Vdc (Equation 13 ′), Ton = T−Toff (n) (Equation 14 ′).

  However, instead of using the average value Vdc for 0.5 seconds, the value of Vdc (i) obtained from the output value ad3 of the AD converter AD3 every 1 mS may be used. In this case, in the discontinuous mode, Ton (n) × Ton (n) = {2 × T × L × Iav (n) × (Vdc (i) −Vac (n))} / {Vac (n) × Vdc (i)} (Equation 9 ″) On the other hand, in continuous mode, Toff (n) × Toff (n) = [2 × T × L × {Iv (n−1) −Iav ( n)} / Vdc (i)] + T × T × Vac (n) / Vdc (i) (Equation 13 ″), Ton = T−Toff (n) (Equation 14 ″) Become. In FIG. 3, for convenience, the case where Vdc (i) is used is indicated by a solid line, and the case where the DC output voltage Vdc is used is indicated by a broken line.

  The predicted value of the average input current Iav (n) is Iav (n) = β × Vac (n) from the relationship with the predicted value of the AC input voltage Vac (n). Here, the gain β is adjusted by PI control so that the difference Verr becomes zero while comparing the reference value (target value) Vo of the DC output voltage with the above-mentioned averaged DC output voltage Vdc.

  That is, Verr = Vo−Vdc (Expression 16), and Vo = Vac (pk) + α (Expression 17). Here, the reference value Vo of the DC output voltage is set to a value obtained by adding the amount of boost α by the coil L to the peak value Vac (pk) of the AC input voltage (pulsating flow). By setting in this way, conversion with high efficiency according to the input voltage value is possible. In addition, since the amount of pressure boosted by the coil L can be reduced, it is possible to select an inexpensive and lightweight coil with an appropriate size that does not increase in size. The optimum inductance value of the coil L is generally determined based on a design formula of L = Vac × Vac × (Vdc−Vac) / {η × Pac × Vdc / T}. Since the output voltage value Vdc is set in correspondence with the voltage value Vac, the inductance value of the coil always maintains an almost optimum value. In the above design formula, η is the ripple content of the input current, T is the control period, Pac is the maximum input power, and Vac is the instantaneous value of the input voltage.

  FIG. 3 is a control block diagram for explaining the above-described control operation. 4 to 6 are flowcharts showing the processing contents of the one-chip microcomputer 3 that realizes the control operation of FIG. In order to briefly explain the operation principle so far, each control cycle has been dealt with without distinction, but in the PWM control of this embodiment, there is only one measurement cycle in two control cycles, and the measurement result The control on time is updated based on the above. FIG. 13 illustrates the relationship. In a measurement cycle (n−1, n + 1,...) That is an odd-numbered control cycle, necessary data is measured, and control is performed based on the measured value. On-time Ton (n + 1) and Ton (n + 3) are determined. On the other hand, in the even-numbered control cycle (n−2, n, n + 2,...), The same control ON time as that of the previous control cycle is used.

  Based on the above, the control process shown in FIG. 3 includes the first timer interrupt INT1 (FIG. 4A) activated every 50 μS and the second timer interrupt INT2 activated every 1 mS (FIG. 5). ) And an AD conversion end interrupt INT3 (FIG. 6) which is activated when the AD conversion operation started by the first timer interrupt INT1 is completed. Note that the first timer and the second timer are asynchronous, and the first timer interrupt INT1 is set to a higher priority than the second timer interrupt INT2. Hereinafter, the operation content of the one-chip microcomputer 3 will be described with reference to FIGS.

<First Timer Interrupt INT1 (FIG. 4 (a))>
As shown in FIG. 4A, when the first timer interrupt INT1 occurs, after the register saving process (POP process), the interrupt request flag clearing process, and the like are finished (ST1), the PWM signal is set to the H level. At the same time as starting up, a down-counting operation of the set value Ton (n−1) of the PWM timer determined based on the measured value in the previous measurement cycle or the previous measurement cycle is started (ST2). This down-counting operation then proceeds independently of the processing of FIGS. 4 to 6, and PWM control is realized by returning the PWM signal to the L level after a predetermined time Ton (n−1) has elapsed. (See FIG. 4 (b)).

  Next, the one-chip microcomputer 3 checks the value of the measurement request flag FG (ST3), and if FG = 0, sets the measurement request flag FG to 1 and ends the interrupt process. On the other hand, if FG = 1, the measurement request flag FG is set to 0 and the process proceeds to step 6. In step ST6, the 4ch AD converters AD1 to AD4 are operated in the continuous scan mode to start the AD conversion process, and then the interrupt process is finished. Here, when the AD conversion process executed in the continuous scan mode is completed, the CPU of the one-chip microcomputer 3 is set to receive an AD conversion end interrupt. As shown in FIG. 4B, there is a time delay Ts from the start of the control cycle by the interrupt of the first timer until the start of the first AD conversion, and from the start of the first AD conversion. There is a delay time Td until the start of the fourth AD conversion. The time delay Ts is, for example, about 2.2 μS, and the delay time Td is, for example, about 20 μS.

  As is clear from the operations in steps ST3 to ST6, the measurement request flag FG is a flag for determining whether or not to start the measurement process (ST6) by the AD converter, and is a cycle of 0 → 1 → 0 → 1. The operation is performed (see ST4 and ST5). Therefore, once every two control cycles is a measurement cycle, and the other one of the measurement request flags FG = 0, the first timer interrupt process is terminated without starting the AD conversion process. However, the PWM control is executed by the process of step ST2.

<AD conversion end interrupt INT3 (FIG. 6)>
When the AD conversion operation is completed for all the AD converters after the process of step ST6, the control operation is executed by the interrupt process INT3 shown in FIG. First, the output values ad1, ad4 (input current to the coil L) of the AD converters AD1, AD4 are acquired (ST10). Next, the average value Iav (n−1) of the input current in the control cycle (n−1) is calculated by the average calculation (ad1 + ad4) / 2 (ST11). As shown in FIG. 4B, the input current value varies depending on the sampling point, so the accuracy as the average value of the input current is not high, but this average value Iav (n−1) Since it is only used for correcting the inductance value described, no particular problem occurs.

  When the average value Iav (n−1) of the input current is obtained, the inductance value of the coil L mounted on the circuit is specified based on the current value Iav (n−1). As shown in FIG. 14, the inductance value of the coil L often changes in accordance with the DC superimposed current (average current) flowing therethrough. Therefore, in this embodiment, the characteristics of the coil L mounted in the circuit are stored in the memory in advance, and the inductance value corresponding to the average value Iav (n−1) of the input current is used in each arithmetic expression. I am doing so. Therefore, highly accurate control can be performed without using a large and expensive coil.

  In order to further increase the control accuracy, the current value Iav (n−1) obtained by the average calculation (ad1 + ad4) / 2 is used as the control on time Ton (n−1) in the control cycle (n−1). It may be corrected accordingly (see FIG. 3). That is, the two sampling points ad1 and ad4 may be divided into the coil charging current ad1 and the coil discharging current ad4, or the control on time Ton (n−1) may be long, and both may be the coil charging current. Therefore (see FIG. 4B), more accurate control can be realized if the average current is corrected based on this point.

  Subsequently, the output value ad2 (AC input voltage Vac (n-1)) of the AD converter AD2 is acquired (ST13). Then, Vac (n) is calculated based on the voltage prediction formula Vac (n) = 2 × Vac (n−1) −Vac (n−2) (ST14).

  Next, the output value ad3 (DC output voltage Vdc (n-1)) of the AD converter AD3 is acquired (ST15), and it is determined whether or not the output voltage value exceeds the limit value LMT (ST16). The limit value LMT is a value that determines whether or not the PWM control is temporarily stopped. In this embodiment, the limit value LMT is set to LMT = Vo + 20 based on the target value Vo of the output DC voltage. Note that stopping the PWM control means turning off the switching element in this embodiment.

  For example, when the converter load suddenly decreases, the DC output voltage Vdc (n−1) may greatly increase and exceed the limit value LMT. When the DC output voltage Vdc (n−1) exceeds the limit value LMT and Vdc (n−1)> = LMT in this way, the overshoot flag OVF is set to 1 (ST17) and set to the PWM timer. The control ON time Ton (n) is set to 0 and the interrupt process is finished (ST18). As a result, in the next control cycle, the switching element Q is always in an OFF state, and the output DC voltage Vdc decreases. As a result of the suspension of PWM control, the overshooted output DC voltage Vdc eventually falls below the limit value LMT.

  The determination of step ST16 is Vdc <LMT not only in the return state from the overshoot but also in the steady control state where the output DC voltage Vdc does not exceed the limit value LMT. Therefore, when it is determined that Vdc <LMT, it is determined whether or not the overshoot flag OVF is 1. This is to determine whether the current state is a return state from overshoot or a steady control state.

  If the overshoot flag OVF = 0 (steady control state), the process proceeds to step ST23. If the overshoot flag OVF = 1 (return state from overshoot), the restart flag RST is set to 1 (ST20). Further, the operation of the return timer BAK is started (ST21), and the overshoot flag OVF is reset to 0 (ST22). The return timer BAK measures a transient operation time (restart time) from the return from the overshoot state to the transition to the steady control state.

  Here, the return timer BAK may have a hardware configuration, but in this embodiment, the return timer BAK is realized by software. Specifically, the value of the timer variable BAK set to 200 in the process of step ST21 is set. Decrement processing (BAK ← BAK-1) is performed in the second timer interrupt (FIG. 5) that occurs every 1 mS (however, not shown in FIG. 5). That is, in this embodiment, a restart time of 200 mS is ensured by the timing process of the return timer BAK.

  Next, the input current command value Iav (n) is calculated by calculating Iav (n) = β × Vac (n) (ST23). Note that. The necessary value of the integration parameter β is updated every 1 mS in the second interrupt process INT2 shown in FIG. 5 and stored in an appropriate work area.

  Thereafter, the acquired value ad1 from the AD converter AD1, the inductance value L of the coil corrected in the process of step ST12, the AC input voltage value Vac (n−1) acquired in the process of step ST13, and this control cycle Based on the control ON time Ton (n−1) and the DC output voltage value Vdc (i), the coil charging start current Iv (n−1) is calculated based on the discriminant of (Expression 5 ′). (ST24). Then, according to the value (whether positive or not) of the coil charging start current Iv (n−1), it is determined whether the control should be controlled as the continuous mode or the discontinuous mode (ST25). As described above, the discriminant of (Expression 5 ″) may be used instead of (Expression 5 ′).

  If Iv (n−1) ≦ 0 and the discontinuous mode is set, Iv (n−1) = 0 is set (ST26), (Equation 9 ′) or (Equation 9 ″). ) To calculate the control on time Ton (n) (ST27). On the other hand, if Iv (n−1)> and the continuous mode is selected, the control on-time Ton (n) is calculated based on the arithmetic expression of (Expression 14 ′) (Expression 14 ″) (ST28). ). However, in this embodiment, since the same control ON time is assigned to two consecutive control cycles, in FIG. 13, Ton (n + 1) is displayed instead of Ton (n).

  If the current state is a transitional state until shifting to the steady control state, the calculated control on time Ton (n) is set to the upper limit value (upper limit) of the control on time that is defined in advance for each elapsed time. It is corrected according to the duty) (ST29). Next, irrespective of the presence or absence of correction, the control on time Ton (n) is set in the PWM timer and the interrupt process is finished (ST30). Note that the control on time Ton (n) set here is used in the process of step ST2 of the first timer interrupt to realize the PWM control of the switching element Q.

  FIG. 6C is a flowchart showing the correction process (ST29) according to the upper limit duty. In the correction process, first, it is determined whether or not the restart flag RST is 1 (ST40). If the restart flag RST is RST = 0, it means that the current state is a steady control state, so nothing is done. The subroutine processing is finished.

  On the other hand, if the restart flag RST is RST = 1, the value of the timer variable BAK that has started the timing process by the process of step ST21 is checked to determine whether the 200 mS restart time has expired (ST41). . If the restart time has expired, the restart flag RST is reset and the subroutine processing is terminated (ST43).

  Conversely, if it is determined in step ST41 that the current time is within the restart time, an upper limit duty corresponding to the elapsed time BAK from the overshoot return is acquired, and this upper limit duty γ (%) is taken into consideration. The control on time Ton (n) calculated in step ST27 or step ST28 is corrected (ST42). Specifically, if the calculated control on time Ton (n) is Ton (n) <γ × T, the control on time Ton (n) is used as it is, but Ton (n)> = γ If xT, the process Ton (n) ← γ × T is executed. Note that T is a control period and is 50 μS in this embodiment.

  The upper limit duty γ (%) is determined according to the elapsed time BAK from the overshoot return, and approaches 100% as the elapsed time BAK increases and approaches 200 mS. The upper limit duty γ may be calculated by an arithmetic expression using the elapsed time BAK as an element, or may be determined based on a reference table TBL as shown in FIG.

  As described above, in this embodiment, it is always checked whether or not the output DC voltage Vdc has greatly overshooted (ST16), and when an excessive overshoot is recognized, the PWM output is stopped (ST18). . Therefore, control delay due to PI control does not become a problem, and the output DC voltage can be quickly lowered.

  Further, in this embodiment, when the output DC voltage drops due to the stop of the PWM output, the restart operation for 200 mS is started (ST20 to 22), and the control on time Ton (n) is set according to the elapsed time BAK. It is close to the theoretical values shown in (Equation 9) and (Equation 14) (ST29). Therefore, as shown in FIG. 15, a smooth transition from the return from the overshoot state to the steady control state is realized.

<Second timer interrupt INT2>
Next, the second timer interrupt INT2 started every 1 mS independently of the above-described first timer interrupt processing INT1 and AD conversion end interrupt INT3 will be described based on the flowchart of FIG.

  In the second timer interrupt INT2, first, Vdc (i) that is the output of the AD converter AD3 is acquired (ST30). The AD converter AD3 performs an AD conversion operation every 50 μS, but the second timer interrupt INT2 acquires the value of the output DC voltage Vdc (i) output based on the first timer interrupt INT1. .

  Next, the calculation of SUM ← SUM + Vdc (i) is executed, and the acquired value of the output DC voltage Vdc (i) is added to the average calculation buffer SUM in the work area (ST31). Further, the AC input voltage Vac (i), which is the output of the AD converter AD2, is acquired (ST32), and the AC input voltage Vac (i) is compared with the peak value Vac (pk) stored in the memory. (ST33).

  If Vac (i)> Vac (pk), the maximum wave height value Vac (pk) stored in the memory is updated by calculating Vac (pk) ← Vac (i) (ST34). After obtaining the peak value Vac (pk) of the AC input voltage in this way, the counter CT is decremented (−1) (ST35), and it is determined whether or not the counter value CT is zero (ST36).

  Here, when the counter value CT becomes CT = 0, the average value Vdc of the output DC voltage Vdc (i) is obtained by multiplying the value of the average calculation buffer SUM by 1/500 (ST37). When the DC output voltage Vdc is obtained, the values of the average calculation buffer SUM and the counter CT are initialized (ST39), and the reference value (target value) Vo of the output DC voltage is calculated by calculating Vo ← Vac (pk) + α. (ST40). α is a boosted amount in the coil L when compared with the peak value Vac (pk) of the input AC voltage. Then, the difference between the output reference voltage Vo and the output average voltage Vdc obtained from the measured value is calculated (ST41). Specifically, Verr (i) ← Vo−Vdc is calculated.

  Based on the above result, the command value β by the PI control is calculated, and the second timer interrupt processing INT2 is finished (ST42). Here, the calculation of the command value β is based on an arithmetic expression of β = Verr (i) × Kp + {Verr (i) × Ki + Verr (i−1) ′ × Ki}, where Verr (i−1) ′ × Ki is , The previous integration control value (i−1), which is a value calculated as Verr (i−1) ′ × Ki = Verr (i−1) × Ki + Verr (i−2) ′ × Ki.

  The contents of the control processing described above are summarized in FIG. 3 as a control block diagram, and the processing without INT1 and INT2 means processing INT3 of AD conversion end interrupt (FIG. 6).

  As described above, the digital converter 1 according to the embodiment determines whether or not the output DC voltage exceeds the limit value for each control cycle. When the output value exceeds the limit value, the PWM control is temporarily stopped. . Therefore, even if sufficient control time is allocated to PWM control by generating a time margin by using the control ON time calculated once twice in succession, a sudden overshoot due to control delay does not occur. .

  Further, in the digital converter 1 according to the embodiment, after the PWM control is temporarily stopped, when the output DC voltage falls below the limit value, the limited PWM control is executed, so that the transition to the steady operation can be smoothly performed.

  By the way, in the above description, only the restart operation in the transient state at the time of recovery from the overshoot has been described, but the same restart operation is performed in the transient state immediately after the power is turned on or at the recovery from the instantaneous power failure. Preferably it is performed. FIG. 16 (a) is a flowchart for explaining the operation at the time of power-on or return after an instantaneous power failure, in other words, the operation after the CPU is reset.

  As shown in the figure, when the CPU is reset, after initialization processing such as setting of each register of the one-chip microcomputer 3, the restart flag RST is set to 1, and 200 is set to the timer variable BAK. . The timer variable BAK is decremented on the condition that the restart flag RST is 1 in the second timer interrupt INT2 that occurs every 1 mS. Note that the timer variable BAK and the restart flag RST are referred to in the AD conversion end interrupt INT3 (ST40, ST42), and the control ON time Ton (n) is corrected on the condition that RST = 1 and BAK> 0. Is done. As a result, only the first 200 mS becomes a control ON time shorter than the calculated time, and the occurrence of overshoot is prevented.

  In addition, in this embodiment, since no hardware control circuit exists and all can be realized by software control, the power factor correction circuit and the like can be omitted, and the size can be reduced at low cost. In addition, since a common coil is used and its inductance value is corrected in real time and reflected in the arithmetic expression, high-precision control is possible at a low cost.

It is a circuit block diagram which shows the digital converter which concerns on an Example. It is a time chart explaining the relationship between a control period and the charging / discharging operation | movement of a coil. It is a control block diagram explaining the control operation of the digital converter which concerns on an Example. It is a flowchart which shows the operation | movement content of a one-chip microcomputer. It is a flowchart which shows the operation | movement content of a one-chip microcomputer. It is a flowchart which shows the operation | movement content of a one-chip microcomputer. It is drawing explaining the derivation | leading-out process of (Formula 5). It is drawing explaining the derivation | leading-out process of (Formula 5). It is drawing explaining the derivation | leading-out process of (Formula 9). It is drawing explaining the derivation | leading-out process of (Formula 9). It is drawing explaining the derivation | leading-out process of (Formula 14). It is drawing explaining the derivation | leading-out process of (Formula 14). It is a time chart explaining the operation | movement content of the digital converter which concerns on an Example. It is a characteristic view which shows the relationship between the inductance value of a coil, and an average current. It is drawing explaining an overshoot and a soft start. It is a flowchart explaining the transient operation after CPU reset.

Explanation of symbols

L Coil Q Switching element (IGBT)
4 Boost chopper 2 Rectifier circuit (Single-phase full-wave rectifier circuit)
3 Computer circuit (one-chip microcomputer)
1 Digital converter

Claims (6)

A digital converter having a step-up chopper provided with a coil and a switching element, a rectifier circuit that supplies an input current to the coil, and a computer circuit that performs PWM control of the switching element in a predetermined control cycle,
In the steady state, it is determined whether the input current to the coil is a continuous mode in which the current is not interrupted during the control cycle or a discontinuous mode in which the current is interrupted in the middle of the control cycle, and PWM control is performed using a different algorithm based on the determination result.
A digital converter characterized in that when the output DC voltage increases and exceeds an upward limit, PWM control is stopped and the output DC voltage is quickly reduced. A digital converter having a step-up chopper provided with a coil and a switching element, a rectifier circuit that supplies an input current to the coil, and a computer circuit that performs PWM control of the switching element in a predetermined control cycle,
In the steady state, it is determined whether the input current to the coil is a continuous mode in which the current is not interrupted during the control cycle or a discontinuous mode in which the current is interrupted in the middle of the control cycle, and PWM control is performed using a different algorithm based on the determination result.
A digital converter that performs PWM control by correcting an original control time calculated by the algorithm in a transient state up to the steady state. The digital converter according to claim 2, wherein the transient state is a time period from when the computer circuit starts to operate until a predetermined time elapses. 3. The digital converter according to claim 2, wherein the transient state is a time period from when the PWM control is stopped and the output DC voltage is lowered to fall below a downward limit value until a predetermined time elapses. 3. The digital converter according to claim 2, wherein in the transient state, a correction algorithm that approximates the original control time as time passes is adopted. A method for controlling a digital converter that realizes the operation according to claim 1.